4 Key Steps to Design a NCL30188-Controlled LED Driver

AND9451/D
4 Key Steps to Design
a NCL30188‐Controlled
LED Driver
Description
This paper proposes the key steps to rapidly design
a NCL30188-driven flyback converter to power an LED
string. The process is illustrated by a practical 10 W,
universal mains application:
• Maximum Output Power: 10 W
• Input Voltage Range: 90 to 265 V rms
• Output Voltage Range: 12 to 20 V dc
• Output Current: 500 mA
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APPLICATION NOTE
Introduction
• Cycle-by-cycle Peak Current Limit: when the current
The NCL30188 is a driver for power-factor corrected
flyback, non-isolated buck-boost and SEPIC converters.
The current-mode, quasi-resonant architecture optimizes
the efficiency by turning on the MOSFET when the
drain-source voltage is minimal (valley). At high line, the
circuit delays the MOSFET turn on until the second valley
is detected to reduce the switching losses. An internal
proprietary circuitry controls the input current in such a way
that a power factor as high as 0.99 and an output current
deviation below ±2% are typically obtained without the
need for a secondary-side feedback. The circuit further
contains a suite of powerful protections to ensure a robust
LED driver design without the need for extra components or
overdesign. Among them, one can list:
• Over Temperature Thermal Fold-back: connecting
a NTC to the SD pin allows for gradual reduction of the
LED current down to 50% of its nominal value when the
temperature is excessive. If the current reduction does
not prevent the temperature from reaching a second
level, the controller stops operating (SD_OTP).
• Over Voltage Protection (SD_OVP): A Zener diode can
further be used on the SD pin to provide an adjustable
OVP protection (SD OVP).
© Semiconductor Components Industries, LLC, 2016
June, 2016 - Rev. 0
•
•
•
•
sense voltage exceeds the internal threshold (VILIM ),
the MOSFET immediately turns off (cycle-by-cycle
current limitation).
Winding and Output Diode Short-circuit Protection
(WODSCP): an additional comparator stops the
controller if the CS pin voltage exceeds (150% ⋅ VILIM )
for 4 consecutive cycles. This feature can protect the
converter if a winding or the output diode is shorted or
simply if the transformer saturates.
Output Short-circuit Protection (AUXSCP): If the ZCD
pin voltage remains low for a 90-ms time interval, the
controller stops pulsating until 4 seconds has elapsed.
Open LED Protection: if the VCC pin voltage exceeds
the OVP threshold, the controller shuts down and waits
4 seconds before restarting switching operation.
Floating/Short Pin Detection: the circuit can detect
most of these situations which helps pass safety tests.
Pin−to−pin compatible to the NCL30088, the NCL30188
provides the same benefits with in addition, an increased
resolution of the digital current−control algorithm for a 75%
reduction in the LED current quantization ripple.
1
Publication Order Number:
AND9451/D
AND9451/D
PRELIMINARY REMARKS
Two NCL30188 Versions
There exist two NCL30188 versions. As summarized by
Table 1, they differ in their respective protection mode.
When the Winding and Output Diode Short Circuit
Protection (WOD_SCP) or the Output and Auxiliary
Winding Short Circuit Protection (AUX_SCP) triggers,
the A version latches-off while the NCL30188B enters the
auto-recovery mode. Similarly, the SD over-temperature
and over-voltage protections (SD pin OTP and SD pin OVP)
are latching-off in the NCL30188A and auto-recovery in the
NCL30188B.
Table 1. PROTECTION MODES
AUX_SCP
WOD_SCP
SD Pin OTP
SD Pin OVP
NCL30188A
Latching Off
Latching Off
Latching Off
Latching Off
NCL30188B
Auto-Recovery
Auto-Recovery
Auto-Recovery
Auto-Recovery
In the case of a latching-off fault, the circuit stops pulsing
until the LED driver is unplugged and VCC drops below
VCC(reset) (5 V typically). At that moment, the fault is
cleared and the circuit can resume operation. In the
auto-recovery case, the circuit cannot generate DRV pulses
for the auto-recovery 4-s delay. The circuit recovers
operation when this time has elapsed.
regulation will then be optimal as long as the lowest line
peak voltage is higher than the inductor demagnetization
voltage, i.e.,:
Duty-Ratio Limitation
where (Vin,rms )LL is the lowest-line rms voltage (85 or
90 V rms in general) and (Vf ) is the output diode forward
voltage.
The NCL30188A/B duty-ratio is internally limited to
50% at the top of the lowest line sinusoid. Output current
• If
ǒǸ2 @ ǒV
• If
ǒǸ2 @ ǒV
Ǔ
Ǔ
w V out ) V f with non-isolated converters,
Ǔ
np
w n ǒV out ) V fǓ in flyback applications,
in,rms LL
in,rms LL
Ǔ
s
Table 2. NCL30188 CONDITIONS OF USING
Output Voltage Range for
Non-Isolated Converters (Note 1)
NCL30188A (Note 2)
NCL30188BA
Output Voltage Range for
Flyback Converters (Note 1)
V out ) V f v Ǹ2 @ ǒV in,rmsǓ
LL
n
V out ) V f v n s @ Ǹ2 @ ǒV in,rmsǓ
p
LL
V out ) V f v Ǹ2 @ ǒV in,rmsǓ
LL
n
V out ) V f v n s @ Ǹ2 @ ǒV in,rmsǓ
LL
p
1. (Vin,rms)LL is the lowest-line rms voltage (e.g., 85 V rms), (Vf), the output diode forward voltage.
2. Please contact local sales representative for availability.
As an example, let’s assume that we must design a 90 to
265 V rms, non-isolated buck-boost converter. For optimal
control accuracy, the LED driver output voltage should not
exceed:
V out,max + Ǹ2 @ ǒV in,rmsǓ
LL
* Vf ^
^ Ǹ2 @ 90 * 1 ^ 126 V
If the duty-ratio limitation is exceeded by your
application, the LED current will be below its nominal value
at the lowest line voltage but will meet the target when the
input voltage level is sufficient. By the way, a symptom of
the duty-ratio limitation effect can be observed as shown by
Figure 1 where the input current is clamped by the
over-current protection during normal load conditions.
(eq. 1)
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2
AND9451/D
V
CC
I LED
The current is
MOSFET current
clamped to
(V
/ R
ILIM
sense
Figure 1. Current Over-Current Limitation
(VILIM is Over-Current Threshold, RSENSE the Current Sense Resistor)
Our application of interest is a flyback converter. Note that
in this case, turns ratio provides some flexibility which can
help meet the condition of using.
LED Driver Dimensioning
D1
RSTUP
Aux
.
RC
RS1
.
DC
RZCD1
COUT
.
CC
NCL30188
RS2
RZCD2
CZCD
CCOMP
CSD
RTH
1
8
2
7
3
6
4
5
DZ
Q1
RLFF
C2
Figure 2. Basic Schematic
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3
C1
CCS
RSENSE
)
AND9451/D
STEP 1: POWER COMPONENTS SELECTION
This leads in our case to:
Basically, the transformer, the output capacitor and the
power silicon devices are dimensioned “as usual”, that is, as
done with any other PF corrected, quasi-resonant flyback
converter. This chapter does not detail this process, but
highlights the major points.
n AUX v n S @ 25.5 ^ 1.3 @ n S
20 ) 1
(eq. 4)
Practically, we will select (nAUX = nS).
In this case, VCC will be in the range of Vout , with some
deviations due to the imperfect coupling.
Transformer Selection
Selecting the Auxiliary Winding Number of Turns
An auxiliary winding is necessary for zero current
detection and to provide the VCC voltage. The output voltage
of a LED driver generally exhibits a large range. The VCC
voltage provided by the auxiliary winding will vary in
a similar manner. The NCL30188 features a large VCC range
to address these variations. Practically, after start-up,
the operating range is 9.4 V up to 25.5 V.
NOTE: (VCC(OVP) )min = 25.5 V is the threshold
minimum value of the VCC over-voltage
protection. This safety feature protects the
circuit if the LED string happens to be
disconnected.
Selecting the Secondary Winding Number of Turns
In general, NPS , the secondary to primary transformer
turns ratio (NPS = nS / nP) [nP designates the primary number
of turns, nS , the secondary number of turns] is selected as
low as possible so that the input current stress is reduced.
NPS cannot be too small however. NPS sets the amount of
voltage reflected during the off-time (see Figure 3) and
hence, must be high enough to limit the voltage stress across
the primary-side MOSFET. Indeed, the voltage to be
sustained by the primary-side MOSFET and the output
diode are:
The auxiliary winding number of turns can be selected so
that the auxiliary voltage is slightly below (VCC(OVP))min
when the output voltage is at a maximum factoring in impact
of the 100/120-Hz ripple. Practically, this criterion turns
into:
V diode,max + N PS Ǹ2 (V in,rms) max ) V out ) V f ) V D*ov
n AUX
n S (V out,max ) V f) v (V CC(OVP)) min ) V f
V DS,max + Ǹ2 (V in,rms) max )
(V CC(OVP)) min ) V f
V out,max ) V f
(eq. 5)
Where:
• NPS is the secondary to primary transformer turns ratio
NPS = nS / nP
• VQ−ov is the MOSFET overvoltage shown in Figure 3.
This overshoot is due to the leakage inductor reset. It is
limited by the clamping network consisting of DC , CC
and RC of Figure 2.
• VD−ov is a similar overshoot that occurs across the
output diode when the MOSFET turns on.
(eq. 2)
Hence:
n AUX v n S @
V out ) V f
) V Q*ov
N PS
(eq. 3)
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4
AND9451/D
The clamping network is often designed so that VQ−ov is
between 50% and 100% of the reflected voltage:
V Q*ov + k C @
v out ) V f
with 0.5 v k c v 1.0
N PS
V ds,max + Ǹ2 @ (V in,rms) HL )
N PS
(eq. 7)
Some derating is generally requested. The
typically-applied 15% safety factor implies that the
MOSFET voltage does not exceed 85% of its breakdown
voltage. Hence:
(eq. 6)
We can estimate the maximum voltage reached on the
drain node, considering Vout(OVP) level as the maximum
output voltage:
V ds,max + Ǹ2 @ (V in,rms) HL )
(1 ) k c)(V out(OVP) ) V f)
(1 ) k c)(V out(OVP) ) V f)
N PS
Where VDSS is the MOSFET breakdown voltage.
v 85%V DSS
(eq. 8)
85%V DSS* Ǹ2 @ (V in,rms) HL
nP
v
nS
(1 ) k c)(V
)V)
Finally:
out(OVP)
(eq. 9)
f
Figure 3. MOSFET Drain-source Voltage (Yellow Trace) and Current (Blue)
In our application, (Vin,rms )HL is 265 V rms and
(Vout(OVP) + Vf ) is about 28 V.
With a 600 V MOSFET,
nP
85% @ 600* Ǹ2 @ 265
^ 4.8
n S (1 ) k c) v
28
f sw +
nP
85% @ 800* Ǹ2 @ 265
^ 10.9 (eq. 11)
n S (1 ) k c) v
28
We select the second option (800 V MOSFET) with
nP
n S + 6 and (kc = 80%), which meets Eq. 11 requirements
n
since nP (1 ) k c) + 6 @ 180% + 10.8 v 10.9 .
S
It can be easily checked that the Table 2 condition of using
the A or B version is met. The NCL30188B will be used for
this application.
ǒ
Ǔ
Ǔ
2
(eq. 12)
The switching frequency is a rising function of the rms
line voltage. At a given line magnitude, the switching
frequency is yet higher near the line zero crossing and
decays as the line voltage rises due to the (nin (t)) term.
Note that when high-line conditions are detected (see
NOTE), the NCL30188 does not operate in quasi-resonant
mode but delays the MOSFET turn on until the 2nd valley is
detected (see Figure 4). This reduces the switching
frequency upper range and optimizes the high-line
efficiency.
NOTE: The input voltage is sensed by the VS pin for
brown-out protection, feedforward and line
range detection. High-line conditions are
detected when the VS pin voltage exceeds 2.4 V
typically. See data sheet for more details.
(eq. 10)
With a 800 V MOSFET,
ǒ
ǒ
(V in,rms) 2
V out ) V f
@
2L PP in,avg
N PS n in(t) ) V out ) V f
Ǔ
Selecting the Primary Inductance
Assuming a quasi-resonant operation and neglecting the
small delay necessary for detecting the MOSFET
drain-source valley, the primary inductance dictates the
switching frequency as follows:
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AND9451/D
Figure 4. Quasi-Resonant Mode in Low Line (Left), Turn On at Valley 2 when in High Line (Right)
The primary inductor will be selected with respect to the
targeted switching frequency range, keeping in mind that:
• High switching frequency levels reduce the size of the
storage elements
• Conversely, increasing the switching frequency leads to
more switching noise and losses. Also, EMI filtering
may be tougher because of the EMI generated at the
switching frequency and close harmonic levels. Most
power supplies have to meet standards which apply to
frequencies above 150 kHz. That is why SMPS
designers often select FSW = 130 kHz to keep the
fundamental component below 150 kHz and then out of
the regulation scope. Often, 65 kHz is also chosen to
not to have to damp harmonic 2 too.
•
nominal line voltage when ǒ n in(t) + V in,pkń2 Ǔ
Our application is a wide-range one.
Let us compute LP so that at 115 V rms, the switching
frequency is below fsw,T = 65 kHz:
ȡ
ȧ
ȢN
(P in,avg) max
(V in,rms) LL
(P in,avg) max
(I L,pk) max + 2
Ǹ3 (V in,rms) LL
LP w
ǒ
@ 1)
Ǹ
ǒ
Ǹ
(I L,rms) max +
2 12
@
@
Ǹ3 90
1)
1)
[email protected]
in,rms
PS
115 2
2 @ 65 @ 10 3 @ 12
@
2
ǒ
) V out
12 ) 1
[email protected]
[email protected]
ȣ
ȧ
) VȤ
(eq. 13)
f
Ǔ
) 12 ) 1
2
^ 2 mH
(eq. 14)
Finally we have to consider the primary current
magnitude constraints:
90
6
N PS(V in,rms) LL
V out ) V f
Ǔ
16 Ǹ2 @ (V in,rms) LL
1)
In our application, assuming an efficiency of 84%
((Pin,avg )max = 12 W), Eq. 15 and Eq. 16 lead to:
12
(I L,pk) max + 2 Ǹ2 @
@
90
V out ) V f
Which leads to:
in,pk
(I L,pk) max + 2 Ǹ2 @
2
(V in,rms) 2
LP w
@
2f sw,TP in,avg
As the rule of thumb, let us select LP as follows:
• In wide mains application: choose LP so that the
switching frequency is below 65 kHz at the low-line
range nominal voltage (typically 115 V rms) over
a large part of the sinusoid. Practically, we can select
that the frequency target will have to meet starting from
(V ń2) that is ǒǸ2 @ 115ń2Ǔ. This arbitrary choice
relies on the idea that for below this line voltage level
the input current is relatively small and easy to filter.
Check that at the high-line nominal voltage (230 V rms
typically), the switching frequency stays below 65 kHz
thanks to the valley-2 operation.
Similarly, in a narrow mains operation case, select LP
so that the switching frequency is below 65 kHz at the
3p @
Ǔ
20 ) 1
V out)V
N
PS
f
(eq. 15)
)
6p @ (V in,rms) LL
4@
ǒ
V out)V
N
PS
^ 0.65 A
16 Ǹ2 @ 90
6p @ 90 2
)
^ 350 mA
3p @ 6 @ 21 4 @ (6 @ 21) 2
We selected transformer 750871144 from Wurth
Elektronik with the following characteristics: LP = 1.9 mH,
nP / nAUX = nP / nS = 6.
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6
Ǔ
2
2
(eq. 16)
f
(eq. 17)
(eq. 18)
AND9451/D
Power Switches
a circuit requires a diode, a resistor and a capacitor (DC , RC
and CC of Figure 2):
• The capacitor CC absorbs the leakage inductor energy
when the MOSFET turns off. The voltage rating of this
capacitor must be equal to the MOSFET breakdown
voltage or higher. Note that similarly to the snubber
capacitor, the CC negative terminal should preferably
be connected to the MOSFET source rather than to
ground.
• The resistor RC loads CC to ensure that the CC voltage
does not drift up but stabilize at a level which ensures
a proper MOSFET protection (the MOSFET voltage is
clamped to the CC voltage by means of DC ).
• The diode DC prevents CC from discharging when the
MOSFET turns on. This diode is generally a
fast-recovery diode. A low-value resistor (R14 of
Figure 9) is inserted to limit the current spike which
otherwise occurs when the MOSFET turning off, CC
abruptly charges. Do not oversize this series resistor.
The leakage current flowing through it creates a voltage
drop. The MOSFET voltage is clamped to the CC
voltage PLUS the series resistor voltage. In our case,
the maximum leakage current is (V ILIMńR sense) that is
(1Vń1.5W ^ 667mA). In our design, a 22 W
resistor is implemented that results in an overshoot less
than (22W @ 0.667 ^ 17V).
MOSFET
The voltage constraints on the MOSFET have been
discussed in the transformer section. Conduction losses
depend on the MOSFET rms current which can be computed
with the following equation:
(P in,avg) max
(I Q,rms) max + 2
Ǹ3 (V in,rms) LL
Ǹ
1)
8 Ǹ2 @ (V in,rms) LL
3p @
V out)V
N
f
PS
(eq. 19)
A NDD03N80 MOSFET is selected (DPAK, 800 V,
4.5 W).
Output Diode
Similarly, the voltage constraints have been discussed in
the previous section.
Losses are mainly produced by the average current
flowing through the diode. This average is simply the LED
current (0.5 A in our case).
A 200-V, 2-A SMB diode is selected (MURS220).
Snubber and Clamping Network
A snubber capacitor can be placed across the MOSFET to
reduce the dV/dt and lower the switching noise. A 47-pF,
1,000-V capacitor is placed in our application (C2 of
Figure 9). Note that for optimal operation, it is
recommended to connect the snubber capacitor between the
MOSFET drain and source terminals rather than between
drain and ground.
When the MOSFET turns off, the magnetizing inductor
energy is conveyed to the secondary side and charges the
output. The leakage inductor current cannot be used by the
output. It must be diverted from the MOSFET. If not, the
MOSFET drain-source voltage would rise to destructive
levels. A clamping network is hence necessary. Such
(1)k c )@(V
)V )
f
out(OVP)
N
PS
ER +
@
Eq. 7 gives the maximum voltage
(1 ) k c)(V out(OVP) ) V f)
Ǹ2 @ (V
in,rms) HL )
N PS
the MOSFET must sustain. This voltage is the CC maximum
voltage. Hence, the energy that RC can consume and
re-inject to the input over a switching cycle is given by:
ǒ
ǒ
Ǔ
(1)k c )@(V
)V )
f
out(OVP)
N
PS
Ǔ
) Ǹ2 @ (V in,rms) HL @ T SW
(eq. 20)
RC
C
The energy due to the leakage inductor must be handled.
V ILIM
It is maximal when the primary current exceeds its
R sense
limit where VILIM is the over-current protection threshold
and Rsense , the current sense resistor:
ǒ Ǔ
ǒ
V ILIM
EL
+ 1 L leak
leak
2
R sense
Ǔ@
2
(1)k c )@(V
)V )
f
out(OVP)
N
PS
(1)k c )@(V
)V )
(V
)V )
f
f
out(OVP)
out(OVP)
*
N
N
PS
PS
+
This energy of Eq. 20 must be equal or higher than the
leakage inductor energy defined in Eq. 21. From this, we can
deduce the following minimum RC value:
(V
RC v
)V )
f
out(OVP)
N
PS
@
ǒ
(1)k c )@(V
1
[email protected]
)V )
f
out(OVP)
N
PS
C
L leak
ǒ
V
ILIM
R sense
Ǔ
@ f SW
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7
) Ǹ2 @ (V in,rms) HL
2
ǒ
1 ) kC
V ILIM
L
2 @ k C leak R sense
Ǔ
2
(eq. 21)
Ǔ
(eq. 22)
AND9451/D
Ǹ
Select the CC capacitor so that the time constant (RC ⋅ CC )
is large compared to a switching period, practically, in the
range of 1 ms.
Since in our application, we have selected (kC = 80%), it
comes:
RC v
28 @ 6 ǒ(180% @ 28 @ 6) ) Ǹ2 @ 265Ǔ
1
[email protected]%
2
1 Ǔ
20m ǒ1.5
@ 65k
ȡ
ȧǒ
Ȣ
C out,min +
^ 315 kW
(180% @ 28 @ 6) 2
^ 290mW
315k
Two 470 kW, 1/2W resistors are placed in parallel for the
following effective resistance: (470 k || 470 k = 235 kW).
A 4.7 nF/100 V capacitor is implemented for CC that with
RC forms a 1.1 ms time constant.
This resistor will dissipate
The power delivered by PFC converters exhibits a large ac
component at twice the line frequency. To some extend, the
output capacitor compensates for it but yet, the output
current exhibits some ripple inversely proportional to the
capacitor value (Cout ).
Below equation expresses the current ripple:
C out,min +
Ǹ
S
2
@
(eq. 26)
Ǹǒ Ǔ *1
2
2
1
2 @ 100p @ 6
^ 460 mF
(eq. 27)
A 470 F/35 V is implemented.
Bulk Capacitor Heating:
It must also be checked that the ESR is low enough to
prevent the rms current that flows through it, from
overheating the bulk capacitor. This capacitor rms current
can be estimated using the following expression.
From Eq. 24, the following minimum value for Cout can
be deduced (Eq. 25):
P
(eq. 25)
4p @ f line,min @ R LED,min
In our application the minimum LED dynamic resistance
is estimated to be 6 W and the minimum line frequency is
50 Hz. In this case, the minimum output capacitor value is:
(eq. 24)
ȡ32 Ǹ2 n
ȧ 9p @ ǒn Ǔ
Ȣ
pk*pk
I out,nom
(DI out) pk*pk
+1
I out,nom
Output Capacitor
(I C,rms) max +
2
DI outǓ
Cout must then be large enough to avoid an excessive
current ripple which could reduce the LED reliability. The
flicker index is commonly specified below 0.15. This
requirement corresponds to a 100% peak-to-peak ripple in
a PF-corrected LED driver with a sinusoidal output current
shape
This criterion (100% peak to peak ripple), leads to:
(eq. 23)
(DI out) pk*pk
2
+
I out,nom
Ǹ1 ) (4p @ fline @ R LED @ C out)2
2
ȣ
ȧ*1
Ȥ
(P in,avg) max
V in,rms @
2
V out)V
N
PS
ȡ 9p V ȣȣ
ȧ1 ) 16 Ǹ2 @ ȧȧ*I
Ȣ
ȤȤ
2
f
in,rms
V out)V
f
N
PS
out,nom
2
(eq. 28)
It remains wise to check the output capacitor heating in the
lab.
STEP 2: OUTPUT CURRENT SETTING
As explained in the data sheet, the output current is
regulated to equal the following Iout,nom nominal output
current:
V REF
I out,nom +
2N PSR sense
R sense +
V REF
2N PSI out,nom
(eq. 30)
The power dissipated by Rsense can be computed by the
following equation:
(eq. 29)
ǒ Ǔ ȡȧ
P in,avg
P Rsense + 4 R sense
3
V in,rms
Where:
• NPS is the secondary to primary transformer turns ratio
NPS = nS / nP
• Rsense is the current sense resistor (see Figure 2)
• VREF is the output current internal reference.
In our application:
• NPS = 1/6
• Iout,nom = 500 mA
• (Pin,avg )max = 12 W
• (Vin,rms )LL = 90 V
• Vout,min = 10 V
Hence once the transformer is designed, NPS is known and
the only current sense resistor dictates the output current
level.
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2
1)
Ȣ
ȣ
ȧ (eq. 31)
Ȥ
8 Ǹ2 @ V in,rms
V out
3p @ N
PS
AND9451/D
Hence:
R sense +
10 *3
250 @
+ 1.5 W
2 @ 16 @ 500 @ 10 *3
And:
ǒ Ǔ
P Rsense + 4 1.5 12
90
3
2
ǒ
1)
Ǔ
8 Ǹ2 @ 90
10
3p @ 1ń6
R S2
@ Ǹ2 (V in,rms) BOH + V BO(on)
R S1 ) R S2
(eq. 32)
Where VBO(on) is the internal threshold (1 V typically) the
VS pin voltage must exceed to allow circuit operation.
In other words,
^ 100 mW
R S1 + R S2
(eq. 33)
Input Voltage Sensing and Feedforward:
A portion of the input voltage must be applied to the VS pin
to provide the circuit with the sinusoidal reference necessary
for shaping the input current (PFC). The obtained current
reference is further modulated so that when averaged over
a half-line period, it is equal to the output current reference
(VREF ). This averaging process is made by an internal
Operational Trans-conductance Amplifier (OTA) and the
capacitor connected between the COMP pin (pin3) and
ground. The recommended minimum COMP capacitance is
1 mF.
R S1 + 47 @ 10 3 @
(eq. 35)
ǒ
Ǔ
Ǹ2 @ 81
*1 [ 5.4 MW
1
(eq. 36)
Feedforward
The NCL30188 computes the current setpoint (Vcontrol )
for power factor correction and proper regulation of the LED
current. Now, the MOSFET cannot turn off at the very
moment when the current-sense voltage exceeds Vcontrol .
There actually exists a propagation delay tprop (Figure 5) for
which the primary current keeps rising. As a result, the
primary current does not exactly peak to the expected
(Vcontrol / Rsense ) value but to a higher level. The output
current is hence also affected. Optimal regulation
performance requires the peak current increase caused by
tprop to be compensated.
Input Voltage Sensing
A resistors divider (RS1 and RS2 of Figure 2) provides
pin 2 with the VS signal. The scale-down factor is computed
in accordance with the brown-out protection. If (Vin,rms )BOH
is the targeted minimum line rms voltage necessary for
entering operation, RS1 and RS2 must comply with:
Rsense
Ǔ
Ǹ2 @ (V
in,rms) BOH
*1
V BO(on)
It is generally recommended not to have a single resistor
placed between a high-voltage rail and a low potential node.
Instead, two or more resistors are to be placed in series. In
our case, we use two 2,700 kW resistors for RS1.
COMP Pin Capacitor
A 1 mF capacitor is to be placed between COMP pin and
ground.
Vcontrol
ǒ
RS2 values in the range of 50 kW generally provide a good
tradeoff between losses and noise immunity. In our
application, we select 47 kW. Our system being supposed to
enter operation when the line voltage exceeds 81 V rms:
Two 3 W resistors are placed in parallel.
IL
(eq. 34)
DIL,pkH
High
Line
DIL,pkL
Low
Line
tprop
tprop
Figure 5. Propagation Delay Effect on Peak Current
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9
time
AND9451/D
However, for a first approximation, we can calculate
RLFF, using tprop = 200 ns.
Then, the offset resistor value can be fine-tuned on the
bench so that the output current characteristic is nearly flat
over the line voltage range.
Using Eq. 39, we can calculate the first value of RLFF for
our design:
The NCL30188 compensates for the propagation delay by
sourcing a current proportional to the VS pin voltage out of
the CS pin during the on-time. Placing a resistor RLFF
between the CS pin and the sense resistor, the following
offset is hence obtained:
V CS(offset) + K LFF n S (t)R LFF
(eq. 37)
Where the VS pin voltage vS (t) equates:
n S(t) +
R S2
n (t)
R S1 ) R S2 in
ǒ
R LFF + 1 )
(eq. 38)
ǒ
R S1
R S2
Ǔ t L RK
prop
P
Sense
LFF
prop
P
Sense
LFF
Ǔ
+ 1 ) 5400 200n
47 1900m
Ǔ
R LFF + 1 )
Ǔ t L RK
ǒ
Since the CS pin offset must compensate for
R sense @ n in(t) @ t prop
R sense @ DI L,pk +
, the offset resistor
LP
value can be computed as follows:
ǒ
R S1
R S2
+
(eq. 40)
1.5 ^ 915W
20m
After experiments in the lab, RLFF value was decreased to
820 W.
Important Note: As indicated in the NCL30188 data sheet,
RLFF must be selected higher than 250 W.
If not, the circuit may improperly detect
that the CS pin is grounded.
(eq. 39)
Where:
• KLFF is the VS pin voltage to CS pin current conversion
ratio. Its typical value is 20 mS.
• RS1 and RS2 are the input voltage sensing resistors (see
Figure 2).
Selecting the CS Pin Capacitor
The shape of the current-sense voltage influences the
output current regulation. If the CS pin filter (RLFF, CCS ) is
too big, the output current setpoint will vary (Iout higher than
expected value). Thus, once RLFF has been chosen, it is
important to keep the value of CCS as small as possible to
have an optimal output current regulation. CCS should be in
the range of 10–100 pF.
Finally: (see Table 3)
Parameter tprop includes the controller internal delay of
the controller (about 50 ns) and the MOSFET turning off
time. Thus, it varies with respect to the chosen MOSFET and
the way it is driven (value of the gate resistors for instance).
As a consequence, it is difficult to predict its exact value
prior to evaluating the LED driver design.
Table 3.
RS1
RS2
CCOMP
RSENSE
RLFF
CCS
5400 kW (two 2.7 MW
placed in series)
47 kW
1 mF
1.5 W (two 3 W
resistors in parallel)
820 W
−
• The output current value depends on the sense resistor
NCL30188 proprietary regulation technique ensures
a very precise LED current control.
Please note that sources of deviation are however to be
considered. They are detailed in [3]. Let’s recall the main
points:
• The NCL30188 regulates the total current provided by
the converter, that are, the LED current plus the VCC
current. Hence, the actual output current is:
I out,nom +
In general, the
N
N P @ V REF
* Aux I CC
NS
2 @ N S @ R sense
ǒNN
Aux
S
•
(eq. 41)
(Rsense ). Select a precise resistor and avoid long tracks
that lead to an additional series resistance. If Rsense is
1 W and that the circuit additionally senses the voltage
across a 20 mW track, the total sensing resistor will be
1.02 W instead of 1 W. Ultimately, the output current
will 2% below target.
Avoid inductive sense resistor. If not, the output current
will be less than the target because of the offset the
series inductor causes on the CS pin voltage:
l Rsense
@ n in(t) where l Rsense is the Rsense parasitic
LP
inductance.
ǒ
Ǔ
I CC term is small compared to the
target LED current and can be ignored. If not, Rsense should
be reduced to compensate for the circuit consumption.
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10
Ǔ
AND9451/D
STEP 3: SD PIN MANAGEMENT
The Thermal Foldback and Shutdown block of the
NCL30188 is inherited from the NCL30082 and its
functioning and design is detailed in application note
AND9131/D ([2]). Only key points will be highlighted here.
voltage at a low enough level. This is not the case in our
application. No Zener diode is hence implemented.
Selecting the Thermistor
The resistance of a Negative Coefficient Temperature
thermistor (NTC) reduces when its temperature rises. An
NTC is to be placed between the SD pin and ground to detect
an over-temperature condition. In response to a high
temperature, the circuit gradually reduces the LED current
down 50% of its nominal value. If despite the current
reduction, the temperature still increases, the circuit will
eventually stop operation. In general a 50% reduction in
current is more than a 50% drop dissipated power as the LED
forward voltage will decrease as the current is folded back.
More specifically, as shown by Figure 6, Rth designating
the NTC resistance:
• The circuit starts to gradually reduce the output current
when Rth drops below RTF(start) and continues
diminishing it until Rth goes below RTF(stop).
• At that moment, it maintains the output current at 50%
of its nominal level as long as Rth is between RTF(stop)
and ROTP(off). If on the contrary, a temperature decay
leads Rth to rise above RTF(stop), the current increases
according the precedent characteristics. If Rth exceeds
RTF(start), full current capability is recovered.
• The LED driver totally stops operating if Rth drops
below ROTP(off) and stays off until the temperature
having reduced, Rth exceeds ROTP(on). At that moment,
the circuit resumes (NCL30188B only – the A version
latches off) and delivers 50% of the nominal current.
• If Rth further rises, the current regulation grows as well
until Rth reaches RTF(start). At that moment, the LED
driver provides the full current.
Selecting the SD Over-voltage Zener Diode
A Zener diode can be placed between the VCC and the SD
pins. The circuit detects an OVP fault if the SD pin voltage
exceeds 2.5 V. Note that the NCL30188 ensures that
a 700 mA minimum current flows through the Zener diode
in this case (see [1]) so that it can be operated far from its
knee region. The SD OVP threshold on VCC is:
(V CC) SD,OVP + V Z ) V OVP
(eq. 42)
Where VOVP is the 2.5-V SD OVP threshold.
An SD OVP fault is detected if VCC exceeds (VCC )SD,OVP.
For instance, if you applied a Zener diode exhibiting an 18 V
Zener breakdown voltage (when biased by a 700 mA
current), the SD_OVP protection will trip when VCC
exceeds (18.0 + 2.5) volts, that is, 20.5 V. In this case, the
NCL30188B stops operating for the auto-recovery 4 s delay.
At the end of this time, the circuit attempts to resume
operation. If the fault is still present, the circuit again detects
an SD OVP fault and stops for 4 s. Finally, the NCL30188B
enters a safe, very low duty-ratio burst mode. An SD OVP
fault leads the NCL30188A to latch off until the LED driver
is unplugged and VCC drops below VCC(reset). At that
moment, the fault is cleared and the circuit can resume
operation.
Such a programmable protection feature is useful if the
fixed VCC OVP protection which trips when VCC exceeds
VCC(OVP) (26.8 V typically) does not clamp the output
Figure 6. Thermal Foldback Characteristics and Over-temperature Protection
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11
AND9451/D
capacitor must not exceed 4.7 nF so that it can charge to its
nominal level before the OTP blanking time has elapsed.
NOTE: At start-up, the controller blanks the SD
function until a delay of 250 ms minimum (OTP
blanking time), has elapsed, to provide CSD with
enough time to properly charge above the 0.5 V
over-temperature threshold. If not, the low SD
pin voltage will be considered as caused by the
low-resistance of an NTC in excessive
temperature conditions.
As an example, if thermistor NB12P00104JBB from AVX
is implemented:
• The circuit starts to reduce the output current at about
82°C ambient temperature.
• The circuit stops operation at about 104°C ambient
temperature.
• The circuit recovers operating at about 90°C ambient
temperature.
Selecting the SD Pin Capacitor
A small capacitor can be placed between the SD pin and
ground to prevent the pin from picking up possible
surrounding noise. Please note that the value of this
Finally: (see Table 4)
Table 4.
DZ
Rth
CSD
N/A
NB12P00104JBB (AVX)
1 nF
STEP 4: AUXILIARY WINDING AND VCC MANAGEMENT
VCC Capacitor Refueling
VCC Capacitor Value and Startup Circuitry
In nominal operation, the auxiliary winding provides the
VCC voltage as shown by Figure 2. The auxiliary winding
number of turns (naux ) is computed in the transformer
section of the “Step 1” paragraph. Note that during the
on-time, diode DAUX of Figure 2 rectifies the auxiliary
voltage to provide VCC . Hence, neglecting the turn on spike,
DAUX must be able to sustain:
VD
AUX
+ V CC )
ǒnn
AUX
P
@ Ǹ2 @ (V in,rms) HL
Ǔ
When off (that is until VCC has reached the 18-V start-up
level), the NCL30188 consumes a very low current (13 mA
typically, 30 mA maximum). Thus, high-impedance, low
dissipation, resistors can be used to charge the VCC capacitor
at start-up.
Note however, that faults like a VCC over-voltage
condition lead the LED driver to stop operation and refrain
from attempting to recover until a 4 s delay is elapsed. A low
duty-ratio burst mode of operation is hence obtained as long
as the fault is present. VCC cycles up and down in such a case.
For this time, the (off-mode) consumption is slightly higher
(75 mA max.). It is hence recommended to have the startup
current (Istartup in Figure 7) above 75 mA. If not, VCC may
collapse and the circuit reset before the 4 s delay has elapsed.
As detailed in application note AND9131/D [2],
the startup resistor Rstartup can either be connected to the
bulk rail or to half-wave (Figure 7). Connecting the startup
resistor to the half-wave allows decreasing the power
dissipated in the startup resistor.
(eq. 43)
The VCC highest value is the maximum voltage the
VCC(OVP) threshold can take (28.5 V). Therefore:
VD
AUX
+ (V CC(OVP)) max )
ǒnn
AUX
P
@ Ǹ2 @ (V in,rms) HL
Ǔ
(eq. 44)
In our case:
VD
AUX
+ (V CC(OVP)) max )
ǒ
ǒnn
AUX
P
Ǔ
Ǔ
@ Ǹ2 @ (V in,rms) HL ^
^ 28.5 ) 1 @ Ǹ2 @ 265 ^ 91 V
6
(eq. 45)
Due to the turn on spike, some significant headroom is
necessary. Selecting a diode exhibiting at least twice the
computed VRRM value seems a good practice.
A 250 V/0.2 A BAV21 diode is implemented in our
application.
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12
AND9451/D
Figure 7. The Startup Resistor can be Connected to the Bulk Rail or to the Half Wave
Calculating the VCC Capacitor
output capacitor (no current flows through the LED string).
treg lasts until the output voltage reaches the level at which
VCC starts to be charged. In general, we try to minimize the
CVcc capacitor by allowing a nearly maximal VCC capacitor
discharge, that is, down to a value close to the UVLO level.
At that moment, the output voltage will nearly be
n
V out + (V CC(off)) max n s and treg can then computed as
aux
follows:
The VCC capacitor value (CVcc ) must be large enough to
feed the controller until the auxiliary winding voltage Vaux
is sufficiently large to supply the controller. The time
duration where the controller is supplied by the only CVcc
capacitor is noted treg (Figure 8).
The circuit enters operation when the VCC capacitor is
charged to the VCC(on) startup level. For the treg duration, the
VCC capacitor must be able to maintain the VCC voltage
above the UVLO level (VCC(off)) while providing the current
consumed by the circuit (ICC2 specified in the data sheet)
and the current necessary to drive the MOSFET.
We can estimate treg by considering that for this period of
time, all the LED driver output current is absorbed by the
C Vcc w
ǒ
Ǔ
t reg ^
ǒ
C out
n
(V CC(off)) max n s
aux
I out
Ǔ
(eq. 46)
Now, using the minimum value of the UVLO hysteresis
(minimum value of VCC(on) − VCC(off)), the minimum VCC
capacitor value comes:
(I CC2 ) Q gf sw)t reg
) Q qf sw) (V CC(off)) max
(I
ns @ C
^ n out @ CC2
@
aux
I out
(V CC(HYS)) min
(V CC(HYS)) min
(eq. 47)
• Qg is the MOSFET total gate charge
• (VCC(HYS) )min is the UVLO hysteresis minimum value
Where:
• ICC2 is the NCL30188 consumption at 65 kHz when the
DRV pin is unloaded (4 mA max)
(8 V)
Figure 8. VCC Waveform during Startup
Once the VCC capacitor value is known, the start-up
current needed to charge CVcc can be computed as a function
of the maximum acceptable start-up time if specified. Recall
anyway that this startup current should not be less than
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13
AND9451/D
75 mA so that as explained at the beginning of this section,
the circuit does not reset in fault mode. Hence:
I startup +
(V CC(on)) max C Vcc
(V CC(on)) max C Vcc
) (I CC(start)) max if
) (I CC(start)) max w 75 mA
t startup
t startup
(eq. 48)
I startup + 75 mA otherwise
Where:
• (VCC(on) )max is the VCC startup threshold maximum
value
• (ICC(start) )max is the maximum value of the NCL30188
startup consumption (30 mA)
• tstartup is the targeted startup time
t reg ^
Ǔ
(eq. 49)
C Vcc w
(I CC2 ) Q gf sw)t reg
(4m ) 19n 65k)9m
^
^ 6 mF
8
(V CC(HYS)) min
(eq. 50)
In our case, assuming a 19-nC gate charge MOSFET,
a 65-kHz operation and a 0.5-s target for the startup time, it
comes: (Eq. 49 and Eq. 50)
I startup +
ǒ
470 mF
C out
n
(V CC(off)) max n s +
(9.4 @ 1) ^ 9 ms
aux
I out
500 mA
We will select a 10 mF/35 V capacitor. Hence:
(V CC(on)) max C Vcc
20 @ 10m
) (I CC(start)) max ^
) 30m + 430 mA w 75 mA
t startup
0.5
Startup Resistor Calculation
P startup +
Bulk Connection
For start-up time, the bulk rail sees the line peak voltage
(the input voltage becomes a rectified sinusoid when the
LED driver starts to operate), the following formula gives
the Rstartup value:
Ǹ2 @ (V
in,rms) LL
R startup +
I startup
ǒǸ2 @ (Vin,rms) HL*VCCǓ
2
v
R startup
(eq. 51)
2 @ (V in,rms) HL
R startup
2
(eq. 53)
Where (Vin,rms )HL is the highest line rms voltage.
Half-wave Connection
If the resistor is connected to the half-wave:
(eq. 52)
Ǹ2@(V
in,rms) LL
R startup1ń2 +
Where:
• Istartup is the startup current
• (Vin,rms )LL is the lowest line rms voltage
p
I Cvcc ) I CC(start)
+
R startup
p
(eq. 54)
The maximum power dissipated by the startup resistor
connected to the half-wave is thus:
The maximum power dissipated by the startup resistor
connected to the bulk rail is:
P startup1ń2 +
ǒ
Ǹ2@(V
)
in,rms HL
*V CC
p
Ǔ
2
R startup1ń2
R startup1ń2 +
p
I startup
+
90Ǹ2
p
430m
2
(eq. 55)
p
In Our Application:
We selected the half-wave configuration. Since we have
computed that the startup current had to be 430 mA or more,
we can deduce:
(V in,rms) LL Ǹ2
2
(V in,rms) HL
(V in,rms) HL
v 22 R
+2
p R startup
p
startup
P startup1ń2 v
ǒ
(V
Ǹ2
)
in,rms HL
p
R startup1ń2
Ǔ
2
+
ǒ
265Ǹ2
p
94
Ǔ
2
k ^ 151 mW
(eq. 57)
Three 33-kW, 1/4 W resistors are placed in series.
^ 94 kW (eq. 56)
The power dissipated for the startup resistor at maximum
input voltage is:
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14
AND9451/D
ZCD Network
I ZCD,dmg +
RZCD1 of Figure 2 limits:
• The current injected into the ZCD pin during the
demagnetization time. As indicated in the data sheet,
this current must remain below 5 mA.
• The current extracted from the ZCD pin during the
on-time. This current must not exceed 2 mA.
R ZCD1
v 5mA
(eq. 59)
Where VCC(OVP)max is the VCC maximum value for VCC
OVP protection tripping (28.5 V).
For optimal output current regulation, it is recommended
to keep the ZCD pin voltage below 5 V. This is the goal of
RZCD2 of Figure 2.
R ZCD2 (V CC,max ) V f)
v 5V
R ZCD1 ) R ZCD2
During the on-time, the ZCD pin current is maximal at the
highest line voltage:
Ǹ2 @ (V
n
in,rms) HL
I ZCD,on + naux
v 2mA
R ZCD1
P
V CC(OVP)max ) V f
(eq. 60)
Where VCC,max is the maximal VCC voltage in normal
operation (20 V in our application).
Finally, this resistor together with the CZCD capacitor
delays the zero-voltage crossing event and helps to tune the
turn-on instant when the drain voltage is in the valley.
Finally: (see Table 5)
(eq. 58)
During the demagnetization time, the auxiliary winding
voltage is maximal when VCC is at its maximum value, that
is, the OVP level. Hence:
Table 5.
CVCC
Rstartup1/2
Rstartup
DAUX
RZCD1 / RZCD2
CZCD
10 mF / 35 V
three 33 kW, 1/4 W
resistors in series
N/A
BAV21
33 kW / 10 kW
22 pF
DETAILED SCHEMATIC FOR OUR 10 W, UNIVERSAL MAINS LED DRIVER
NCL30188
Figure 9. Application Schematic
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15
AND9451/D
EXPERIMENTAL DATA
Output Current Regulation
Figure 10 shows the output current as a percentage of its
nominal value. We can see that its characteristic is very flat
with respect to the temperature.
Thermal Foldback starts at about 80°C. As a result, the
output current linearly decays to reach 50% of its nominal
value at nearly 92°C. The circuit stops operating (SD pin
Over Temperature Protection) at approximately 105°C and
resumes operation when the temperature drops down to
about 90°C. These temperature thresholds depend on the
thermistor connected to the SD pin. Below characteristics
were obtained with a NB12P00104JBB thermistor
manufactured by AVX.
110
100
90
Iout / Iout,nom (%)
80
Start of
Thermal
Foldback
70
60
50
40
30
SD pin
OTP
activation
115 Vrms
230 Vrms
20
10
0
−40 −30 −20 −10
0
10
20
30
40
50
60
70
80
90
100 110
TEMPERATURE (°C)
Figure 10. LED Current Characteristics over the Temperature Range
Power Factor
Figure 11 shows the power factor measured at two
different line magnitudes (115 V rms and 230 V rms). The
power factor is extremely stable over the considered
temperature range from −40°C to 80°C. Above 80°C, the
performance is affected by the thermal foldback which
reduces the output current.
1.05
1.00
0.95
PF (−)
0.90
0.85
0.80
0.75
Thermal
Foldback
and
SD pin
OTP
0.70
0.65
115 Vrms
230 Vrms
0.60
0.55
0.50
−40 −30 −20 −10
0
10
20
30
40
50
60
70
80
90
100 110
TEMPERATURE (°C)
Figure 11. Power Factor Performance over the Temperature Range
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16
AND9451/D
application discussed in this document), illustrate the circuit
capability to address shorted/open situations of the LED
string or an output diode failure. Application note
AND9204/D discusses in details the NCL30088 behavior
under safety tests [4]. The NCL30188 only differs from the
NCL30088 in an increased resolution of the digital
current−control algorithm. Thus, this application note also
highlights the NCL30188 protection functions that help pass
the safety tests.
VCC
VRsense
Iline
(100 mA/div)
Open LED String Situation
The LED string being disconnected, the VCC voltage rises
and the VCC(OVP) protection trips when VCC exceeds 26.8 V
(typically). At that moment, the circuit stops operating for
the 4-s auto-recovery delay. The LED driver recovers
normal operation when the LEDs are again connected.
Figure 12. Current Waveform at 115 V rms
Fault Situations
The NCL30188 incorporates a large suite of protections.
Next figures (obtained using the NCL30088B in the
VCC
The LED string is disconnected
Iout (0.5 A/DIV)
Figure 13. The System Enters a Safe Low Duty-ratio Burst Mode when the LED
String is Disconnected (the LED Driver is Unloaded for about 15 s for this Test)
LED String Short Situation
In the case of Figure 13, the only VCC(OVP) protection is
in play. Note that a Zener diode could have been placed
between the VCC and SD pins to select a lower VCC threshold
for fault detection (see the SD pin OVP function in the
NCL30188 data sheet). Further note that if the VCC(OVP)
protection is auto-recovery in all NCL30188 versions, the
SD pin OVP can be a latching-off (NCL30188A) or an
auto-recovery (NCL30188B) protection.
As illustrated by Figure 14, if the output is shorted, the
AUX_SCP protection makes the LED driver enters a safe,
low duty-ratio burst mode. Normal operation is recovered
when the short is removed. Note that the NCL30188A
latches off when an output short is detected. No recovery is
hence possible with these versions until the LED driver is
unplugged and VCC drops below VCC(reset) . At that moment,
the fault is cleared and the circuit can resume operation.
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17
AND9451/D
VCC
Iout
(0.5 A/DIV)
The output is shorted
Figure 14. The System Enters a Safe Low Duty-ratio Burst Mode when the
Output is Shorted (the LED Driver Output is Shorted for about 10 s for this Test)
Output Diode Short
The LED driver stops operation as soon as 4 faulty DRV
pulses are detected (see Figure 16). In this situation, the
NCL30188B attempts to resume operation when the 4-s
auto-recovery delay is elapsed. The NCL30188A remains
latched off until the system is reset.
Iout (500 mA/div)
VCC
VDS
Rsense voltage
Figure 15. NCL30188B Operation when the Output Diode is Shorted
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18
AND9451/D
Iout (500 mA/div)
VCC
VDS
Rsense Voltage
Figure 16. The Winding or Output Diode Short Circuit Protection (WODSCP) Trips
as soon as 4 Consecutive Faulty DRV Pulses are Detected
REFERENCES
[3] Stéphanie Cannenterre, Understanding sources of
LED current deviations…
[4] Joel Turchi, “NCL30188 and NCL30085 Safety Tests
Consideration”, Application Note
http://www.onsemi.com/pub_link/Collateral/
AND9204−D.PDF
[1] NCL30188 Data Sheet,
http://www.onsemi.com/pub_link/Collateral/
NCL30188−D.PDF
[2] Stéphanie Cannenterre, Application Note
AND9131/D, Designing a LED Driver with the
NCL30080/81/82/83,
http://www.onsemi.com/pub_link/Collateral/
AND9131−D.PDF
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