90 W, Single Stage, Notebook Adaptor

AND8209/D
90 W, Single Stage,
Notebook Adaptor
Prepared by: Terry Allinder
ON Semiconductor
Sr. Applications Engineer
http://onsemi.com
APPLICATION NOTE
Detailed Circuit Description
General Description
The 90 W demo board demonstrates the wide range of
features found in the NCP1651. It provides an 18.5 V, 4.86 A
isolated output, foldback current limit which is ideal for
low−cost battery charger and notebook adaptor applications.
This unit will provide an isolated 18.5 V output from an
input source with a frequency range from 47 Hz to 63 Hz,
and a voltage range of 90 Vrms to 265 Vrms. It is fully
self−contained and includes an internal high voltage startup
circuit, and bias supply that operates off of the Flyback
transformer auxiliary winding.
In addition to excellent power factor, this chip offers fixed
frequency operation in continuous and discontinuous modes
of operation. It has a wide variety of protection features,
including instantaneous current limiting, average current
limiting, and an accurate secondary side power limit.
The detailed operational description and design equations
are contained in the NCP1651 data sheet and in application
note AND8124/D. This application note relates to this
18.5 Vdc 90 W adaptor design.
The 18.5 Vdc 90 W adaptor was designed using the
Excel Design Spreadsheet which can be downloaded from
the ON Semiconductor website (www.onsemi.com). The
design steps for the adaptor are listed below. The schematic
for the 90 W demo board, Figure 7, is located at the end of
the technical write up.
Design Steps
1. Specifications, refer to Table 1 (90−265 V,
18.5 Vout, 90 W).
2. Determine primary inductance.
3. Determine turns ratio.
4. Select the MOSFET.
5. Select the output rectifier.
6. Build transformer with the lowest leakage
inductance.
7. Select the output capacitor for ripple and transient
response.
8. Complete control circuit design.
9. Build and test!
Features
•
•
•
•
•
Fixed Frequency Operation
Operation Over the Universal Input Range
Multiple Protection Schemes
Single Power Stage with Isolated Output
Startup and Bias Circuits Included
Table 1. Demonstration Board Specifications
Requirements
Symbol
Min
Max
Input
Vac
90
265
Frequency
Hz
47
63
Vo (Static Regulation)
Vdc
18.4
18.6
Io
Adc
0
4.86
Output Power
W
−
90
Efficiency
%
84
−
mW
−
500
Standby Power
Vin 230 Vac
 Semiconductor Components Industries, LLC, 2005
May, 2005 − Rev. 0
Figure 1 shows a sample of the System Input parameters
from the NCP1651 Design Excel Spreadsheet. In the
“Limits” column you enter your System Requirements.
Below this is a column labeled “Evaluation”. This is where
you would like to evaluate your design. Normally this is
done at full load and with the minimum input AC line
voltage. To the right you have the spreadsheet plot with the
average input current with respect to phase angle.
1
Publication Order Number:
AND8209/D
AND8209/D
Average Current vs. Phase Angle
Limits
POmax = 90 W
Vinmax = 265 Vrms
Vinmin = 90 Vrms
Vo
= 18.5 V
Lp
= 600 H
fswitch = 100 kHz
Np/Ns = 8.43
1.8
1.6
1.4
CURRENT (A)
Limits info should not
change for a given
design. Evaluation data
may be changed as
desired for various line
and load conditions.
1.2
1.0
0.8
0.6
0.4
0.2
0.0
0
45
Peek switch voltage is
approximately (V). 560.7
90
PHASE ANGLE (°)
135
180
Switching Current vs. Phase Angle
Evaluation
= 90 W
= 90 V
= 0.85
= 60 Hz
= 10 s
= 106 W
4.0
3.5
Peak Current
3.0
CURRENT (A)
Pout
Vi
Effic
fLINE
T
Pin
2.5
2.0
Pedestal Current
1.5
1.0
0.5
0.0
0
45
90
135
180
PHASE ANGLE (°)
Figure 1.
Primary Inductance Selections
For most applications ON Semiconductor recommends
CCM operation at low line and full load to minimize losses
(deciding on the boundary conditioned from CCM to DCM
depending on your magnetics size trade−offs). Using the
Design Spreadsheet we selected a primary inductance of
600 H. Using 600 H the input current is CCM at low line
and full load, and starts to go discontinuous at 230 Vac near
the zero crossing of the line. Selecting this operating mode
allows for a lower Total Harmonic Distortion (THD) and a
high Power Factor (PF), and lower peak current. A lower
primary inductance can be used to reduce the size of the
Flyback transformer, understanding that it will result in a
higher THD, lower PF, and higher peak current which results
in higher losses. Refer to the section labeled “Demo Board
Test Result” for final Demo Board performance.
To determine the required primary inductance you must
first determine if you want to operate in the continuous or
discontinuous mode.
• CCM Operation
− Lower peak and rms currents
− Smaller input filter
− Requires higher primary inductance (more turns)
• DCM Operation
− Higher peak and rms currents
− Larger input filter
− Smaller inductor size
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2
AND8209/D
Transformer Turns Ratio
RDSON Infineon SPA11N80C MOSFETs were used. In
addition to reduce the switching loses the oscillator
frequency of the NCP1651 controller is set to run at 70 kHz.
700
250
600
Max Recommended MOSFET VDS
200
500
150
400
Max Recommended
Diode VR
300
100
200
50
Area of Operation
100
0
Output Diode Selection
For this application we selected an ON Semiconductor
MBR20100CT Schottky diode. The MBR20100CT diode
has a peak inverse voltage rating of 100 V and an average
forward current rating of 10 A.
Leakage Inductance
To minimize the effect of the leakage inductance spike,
the coupling between the primary and secondary of the
transformer needs to be as high as possible. This can be
accomplished, if your transformer requires a primary with
multiple layers, by interleaving the primary secondary
windings. In our 18.5 Vdc application, the transformer
primary has fifty−nine turns, and the secondary has seven
turns. The manufacturer of the transformer, TDK, wound
one layer of the primary with thirty turns and then the seven
turn secondary (two seven turns secondary in parallel), and
the remaining twenty−nine turns of the primary. The results
were a leakage inductance of approximately 8.5 H. Refer
to application note AND8147/D where a comparison of
transformer winding techniques versus leakage inductance
was preformed.
Output Voltage Ripple
The output voltage ripple (V) on the secondary of the
transformer has two components, the traditional high
frequency ripple associated with a flyback converter, and the
low frequency ripple associated with the line frequency
(50 or 60 Hz).
VR, OUTPUT DIODE REVERSE VOLTAGE (V)
VDS, POWER SWITCH DRAIN TO SOURCE
VOLTAGE (V)
There are several tradeoffs that must be considered when
selecting the transformer turns ratio (n). The first is the peak
primary current, the second is the maximum voltage stress
on the Flyback MOSFET (for more details refer to
application note AND8124/D), and the third is the output
diode reverse voltage. Figure 2 graphical shows the
relationships, on the left axis is the MOSFET Drain to
Source voltage (VDS). The horizontal axis is the
transformer turns ratio, and the right vertical axis is the
output diode reverse voltage. For an 18.5 Vdc output the
graph shows that the transformer turns ratio can be between
4.6 and 9.4 (operating from the universal AC input).
The MOSFET in our design has a VDS rating of 800 V,
this is the peak voltage across the device at turn−off
(excluding the leakage inductance spike) is:
VDS = Vinmax 1.414 + (Vo + Vf) n
Where:
Vinmax = 265 Vac
Vo = 18.5 Vdc
To provide some margin for the leakage inductance spike,
the design goal is to keep VDS below 550 V. Based on the
above design goals and the requirement to keep the peak
current as low as possible, our turns ratio was selected to be
8.4. This will limit the MOSFET VDS to approximately
525 V (excluding the turn−off leakage inductance spike),
and the output diode reverse voltage to approximately 55 V.
V Vcap2 Vesr2
The high frequency ripple can be calculated by:
Vcap Ioavg 3
4
5
6
7
8
9 10 11
TRANSFORMER TURNS RATIO
Ipk Iped
2
Vesr Ipk · esr
If we divided the output ripple into 10° increments over
one cycle (180°) the sinusoidal ripple voltage with respect
to phase angle is:
The low frequency ripple can be calculated by:
0
2
Ioavg · dt
Co
12
Figure 2. Turns Ratio Tradeoff
V Transformer Turns Ration Summary
• Higher N leads to lower Iprim, Vsec
• Lower N allows lower Vds, lower leakage inductance,
2 · Co · VoPo· 2 · · fline · sin Where:
Ipk = Peak current (secondary)
Iped = Pedestal of the secondary current
Co = Output capacitance
esr = Output capacitor equivalent series resistance
T = Switching frequency
and capacitor ripple current
MOSFET Selection
For our application one of the primary concerns is the
system efficiency. To reduce the conduction losses two low
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3
AND8209/D
Control Loop
Using the NCP1651 Excel Design Spreadsheet the output
voltage ripple is plotted versus phase angle in Figure 3 and
is approximately 800 mV pk–pk with an output capacitance
of 15,600 F.
The control loop is set up to limit the loop bandwidth at
high line (230 Vac) to approximately 15 Hz with a minimum
phase margin of 45°. The simplest way do this is with the
Excel Design Spreadsheet. In our application the final
components selected were based on cost and standard
components values. The loop crosses 0 dB at 12 Hz at high
line with a phase margin of 50° (refer to Figures 5 and 6).
Step 5, the “Error Amplifier Loop Design”, below, is a
captured screen from the Excel Design Spreadsheet. The
spreadsheet will recommend the compensation values to
provide a stable control loop, but the user typically should
select the closest standard value. For this application, the
18.5 Vdc Demo Board, the spreadsheet recommended
values were not used so we could reduce the loop bandwidth
to provide a higher power factor and lower distortion.
Output Ripple Envelope
0.50
0.40
0.30
RIPPLE (V)
0.20
0.10
0.00
−0.10
−0.20
−0.30
−0.40
−0.50
Step 5 − Error Amplifier Loop Design
0
45
90
135
180
NCP1651 Design Spreadsheet
Provided by ON Semiconductor
DEGREES
Loop Stability
Figure 3. Excel Spreadsheet Output Voltage
Ripple
Suggested
Value
For a comparison, Figure 4 shows the measured output
voltage ripple from the NCP1651 Demo Board. The results
show that the Excel Design Spreadsheets provide very
accurate results.
CTRopto =
VCC =
Optocoupler Current
Transfer Ratio
(Icoll/Idiode)
12 V
Bias Voltage for
Secondary
Operational Amplifier
8,333
4,700 Optocoupler Series
Resistor
Rfb =
455
560 Volt Error Amp
Stability (suggested
value for 10 Hz
crossover)
Cfb =
70.6
22 F
Volt Error Amp
Stability (see bode
plots for Cfb and Rfb)
Ropto =
fz error amp =
12.92 Hz
fp output =
2.68 Hz
Figure 4. Measured Output Voltage Ripple
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4
2.5
AND8209/D
Open Loop Gain
Loop Phase
100
−75
80
−90
High Line
PHASE (°)
GAIN (dB)
60
40
20
−105
−120
−135
0
−150
−20
Low Line
−40
−60
0.01
0.1
1
10
−165
100
−180
0.01
1000
FREQUENCY (Hz)
0.1
1
10
FREQUENCY (Hz)
Figure 5. Loop Gain
Figure 6. Phase Margin
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5
100
1000
7
5
11
10
3
2
D4
1N4006
C16
2.2 F
R5
100 k
R34
100 k
R35
100 k
J2
C27
1.2 F
C26
D3
0.47 F 1N4006
4
0
D5
MUR20100CT
14
C25
6
0.01 F
0
C22
3900
F
15
C23
3900
F
R20
2k
R2
330 k
+Vccsec
R21
8.2 k
U4
SFH6155AA
12
Vref
AC in
C10
C11
R4
C3
R7
C6
0.02
F
1 nF
12 nF
33 k
470
pF
8.6 k 470
pF
1
MC3303
R22
560
6
C8
7
C12
C13
0.1
F
0.1
F
R13
0.2
F
D11
BAS19LT1
R10
0.2
C20
1 F
F
R23
2.67 K
Ilftr
Iavg
7
3
Ct
Ramp
R11
1.0 K
U38
+
−
C24
0.01 F
AC cmp
Gnd
11
5
Is+
2
C17
22 F
R26
3.3 k
C18 0.22 F
C2
1 nF
F
R27
7.5 k
C19
0.01 F
TL431
U2
R29 2 K
R30
100
0
R31
0.006
R6
0.006
12
13
+
−
+Vccsec
U3D
14
MC3303
11
0.01
F
R9
6.6 k
5
6
R1
2.7
AC ref
4
10
C9
R33
17.4 K
Q2
SPP11N80C3
1
Out
Q1
SPP11N80C3
4
D14
BAS19LT1
11
9
FB/SD
Vcc
8
Startup
16
U1
vfb
F
Figure 7. NCP1651 Applications Circuit Schematic
R28
3.3 k
C28
1 F
AND8209/D
6
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R8
680
13
vfb
4
D6
MUR160
R3
330 k
1
2
C31
C1
3900 3900
F
F
C5
2
3
F1
3
470 pF
1
Input
1
C29
0.47 F
D13
AZ23CK18
C21
2200 F
2
J1
1
2
D7
1N4006
2
D2
1N4006
L2
D8
BAS19LT1
Output
D1
1N4006
AND8209/D
DEMO BOARD TEST PROCEDURE
Table 2. Test Equipment
AC Source 85−265 Vac, 47−64 Hz
Variable Electronic Load
Digital Multimeter
Voltec Precision Power Analyzer
1. Connect the AC source to the input terminals J1.
2. Connect a variable electronic load to the output terminals J2, the PWB is marked +, for the positive output, and − for
the return.
3. Set the variable electronic load to 45 W.
4. Turn on the AC source and set it to 115 Vac at 60 Hz.
5. Verify that the NCP1651 provides 18.5 Vdc to the load.
6. Vary the load and input voltage. Verify output voltage as shown in Table 3.
Table 3. Expected Values for Varying Input Voltages and Loads
Vin (Vac)
Vo (Vdc)
@ No Load
Vo (Vdc)
@ 45 W
Vo (Vdc)
@ 90 W
THD (%)
PF
90 W
90
18.7
18.6
18.5
8.0
0.995
115
18.7
18.6
18.5
10
0.990
230
18.7
18.6
18.5
20
0.920
Table 3 shows typical values, the initial set point (18.5 Vdc may vary).
7. To verify total harmonic distortion (THD) first, shut off the AC power supply.
8. Connect the Voltec Precision Power Analyzer as shown in Figure 1.
9. Turn on the AC source to 115 Vac at 60 Hz and set the electronic load to 90 W (only measure the THD at full load).
10. Verify the voltage and current Harmonics of the circuit as shown in Table 3.
11. Shut off the power AC power supply.
12. Set the variable electronic load to 90 W.
13. Turn on the AC source and set it to 230 Vac at 60 Hz.
14. Verify the voltage and current Harmonics of the circuit as shown in Table 3.
Figure 8. NCP1651 Test Setup
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7
AND8209/D
NCP1651 DEMO BOARD TEST RESULTS
PERFORMANCE DATA
Regulation
Vin (Vac)
Pin (W)
Vo (Vdc)
IO (Adc)
PO (W)
Eff (%)
90
106.03
18.55
4.85
89.97
84.85
115
105.21
18.55
4.85
89.85
85.40
230
105.1
18.57
4.85
90.1
85.69
Standby Power
Vin (Vac)
Pin (mW)
115
372
230
455
Power Factor and THD
Vin (Vac)
PF (W)
THD (%)
PO (W)
90
0.995
8.5
90
115
0.990
9.18
90
230
0.940
19.45
90
Vendor Contact List
Vendor
U.S. Phone/Internet
ON Semiconductor
1−800−282−9855
www.onsemi.com/
TDK
1−847−803−6100
www.component.tdk.com/
Vishay
www.vishay.com/
Bussman (Cooper Ind.)
1−888−414−2645
www.cooperet.com/
Coiltronics (Cooper Ind.)
1−888−414−2645
www.cooperet.com/
Fairchild
www.fairchildsemi.com/
Panasonic
www.eddieray.com/panasonic/
Weidmuller
www.weidmuller.com/
Keystone
1−800−221−5510
www.keyelco.com/
HH Smith
1−888−847−6484 www.hhsmith.com/
Aavid Thermalloy
www.aavid.com/
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8
AND8209/D
Table 4. NCP1651 Application Circuit Parts List
Ref Des
Description
Part Number
Manufacturer
C2
Cap, Ceramic, Chip, 0.001 F, 25 V
VJ0603Y102KXXET
VISHAY
C3
Cap, Ceramic, Chip, 470 pF, 25 V
VJ0603Y471JXXET
VISHAY
C4
Cap, Aluminum Elec., 100 F, 35 V
EKB00BA310F00
VISHAY
C5
Cap, Ceramic, Chip, 470 pF, 25 V
VJ0603Y471JXXET
VISHAY
C6
Cap, Ceramic, Chip, 470 pF, 25 V
VJ0603Y471JXXET
VISHAY
C8
Cap, Ceramic, Chip, 0.022 F, 25 V
VJ0603Y223KXXET
VISHAY
C9
Cap, Ceramic, Chip, 0.01 F, 25 V
VJ0603Y103KXXET
VISHAY
C11
Cap, Ceramic, Chip, 0.012 F, 25 V
VJ0603Y123KXXET
VISHAY
C10
Cap, Ceramic, Chip, 0.001 F, 25 V
VJ0603Y102KXXET
VISHAY
C12, C13
Cap, Ceramic, Chip, 0.1 F, 25 V
VJ0606Y104KXXET
VISHAY
C16
2.2 F, Alum Elect, 450 V
(0.394 dia x 0.492H) (0.394 dia x 0.492H)
ECA−2WHG2R2
EKA00DC122P00
Panasonic (Digi–P5873)
Vishay Sprague (20)
C17
Cap, Ceramic, Chip, 22 F, 10 V
C3225X5R0J226MT
TDK
C18
Cap, Ceramic, Chip, 0.22 F, 25 V
VJ0603Y224KXXET
VISHAY
C19
Cap, Ceramic, Chip, 0.01 F, 25 V
VJ0603Y103KXXET
VJ0603Y103KXAAT
C20
Cap, Ceramic, Chip, 1.0 F, 25 V
C3216X7R1E105KT
TDK
C21
220 F, Alum Elect, 25 V
ECA1EM331
Panasonic
C1, C22, C23, C31
Cap, Alum, 3900 F, 25 V
25YXH3900M16X35.5
UHE1E392MHD
Rubycon
Nichicon
C24
Cap, Ceramic, Chip, 0.01 F, 50 V
VJ0603Y103KXAAT
VISHAY
C25
Cap, Ceramic, 0.01 F, 1.0 KV
225261148036
Vishay
C27
Cap, 1.2 F, 275 Vac
F1778−512K2KCT0
Vishay
C26, C29
Cap, Polypropylene, 0.47 F, 400 VDC
MKP1841−447−3000
Vishay−Sprague
C28
Cap, Ceramic, Chip, 1.0 F, 25 V
C3216X7R1E105KT
TDK
D1–D4
Diode, Rectifier, 800 V, 1.0 A
1N4006
ON Semiconductor
D5
Diode, Ultrafast, 200 V, 16 A
MUR20100CT
ON Semiconductor
D6
Diode, Ultrafast, 600 V, 1.0 A
MUR160
ON Semiconductor
D7
Diode, Rectifier, 800 V, 1.0 A
1N4006
ON Semiconductor
D8, D11, D14
Diode, Switching, 120 V, 200 mA, SOT−23
BAS19LT1
ON Semiconductor
D13
Zener Diode, 18 V
AZ23C18
VISHAY
F1
Fuse, 2.0 A, 250 Vac
1025TD2A
Bussman
L2
5.0 A Sat, 3.0 mH Inductor, Common Mode
Q4007−A
Coilcraft
Q1, Q2
FET, 11 A, 800 V, 0.45 ?, N−channel
SPA11N80C3
Infineon
R1
Resistor, SMT1206, 2.7
CRCW1206270JRE4
Vishey
R2
Resistor, Axial Lead, 270 k, 1/4 W
CMF−55−270K00FKRE
Vishey
R3
Resistor, Axial Lead, 270 k, 1/4 W
CMF−55−270K00FKRE
Vishey
R5
Resistor, 100 k, 3.0 W, 5%
CFP−3104JT−00K
VISHAY
R4
Resistor, SMT1206, 33 k
CRCW120633KOJNTA
Vishey
R7
Resistor, SMT1206, 8.66 k
CRCW12068661F
Vishey
R8
Resistor, SMT1206, 680
CRCW12066800F
Vishey
R10
Resistor, SMT, 0.2, 1.0 W
WSL2512 .20 1%
Vishey Dale
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AND8209/D
Table 4. NCP1651 Application Circuit Parts List (continued)
Ref Des
Description
Part Number
Manufacturer
R9
Resistor, Axial Lead, 5.4 k, 1/4 W
CMF−55−5K400FKBF
Vishey
R11
Resistor, SMT1206, 1.0 k
CRC12061K00JNTA
Vishey
R12
Resistor, 100 k, 3.0 W, 5%
CFP−3104JT−00K
VISHAY
R13
Resistor, SMT, 0.2, 1.0 W
WSL2512 .20 1%
Vishey Dale
R14
Resistor, SMT1206, 100
CRC12062K100JNTA
Vishey
R20
Resistor, SMT1206, 2.0 k
CRC12062K00JNTA
Vishey
R21
Resistor, SMT1206, 8.2 k
CRC12068K20JNTA
Vishey
R22
Resistor, SMT1206, 2.0 k
CRC12062K00JNTA
Vishey
R23
Resistor, SMT1206, 2.67 K, 1%
CRCW12062670F
Vishey
R26
Resistor, SMT1206, 3.3 k
CRC12063K30JNTA
Vishey
R27
Resistor, SMT1206, 7.5 k
CRC12067K50JNTA
Vishey
R28
Resistor, SMT1206, 3.3 k
CRC12063K30JNTA
Vishey
R29
Resistor, SMT1206, 2.0 k
CRC12062K00JNTA
Vishey
R30
Resistor, SMT1206, 100, 1%
CRCW12061000F
Vishey
R31
1.0 W, 0.006 Resistor
WSL251R006FTB
Vishey
R6
1.0 W, 0.006 Resistor
WSL251R006FTB
Vishey
R33
Resistor, SMT1206, 17.4 k, 1%
CRCW120617400F
Vishey
R34
Resistor, 100 k, 3.0 W, 5%
CFP−3104JT−00K
VISHAY
R35
Resistor, 100 k, 3.0 W, 5%
CFP−3104JT−00K
VISHAY
T1
Transformer, Flyback (Lp 600 H)
SRW42EC−U10H014
TDK
U1
PFC Controller
NCP1651
ON Semiconductor
U2
2.5 V Programmable Ref, SOIC
TL431ACD
ON Semiconductor
U3
Quad Op A
MC3303D
ON Semiconductor
U4
Optocoupler, 1:1 CTR, 4 Pin
SFH615AA−X007
Vishay
Part Number
Manufacturer
Hardware
Ref Des
Description
H1
Printed Circuit Board
H2
Connector
171602
Weidmuller
(Digi 281−1435−ND)
H3
Connector
171602
Weidmuller
(Digi 281−1435−ND)
4672
Keystone
4.1″ X 1.0″ X 0.05″
Manufactured
Insulator
H8, H9
Aluminum Heatsinks
ON Semiconductor and
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to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
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