A 5 V/2 A Standby Power Supply for INTEL Compliant ATX Applications

AND8241/D
A 5.0 V/2.0 A Standby
Power Supply for INTEL
Compliant ATX Applications
Prepared by: Christophe Basso
ON Semiconductor
http://onsemi.com
APPLICATION NOTE
ATX power supply units (PSU) require a standby section
who keeps alive some particular areas of the motherboard.
Among the live sections are the USB ports, the Ethernet
controller and so on. The INTEL ATX Power Supply Design
Guide 2.01 (rev. June 04) describes the standby voltage rail
needed for this purpose. This is the +5.0 VSB section (3.3.3):
• Output voltage: 5.0 V, 5%
• Nominal load current: 2.0 A
• 500 ms pulses current (USB wake--up event): 2.5 A
• Input power less than 1.0 W at 230 Vac for an output
power of 500 mW
• Power on time: 2.0 s maximum
• Short circuit protection with auto--recovery
subharmonic mode. A simple resistor to ground injects
the right compensation level.
• Over Power Protection: a resistive network to the bulk
reduces the peak current capability and accordingly
harnesses the maximum power at high line. As this is
done independently from the auxiliary VCC, the design
gains in simplicity and execution speed.
• Latch--off input: some PC manufacturers require a
complete latch--off in presence of an external event, e.g.
overtemperature. The controller offers this possibility
via a dedicated input.
• Frequency dithering: the switching frequency (here
65 kHz) is modulated during operation. This naturally
spreads the harmonic content and reduces the peak
value when analyzing the signature.
On top of these requirements, most of PSU designers use
the standby power supply as an auxiliary rail to power the
main controller. This is an auxiliary 12--13 V rail or, if an
UC384X is implemented, this rail can go up to 20 V. When
put in standby, a small switch shuts down the controller by
interrupting this rail.
To help designers quickly fulfilling the above needs, the
NCP1027 has been introduced. This DIP 8 package hosts a
high performance controller together with a low RDS(on)
700 V BVdss MOSFET. On top of the standby needs, we
have packed other interesting goodies in this circuit. They
are summarized below:
• Brown--out detection: the controller will not allow
operation in low mains conditions. You can adjust the
level at which the circuit starts or stops operation.
• Ramp compensation: designing in Continuous
Conduction Mode helps to reduce conduction losses.
However, at low input voltage (85 Vac), the duty--cycle
might exceed 50%, and the risk exists to enter a
© Semiconductor Components Industries, LLC, 2005
December, 2005 -- Rev. 0
Design Description
A full CCM operation gave us an adequate performance
in this particular case, with good full load efficiency results
as we will see. The part switches at 65 kHz which represents
a good trade--off between switching losses and EMI control.
A brown--out circuit was implemented, turning the SMPS on
around 80 Vac and turning it off at 60 Vac. Different values
can easily be selected by altering the dedicated resistive
network. Please note that this network impedance has a
direct influence on the standby power. To limit the amount
of current the supply can deliver at high line, it is necessary
to limit the propagation delay effects. The NCP1027 hosts
an exclusive circuitry used to reduce the maximum peak
current as the line increases. We will see that, once
implemented around the auxiliary diode, it does not affect
the standby power and nicely harnesses the maximum
power. The electrical schematic of the board appears on
Figure 1.
1
Publication Order Number:
AND8241/D
AND8241/D
TR1:
Lp = 3.4 mH
Np:Ns = 1:0.06
Np:Naux = 1:0.15
Bulk
R10
2.8 Meg
C7
100 mF
16 V
R11
2.2 Meg
R7
1k
C5
10 nF
R6
150 k
0.5 W
13 V
+
L1
2.2 mH
D1
MBRD835L
+
C1
1500 mF
R5
47
+ C2
1500 mF
R3
100
U3
D4
1N4937
+ C10
47 mF
400 V
1
R14
560 kΩ
U1
NCP1027
2
+
C13
1 mF
8
7
R4
1k
C6
10 nF
R9
27 k
+
+
U2
TL431
R2
10 k
5
4
C9
100 pF
R8
78 k
R1
10 k
C4
0.1 mF
3
C8
10 mF
10 V
0
TR1
D4
1N4937
Aux.
C3
220
mF
5 V--2 A
R13
47 kΩ
U3b
C11
2.2 nF
Type = Y1
Figure 1. The application board electrical schematic without the EMI filter for simpler representation.
Let us start the review by the transformer description.
where:
Lp, the primary inductance, 3.4 mH
N, the turn ratio, 0.06
Fsw, the switching frequency, 65 kHz
Vin, the input voltage
Vout, the output voltage
Calculations lead to the following mode transition load
values:
Transformer
The design of the transformer section represents the most
difficult part as standby power in no--load and 0.5 W output
must be respected. Various iterations have lead us to adopt
the following characteristics:
Lp = 3.4 mH
Np:Ns_power = 1:0.06
Np:Ns_aux = 1:0.152
The auxiliary winding in this case delivers 12.5 V but it
can be set to any other value, depending on the main
controller VCC. The turn ratio limits the reflected value on
the drain to less than 100 V, without risks of biasing the
MOSFET body diode at the lowest input voltage. This
transformer is available from Coilcraft under a reference
detailed in the Bill Of Material (BOM).
As we have seen, our design operates in CCM at full load
but obviously enters the Discontinuous Conduction Mode
(DCM) at light loads. The transition point can be evaluated
using the following formula:


NVin + Vout 2
Rc = 2LpN2Fsw
NVin
Lowest line, Vin = 120 Vdc: Rc = 4.56 Ω or Iout = 1.0 A
Highest line, Vin = 370 Vdc: Rc = 2.4 Ω or Iout = 2.0 A
Please note that low line is actually 85 Vac but the ATX
PSU including a large bulk capacitor, we can neglect the
ripple given the low output power of this converter, hence a
120 V value. At the highest line, the design operates at the
boundary between CCM and DCM. The duty--cycle at full
load and lowest input line can be evaluated via the flyback
static transfer function:
D=
Vout
NVin + Vout
(eq. 2)
The highest duty--cycle is found to be 41%. However, this
number is likely to increase a little, given the presence of the
leakage inductance.
(eq. 1)
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AND8241/D
Ramp Compensation
If the limit is set to 750 mA, then we have:
Being in CCM with a duty--cycle close to 50% (or above
50% in transient conditions), we need ramp compensation.
Several ways exist to evaluate the amount of ramp
compensation. The quickest one evaluates the off slope of
the secondary inductance and injects around 50% of this
slope through a compensating ramp (Sa ) into the controller.
With the NCP1027, the ramp compensation level is set via
a simple resistor connected from pin 2 to the ground, as
shown in Figure 1.
We can calculate the off slope, the one actually needed to
evaluate Sa , by reflecting the output voltage over the primary
inductance. The slope is projected over a complete
switching period.
Ifinal = 750 m + 100 100 n = 753 mA
3.4 m
(eq. 8)
Ifinal = 750 m + 374 100 n = 761 mA
3.4 m
(eq. 9)
The difference looks small but it often leads to a
significant different in output current capabilities, especially
with larger propagation delays. Hence, the need to act in
high line conditions via a dedicated circuitry.
The NCP1027 hosts a special section used to reduce the
maximum peak current limit as the mains increases. The
system works by connecting a resistive network to the bulk
capacitor, as Figure 2 depicts:
V
+ Vf
6 × 15 m
Soff = out
Tsw =
= 441 mA∕15 ms
0.06 × 3.4 m
NLp
Bulk
(eq. 3)
ROPPH
The NCP1027 features a current--mode architecture using
a SENSEFETt device. That is to say, the controller does not
directly sense the current via a resistor but through a Kelvin
cell. For this particular circuit, the cell ratio can be modeled
as an equivalent sense resistor of 350 mΩ. This current slope
will thus become a voltage slope having a value of:
S′off = 0.441 × 0.375 = 165 mV∕15 ms
+
Cbulk
2
(eq. 4)
8
4
lb1
7
lb3
3
If we chose 50% of this down slope, then the final
compensation ramp will present a slope of:
Sa =
1
U1
NCP1027
5
lb2
S′off
= 83 mV∕15 ms = 5.53 kV∕s (eq. 5)
2
Following the data sheet indications, we evaluate the
resistor value to be:
Rramp = 7562 = 91 k
0.083
ROPPL
(eq. 6)
In case no ramp compensation is required, pin 2 must be
tied to VCC , the adjacent pin. As experiments were carried
at lower input voltages (Cbulk is small on the demo board),
it was decided to slightly increase the amount of ramp
compensation by reducing Rramp to 78 kΩ.
Figure 2. A possible option to reduce the
peak excursion consists of connecting a
resistive divider to the bulk capacitor.
By injecting a current Ib3 proportional to Vbulk , the
maximum peak current limits reduces. Analytically
obtaining the right value for Ib3 might represent a
complicated exercise as many parameters play a role. To the
opposite, a simple experimental method can be setup to
obtain the value of the needed current:
1. Configure your working power supply as
suggested by Figure 3 or 4. These figures are
purposely simplified for ease of understanding of
the added circuitry. Figure 3 represents the safest
way to run the measurement as the dc power
supply injects current via an optoisolator without
sharing the converter’s ground. An amp--meter is
inserted to read the current Ib3. Leave the dc bias
to zero for now.
2. Power the converter and set the input voltage to
the highest of your specification. Let us assume it
is 375 Vdc.
Over Power Protection (OPP)
Power supply controllers sensing the primary current to
check whether it goes over a certain limit often face
propagation delay problems. That is to say, despite the
current limit detection by a dedicated comparator, the
information takes a certain amount of time to propagate
through the logic circuits and eventually reset the latch.
During this time, the primary current keeps increasing by a
rate given by the primary inductance and the input voltage.
Hence, we can quickly see the effects of a 100 ns delay at
high line or low line:
V
Ifinal = Ipeak_max + in tproper
Lp
(eq. 7)
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AND8241/D
5. At a certain time, if you continue to increase the
current in pin 7, the converter enters in protection
mode. The current flowing in pin 7 and the voltage
on it prior to the shutdown, correspond to the
variables you look for. In this example, we measured
a Vf voltage of 2.45 V and a current of 31 mA.
3. Increase the output current to the point where the
system should shut down per specification. Let’s
say 3.2 A for this design.
4. Start to increase the variable power supply dc
voltage until the amp--meter deviates. Increase
carefully because you deal with hundred of mA only.
R4
10 k
1
U1
NCP1027
2
R2
10 k
+
8
7
S
3
+
Aux
5
4
+
Variable
Power Supply
+
IB3
R3
10 k
Figure 3. An isolated way to safely inject current into pin 7 at high line.
1
2
U1
NCP1027
8
R2
100 k
7
S
3
Aux
+
+
4
5
+
IB3
R3
10 k
+
Variable
Power Supply
Figure 4. A non--isolated way to inject current into pin 7 at high line.
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AND8241/D
The preliminary curve showing the relationship between
the injected current and the maximum peak current setpoint
appears in Figure 5. We can see that 31 mA corresponds to
a 20% reduction in the maximum peak current capability.
The Vf parameter specifies the voltage at which pin 7 starts
to pump in current. It is around 2.5 V as we can see it.
(IPEAK--IPEAKOPP)/IPEAK (%)
90.0%
80.0%
70.0%
60.0%
50.0%
40.0%
30.0%
20.0%
10.0%
0.0%
0
50
100
150
200
LOPP CURRENT (mA)
Figure 5. Maximum Peak Current Setpoint Reduction
vs. Pin 7 Injected Current
To compute the resistor values, we need to define the
range within which the OPP reduction should activate. As
we do not want any OPP action at low input voltages, we will
select resistors to start reducing at Vin = 200 Vdc, with a
clamp at Vin = 375 Vdc. Using the following equations and
our collected data leads us to the final values:
VbulkH = 375 Vdc
VbulkL = 200 Vdc
= 31 mA
IOPP
Vf
= 2.45 V
VbulkH --VbulkL
V = 70 kΩ
IOPP(VbulkL --Vf) f
(eq. 10)
V
--V
ROPPH = ROPPL bulkL f = 5.6 MΩ
Vf
(eq. 11)
ROPPL =
Unfortunately, despite the good behavior of the network,
its permanent presence on the bulk rail will affect the
consumption in standby. When you chase every hidden
milli--watt, it can become a nasty problem. To avoid this
trouble, a simple solution around the auxiliary winding can
be worked out. This is presented in Figure 6. During the on
time, where the power switch is turned on, the input voltage
appears across the primary transformer. Given the auxiliary
diode configuration, N . Vin + Vout also appears on the
cathode, N being the primary to secondary turn ratio. As this
voltage moves up and down with the bulk level, we can
perfectly use it for our OPP purposes. Due to its pulsating
low voltage nature, power consumption will be the smallest.
VCC
C7
100 mF
16 V
R7
1k
+
Aux.
D4
1N4937
1
ROPPH
560 kΩ
U1
NCP1027
2
8
7
3
C8
10 mF
10 V
+
4
C9
100 pF
R8
78 k
5
ROPPL
47 k
U3b
Figure 6. The auxiliary diode lends itself very well to
a cheap and energy efficient OPP implementation.
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AND8241/D
Hooking an oscilloscope probe on D4 cathode gives us the
bulk image evolution between its minimum and maximum
values. We can then run the calculation again to obtain
ROPPH and ROPPL :
VbulkH = 55 Vdc
VbulkL = 37 Vdc
= 31 mA
IOPP
Vf
= 2.45 V
V
--V
Rupper = Rlower bulk1 BO
VBO
Rlower = VBO
Rupper
Rlower
2
PBO = 330
= 27 mW.
4.02 Meg
On the board, we purposely modified these values to
further improve the standby power.
6
OUTPUT CURRENT (A)
Auxiliary Winding
Without OPP
The auxiliary winding is set to deliver 13 V nominal when
the converter is fully loaded. To avoid any VCC drops during
transient loading, e.g. sudden load removal, the main VCC
capacitor C7 has been increased to 100 mF, leading to a
stable VCC in no--load conditions. The resistor R7 sets
the Overvoltage Protection (OVP) level by adjusting
the injected current in pin 1 internal shunt in case of
problems. With 1.0 kΩ, the OVP level is set to
VOVP = 1.0 k · 7.0 m + 8.7 = 15.7 V typical on the
auxiliary winding. Different values can be obtained by
changing the resistor values or the ratio between the power
and auxiliary windings.
4
With OPP
2
1
0
240
260
280
300
320
= 4.0 MΩ
= 22 kΩ
Total power dissipation at nominal line is then
Figure 7 shows the results before and after OPP
implementation. The output current stays below 3.5 A in all
cases which is well within specs.
3
(eq. 13)
Suppose we want to start at Vin = 110 Vdc (77 Vac) and
stop operation at Vin = 70 Vdc (50 Vac), then applying
Equations 12 and 13 leads to the following values:
ROPPH = 580 kΩ → 560 kΩ after tweak
ROPPL = 41 kΩ → 47 kΩ after tweak
5
Vbulk1 --Vbulk2
IBO × (Vbulk1 --VBO)
(eq. 12)
340
360
INPUT VOLTAGE (Vdc)
Clamping Section
Figure 7. Over Power Protection at work keeps the
output current below 3.5 A.
The converter uses an RCD clamp to safely limit the
voltage excursion on the MOSFET drain. Failure to keep
Vds (t) below 700 V will permanently damage the circuit. In
application where the input line can be subject to strong
high--voltage parasitic pulses, we recommend the usage of
a Transient Voltage Suppressor (TVS), that will hard clamp
all potentially lethal spikes on the drain. Figure 8 portrays
the way to wire the TVS. Finally, the TVS offers superior
performance in standby power compared to the RCD clamp.
Simply because the RCD clamp always activates since the
capacitor discharges via the resistor R as soon the leakage
inductance is reset. Unfortunately, even in standby, this
mechanism generates light losses. If you want to further save
50 mW, a TVS is of good usage. A 200 V TVS has shown
to be a good solution here. You can use the 1.5KE200A from
ON Semiconductor for instance.
Brown--out
Brown--out (BO) detection offers a means to protect the
converter in presence of low input voltages by stopping
switching operation until the mains comes back to a normal
value. The circuit works by observing a fraction of the bulk
level via a resistive divider routed to pin 2. When the level
on pin 2 lies below 0.6 V, the controller does not allow
switching operation, but the high--voltage current source
maintains the VCC on pin 1. Then, as soon as pin 2 voltage
crosses 0.6 V, the VCC is ready to power the chip and
switching can start. As this occurs, pin 2 injects around
12 mA (IBO ) in the resistive bridge to create a hysteresis. The
designer must thus select the turn--on and turn--off voltages
to further apply the following equations:
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AND8241/D
Secondary Side
Vbulk
D2
1.5KE220A
The secondary side uses a Schottky diode featuring an
8.0 A capability and a breakdown voltage of 35 V. This is
enough since high line conditions imply a Peak Inverse
Voltage (PIV) of:
3
PIV = NVin + Vout = 0.06 × 370 + 5 = 27.2 V
(eq. 14)
5
D1
1N4937
1
The forward voltage Vf goes down to 0.41 V @ Tj = 125°
which, neglecting ohmic losses, induces conduction losses
of:
4
Pdiode = IRMS2Rd + IavgVf ≈ 2 × 0.41 = 0.8 W
(eq. 15)
5
The output capacitor Cout is selected to a) pass the
adequate RMS current b) limit the undershoot ΔV when the
output is banged by a current step ΔI. The undershoot depth
of a closed--loop converter having a bandwidth fc can be
evaluated via the following formula:
Figure 8. A TVS must be used if the input voltage
can be subject to high voltage spikes.
ΔV =
ΔI
fcCout
Calculations gave us a value of 2.4 mF, made of two
low--impedance 10 V/1200 mF capacitors.
A TL431 ensures a stable regulation at 5.0 V via a type--2
amplifier. R3 and the NCP1027 internal pullup resistor set
the midband gain, whereas C4 sets the zero position. The
small capacitor C9 filters out residual noise and adds a high
frequency pole, acting together with the optocoupler one.
The step load response is clean, without any ringing at both
high and low line conditions. The circuit can be slightly
modified to add some more soft--start, in case designers
would fear output overshoots. Figure 10 shows how to
connect a 1.0 mF/10 V capacitor to soften the start--up
sequence:
Figure 9. Always check the voltage on the drain at
the highest line condition (265 Vac).
D1
MBRD835L
L1
2.2 mH
+5 V
C1
1500 mF
+
+
+
C2
1500 mF
C3
220 mF
0
R3
100
R4
1k
U3
+
(eq. 16)
C4
0.1 mF
X3
TL431
Css
1 mF
R1
10 k
R2
10 k
Figure 10. A 1.0 mF capacitor connected on the TL431
strengthens the startup sequence.
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AND8241/D
The basic description being done, let us have a look at the board performance. Standby measurements were captured with
a Yokogawa WT210 on a board after it warmed up for 15 minutes. Please make sure the WT210 volt--mater is placed before
the current shunt, otherwise its input impedance will degrade the low--power measurements by 30 mW at high line.
Static Measurements
Vin = 230 Vac
Tambient = 25°C
Pout = 0
Pout = 0.5 W
Pout = 10 W
Pin = 88 mW
Pin = 779 mW
Pin = 12.3 W
Vaux = 10.14 V
Vaux = 11.64 V
Vaux = 12.14 V
η = 64.2%
η = 81.3%
→ Please note that replacing R6 by a 200 V TVS as mentioned in the text, reduces the input power at 0.5 W @ 230 Vac down
to 715 mW and 73 mW at no load.
Pout = 0
Pout = 0
Pin = 98 mW at Vin = 265 Vac with RCD
Pin = 85 mW at Vin = 265 Vac with TVS
Tcase NCP1027 = 50°C at Pout = 10 W
Short--circuit temperature: NCP1027 Tcase = 42°C, Tdiode = 52°C
Maximum output current: 3.0 A @ 265 Vac
Vin = 85 Vac
Tambient = 25°C
Pout = 0
Pout = 0.5 W
Pout = 10 W
Pin = 54 mW
Pin = 704 mW
Pin = 12.6 W
Vaux = 10.15 V
Vaux = 11.6 V
Vaux = 12.6 V
η = 71%
η = 79.5%
Tcase NCP1027 = 60°C at Pout = 10 W
Short--circuit temperature: Tcase NCP1027 = 44°C, Tdiode = 52°C
Maximum output current: 3.1 A
Dynamic Measurements
Some critical waveforms have been captured on the demonstration board and are reproduced below:
Figure 11. Short--Circuit Protection, Drain--Source Waveform
In Figure 11, we can see the power supply operating in a
so--called hiccup mode, trying to re--start as soon as the
internal timer has elapsed. The resulting duty--burst stays
below 8%, keeping all component temperatures at a moderate
level. This is an auto--recovery type of protection, implying
a re--start when the fault is removed (Vin = 230 Vac).
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AND8241/D
10 W
0W
0.5 W
Figure 12. Startup Sequence at Three Loading Conditions
Figure 12 shows a typical startup sequence, captured at
different output levels (0, 0.5 W and 10 W) for Vin = 85 Vac.
Changing the input voltage does not modify the shape of the
waveform. This waveform has been captured with the
1.0 mF wired as suggested by Figure 10.
Figure 13. Load step where Vout is banged from 0.1 A to 2.5 A with a
1.0 A/ms slew--rate (Vin = 85 Vac).
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AND8241/D
Figure 14. Load step where Vout is banged from 0.1 A to 2.5 A with a
1.0 A/ms slew--rate (Vin = 230 Vac).
Figures 13 and 14 represent the step load response from
the standby mode to the wake--up mode, featuring a
variation from 100 mA up to 2.5 A. The spike you can see
on the waveforms comes from the current discontinuity
indured by the LC filter inductor L1 .
Figure 15. Peak--to--peak ripple, Pout = 0.5 W, Cable length = 30 cm, Vin = 85 Vac.
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AND8241/D
Figure 16. Peak--to--peak ripple, Pout = 0.5 W, Cable length = 30 cm, Vin = 230 Vac.
Figures 15 and 16 display the ripple amplitude when the
board enters skip cycle, mainly at a 0.5 W output power. The
measurement has been carried at two line levels and using
30 cm long cables to mimic a real PSU cabling connector.
High Line
Low Line
Figure 17. EMI signature, low line and high line in quasi--peak, Pout = 10 W.
Figure 17 shows the advantage of EMI jittering, offering
a clean signature at both line levels. As the sweep is
successfully made in Quasi--Peak (QP) it implies immediate
compliance in average mode.
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AND8241/D
High Line
Low Line
Figure 18. Peak Output Current in Short--Circuit at Both Line Conditions: 6.3 A
54 ms
676 ms
Figure 19. Output Current Waveform in Short--Circuit
Some PC application specs require the output average and
RMS currents in short--circuit to stay within given limits.
With this design, we have obtained the following results at
high--line:
S Ipeak = 6.4 A
S Iout,av = 6.4 x 54 / 676 = 511 mA
S Iout,rms = 6.4
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54 = 1.8 A
676
AND8241/D
PCB Views
The PCB routing includes large copper areas around the
integrated circuit to maximize its power dissipation. The
diode is placed on the copper side and due to its low Vf, it
leads to good thermal performance.
Figure 20. The PCB Copper Area of the Demonstration Board
Figure 21. The SMD Positions on the Copper Side
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AND8241/D
Figure 22. Component Side View
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AND8241/D
Bill of Material
Manufacturer
Part Number
Substitution
Allowed
Lead
Free
General
Semiconductor
DF08M
Yes
No
20%
Panasonic
EEUFC1C122
Yes
47 mF/400 V
20%
Panasonic
ECA2GHG470
Yes
Capacitor, Y1 Class
2.2 nF/400 V
20%
Vishay
WKP222
No
1
Capacitor, X2 Class
220 nF/275 V
20%
Evox Rifa
PHE840MX6220M
No
C13
1
Radial Lead Electrolityc Capacitor
1 mF/50 V
20%
Panasonic
EEUFC1H1R0
Yes
C3
1
Radial Lead Electrolityc Capacitor
220 mF/16 V
20%
Panasonic
EEUFC1C221
Yes
C4
1
SMD Capacitor
100 nF
10%
Epcos
B37872--K5104--K60
Yes
C5
1
Polyester Chip Capacitor
10 nF/630 V
10%
Vishay
MKT1822--310
Yes
C6
1
SMD Capacitor
10 nF
10%
Epcos
B37872--K5103--K60
Yes
C7
1
Radial Lead Electrolityc Capacitor
10 mF/63 V
20%
Elna
RE3--63V100M
Yes
C8
1
Radial Lead Electrolityc Capacitor
100 mF/10 V
20%
Panasonic
EEUFC1A101
Yes
C9
1
SMD Capacitor
1 nF
10%
1206
Epcos
B37872--K5102--K60
Yes
DPAK
Designator
Quantity
Description
Value
Tolerance
B1
1
Single--Phase Bridge Rectifier
800 V
NA
C1,C2
2
Radial Lead Electrolityc Capacitor
1200 mF/16 V
C10
1
Radial Lead Electrolityc Capacitor
C11
1
C12
Footprint
1206
1206
Manufacturer
D1
1
Schottky Barrier Rectifier
35 V/8 A
NA
ON Semiconductor
MBRD835L
Yes
Yes
D2,D4
2
Fast--Recovery Rectifier
600 V/1 A
NA
ON Semiconductor
1N4937
Yes
Yes
J1
1
PCB Connector
NA
NA
Multicomp
JR--201S
Yes
J2
1
PCB Connector
NA
NA
Weidmuller
PM5.08
Yes
L1
1
Inductor, 2.2 mH, 2.5 A
2.2 mH
NA
Wurth Elektonic
744772022
Yes
L2
1
Common Mode Inductor, 2*27 mH
27 mH
NA
Schaffner
RN114--0,8/02
Yes
R1
1
Axial Lead Resistor 1/4w
10 kΩ
5%
Neohm
CFR25J10K0
Yes
R10
1
Axial Lead Resistor 1/4w
2.8 MΩ
5%
Neohm
CFR25J2M8
Yes
R11
1
Axial Lead Resistor 1/4w
2.2 MΩ
5%
Neohm
CFR25J2M2
Yes
R13
1
Axial Lead Resistor 1/4w
47 kΩ
5%
Neohm
CFR25J47K0
Yes
R14
1
Axial Lead Resistor 1/4w
560 kΩ
5%
Neohm
CFR25J560K
Yes
R2
1
SMD Resistor
10 kΩ
1%
1206
Vishay
CRCW1206101J
Yes
R3
1
SMD Resistor
100 Ω
1%
1206
Vishay
CRCW1206100RJ
Yes
R4
1
Axial Lead Resistor 1/4w
680 Ω
5%
Neohm
CFR25J680
Yes
R5
1
Axial Lead Resistor 1/4w
47
5%
Neohm
CFR25J47
Yes
R6
1
Axial Lead Resistor 1w
150 kΩ/1 W
5%
Neohm
CFR100J150K0
Yes
R7
1
Axial Lead Resistor 1/4w
1 kΩ
5%
Neohm
CFR25J1K0
Yes
R8
1
SMD Resistor
78 kΩ
1%
1206
Vishay
CRCW1206781J
Yes
R9
1
SMD Resistor
27 kΩ
1%
1206
Vishay
CRCW1206271J
Yes
U1
1
NCP1027
NA
NA
DIP8
ON Semiconductor
NCP1027
No
Yes
U2
1
Adjustable Shunt Regulator
2.5--36 V/
1--100 mA
2.20%
TO92
ON Semiconductor
TL431
No
Yes
U3
1
Optocoupler
NA
NA
DIP4
Vishay
SFH615A
No
T1
1
Transformer
NA
NA
Coilcraft
DA2077--AL
No
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AND8241/D
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