A 800 W Bridgeless PFC Stage

AND8392/D
A 800 W Bridgeless PFC
Stage
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APPLICATION NOTE
Bridgeless PFC is one of the options to meet these new
requirements. The main goal of this paper is to present a
bridgeless solution that is relatively easy to implement in the
sense that it does not require any specific controller and the
operation remains very similar to that of a conventional
PFC. The solution is illustrated by a 800 W, wide mains
application driven by the NCP1653.
Environmental concerns lead to new efficiency
requirements when designing modern power supplies. For
instance, the 80 plus initiative and moreover its bronze,
silver and gold derivatives force desktops and servers
manufacturers to work on innovative solutions. An
important focus is on the PFC stage that with the EMI filter
can be easily consume 5% to 8% of the output power at low
line, full load.
Why remove the bridge?
D1
D2
+
AC Line
EMI
Filter
PFC
Stage
+
D3
D4
Figure 1. The Input Current Flows Through Two Diodes
Figure 1 portrays the diodes bridge that is usually inserted
between the EMI filter and the PFC stage. This bridge
rectifies the line voltage to feed the PFC stage with a
rectified sinusoid input voltage. It is well known that as a
result of this structure, the input current must flow through
two diodes before being processed by the PFC boost:
♦ For one line half−wave, D1 and D4 conduct (red
arrows of Figure 1)
♦ For the other one, D2 and D3 convey the current
(blue arrows of Figure 1)
As a matter of fact, two diodes of the bridge are
permanently inserted in the current path. Unfortunately,
these components exhibit a forward voltage that leads to
conduction losses.
The mean value of the current seen by the bridge is the line
average current. Hence we can write the following equation:
© Semiconductor Components Industries, LLC, 2015
February, 2015 − Rev. 1
ǂIbridgeǃ
T
+ ǂI line(t)ǃ
line
T
+
line
2 Ǹ2
@ I line(rms)
p
(eq. 1)
The line rms current can be easily expressed as a function
of the power and of the line voltage:
I line(rms) +
P out
h @ V in(rms)
(eq. 2)
Where:
• Pout is the output power
• h is the efficiency
• Vin(rms) is the rms line voltage
1
Publication Order Number:
AND8392/D
AND8392/D
Since two diodes permanently see the average line
current, the bridge consumes a power that can be computed
as follows:
In other words, an input bridge consumes about 2% of the
input power at low line of a wide mains application. Hence,
if one of the series diodes could be suppressed, 1% of the
input power could be saved and the efficiency could for
instance, rise from 94% to 95%. Also, the major hot spot
produced by a traditional diodes bridge would be eliminated
at the benefit of an improved reliability of the application.
Here are the motivations behind the bridgeless approach.
2 Ǹ2 @ P out (eq. 3)
P bridge + 2 @ V f @ I bridge
^ 2 @ Vf @
T
h @ p @ V in(rms)
line
ǂ
ǃ
Finally, if we assume a 1 V forward voltage per diode and
computing the losses at the usual low line rms voltage
(90 V), it comes:
P bridge ^ 2 @ 1 @
2 Ǹ2 @ P out
h @ p @ 90
^ 2% @
P out
(eq. 4)
h
”Basic” Bridgeless Architecture
Switching cell when PH2 is high
M1 is Off
D1
PH1
D2
L
AC Line
PH2
+
M1
M2
Switching cell when PH1 is high
M2 is Off
Figure 2. ”Traditional” Bridgeless Solution
The line (Note 1) and the PFC inductor are placed in series
and the arrangement they form is connected to the switching
nodes of the two switching cells.
Figures 3 and 4 show the functioning of the bridgeless
PFC when the ”PH1” is high. Figure 5 summarizes the
operation for the other half−line cycle.
Figure 2 portrays a classical option for bridgeless PFC.
They are two switching cells. Each of them consists of a
power MOSFET and of a diode:
♦ The first cell (M1, D1) processes the power for the
half−line cycle when the terminal ”PH1” of the line
is high and is in idle mode for the rest of the line
period.
♦ The second cell (M2, D2) is active for the other
half−wave when ”PH1” is low compared to terminal
”PH2”.
1. The EMI filter is not represented for the sake of simplicity.
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AND8392/D
D1
PH1
Ac Line
M2
Body
Diode
PH2
+
M1
Figure 3. MOSFET Conduction Phase (”PH1” Half−Wave)
D1
PH1
Ac Line
M2
Body
Diode
PH2
+
M1
Figure 4. Current Path When the MOSFET is Open (”PH1” Half−Wave)
M2 is On: Conduction Time
M1 is Open: Off Time
D2
PH1
Ac Line
PH2
M2
M1
Body
Diode
Figure 5. Operation for the Line Half−Wave when ”PH2” is High
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AND8392/D
”dV/dt” leads to an increased common−mode noise that is
difficult to filter. This is probably the major drawback of the
solution ([1] and [2]).
An improvement consists in splitting the inductor into two
smaller ones and to insert one of them between one line
terminal and the switching node of cell 1 and the other one
between the second line terminal and the switching node of
cell 2. Doing so, the large aforementioned ”dV/dt” is no
more directly applied to the input terminals and thus, the line
potentials can be more stable with respect to the board
ground. However, such a solution still exhibits a worse
signature in the high frequency part of the spectrum.
As sketched by Figures 3 to 5, the input current is
processed by the switching cell that is active for the
considered line half−wave. The MOSFET of the inactive
cell has a role anyway, since its body diode serves as the
current return path.
Compared to a conventional PFC stage, the losses due to
the bridge are saved but the body diode of the inactive
MOSFET conveys the coil current. Finally, this structure
eliminates the voltage drop of one diode in the line−current
path for an improved efficiency.
However, the presented architecture presents several
inconveniences that actually, result from the fact that the line
is not referenced to the ground as it is the case in a
conventional PFC. Instead, in this structure, the line is
floating compared to the PFC stage ground, leading to the
following difficulties:
♦ Certain PFC controllers need to sense the input
voltage. In this structure, a simple circuitry cannot
do the job.
♦ Similarly, the coil current cannot be easily
monitored.
Besides these difficulties in the circuit implementation,
EMI filtering is the main issue. When ”PH1” is high, the
negative terminal ”PH2” is attached to ground by the M2
body diode. Hence, the application ground is connected to
ac line as it happens in a conventional PFC. Now, when
”PH2” is high, the MOSFET M2 switches and the voltage
between the line terminals and the application ground pulses
as well. More specifically, the potential of the ”PH2” node
nearly oscillates between 0 (when the MOSFET on) and the
PFC output voltage (when the MOSFET is off). This large
Figure 6 portrays another option for bridgeless PFC. This
solution was proposed by Professors Alexandre Ferrari de
Souza and Ivo Barbi ([3]). As shown by Figure 6, there is no
full bridge. Instead, the ground of the PFC circuit is linked
to the line by diodes D1 and D2 and each terminal feeds a
PFC stage. Hence, the solution could be viewed as 2−phase
PFC where the two branches operate in parallel:
♦ For the half−wave when the terminal ”PH1” of the
line is high, diode D1 is off and D2 connects the PFC
ground to the negative line terminal (”PH2”). D2
grounds the input of the ”PH2 PFC stage” branch
that thus, is inactive and the ”PH1 PFC stage”
processes the power.
♦ For the second half−line cycle (when ”PH2” is
high), the ”PH2 PFC stage” branch is operating and
”PH1 PFC stage” that has no input voltage, is
inactive.
“PH1”
PFC
Stage
PH1
+
Ac Line
2−Phase Approach
“PH2”
PFC
Stage
PH2
DRV
D2
+
D1
Figure 6. 2−Phase Architecture
Figure 7 gives an equivalent schematic for the two
half−waves.
Similarly to the traditional bridgeless structure presented
in the previous paragraph, this architecture eliminates one
diode in the current path and hence improves the efficiency.
Another interesting characteristic of this structure is that
the PFC stage that is active, behaves as a conventional PFC
boost would do:
♦ When the ”PH1” terminal is positive (see
Figure 7a), diode D1 opens and D2 offers the return
♦
path. The input voltage for the ”PH1” PFC stage is a
rectified sinusoid referenced to ground.
For the other half−wave (see Figure 7b), when
”PH2” is the positive terminal, D1 offers the return
path. Diode D2 is off and sees a rectified sinusoid
that inputs the ”PH2” PFC stage. Again, we have a
conventional PFC where the input voltage and the
boost are traditionally referenced to ground.
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AND8392/D
PH1 PFC Stage
+
PH1
−
PH1
PH2 PFC Stage
Ac Line
AC Line
PH2
+
DRV
−
PH2
+
D1
D2
+
DRV
b) terminal PH2 is the high one
a) Terminal PH1 is the High One
Figure 7. Equivalent Schematic for the Two Half−Waves
It is also worth noting that the 2−phase structure does not
require any specific controller. The MOSFETs of the two
branches are referenced to ground and they can be
permanently driven even when their phase in idle phase. The
MOSFET of the inactive branch would then be turned on and
off useless but:
♦ At the benefit of a simplified circuitry since there is
no need for detecting the active phase and for
PH1
Ac Line
♦
directing the drive signal to the right MOSFET
according to the half−line cycle.
At the price of the additional losses due to the
inactive MOSFET drive. The loss is not very high
anyway since the voltage across the MOSFET is null
when its input voltage is zero. Hence, the gate
charge to be provided is approximately halved
compared to that of the active MOSFET.
0V
PH2
DRV
D2
D1
+
Rsense
Figure 8. Operation for the ”PH2” Half−Wave
Actually, a large portion of the current flows as indicated
in black.
This is because the body diode of the supposedly inactive
MOSFET provides the current with another path. The coil
exhibiting a low impedance at the line frequency, we have
two diodes in parallel and the current share between them.
One should however note that the current does not
necessary return by the D1 and D2 diodes. Figure 8 portrays
the ”expected” current path when ”PH2” is high (the same
analysis could have been done for ”PH1” high):
♦ The blue path is supposed to be the current path
when the MOSFET is on
♦ The red one, that of the current when the MOSFET
is open.
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AND8392/D
5 ms/div
Phase 1 current (5 A/div)
0A
Part of the active
phase current flows
through the inactive
MOSFET and coil!
0A
Phase 2 current (5 A/div)
Figure 9. Part of the Current Flows through the Supposedly Inactive MOSFET and Coil
Figure 9 portrays the input current for each branch. One
can see on this plot that a negative current takes place
through the body diode during the ”inactive” half−wave.
The main inconvenience of this behavior is that the input
current cannot be sensed by inserting a RSENSE resistor in
the supposed return path (as shown by Figure 8) since part
of current takes another road.
That is why current sense transformers can be of great help
to measure the current in such a structure.
Implementation of the Bridgeless PFC
Figure 10 highlights the main parts of our 800 W
prototype.
Branch 1
PH1
Current sense
transformer
AC Line
+
Bulk
Capacitor
EMI
Filter
Branch 2
PH2
Return
Path
Diodes
D2
D1
FB
VCC
2
3
4
RETURN
D3
1N5406
8
NCP1653
1
Current sense
transformer
7
6
5
In−rush
current
detection
(optional)
RSense
RETURN
Input voltage
sensing
CS
One control circuitry
Figure 10. Simplified Application Schematic
Input Voltage Sensing
Branch 1 and Branch 2 PFC Boost:
The NCP1653 monitors the input voltage for
feed−forward purpose. The NCP1654 further features a
brown−out protection. As shown by Figure 10, two small
diodes (e.g. 1N4007) are used that re−construct the rectified
line voltage that can then be monitored by the circuit.
Two PFC boost converters are to be designed. This
application note does not focus on the dimensioning of the
power components since it is relatively traditional.
However, the fact that each branch is active for one half−line
cycle only, improves the heating distribution. Also, the rms
current being halved in each branch, the power components
does not need to be as large as those of a conventional PFC.
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AND8392/D
Inrush Detection
Control Circuitry
Instead of a third current sense transformer, we made the
choice to keep a current sense resistor to monitor the current
that re−fuels the buck capacitor, i.e.:
♦ In−rush currents during the start−up phase or in
over−load situations
♦ The current provided by the boost diodes in normal
operation
Due to that, the MOSFET and the input bridge are
referenced to the ”RETURN” potential instead of ground.
The voltage between the RETURN and ground potentials is
the negative voltage engendered by the RSENSE resistor. If
this voltage becomes too large (during in−rush sequences for
instance), the MOSFETs’ source potential may dramatically
drop and some accidental MOSFET turn on may follow.
That is why the voltage across the RSENSE resistor is limited
by a diode.
This diode must be able to sustain the in−rush current and
its forward voltage must high enough so that the RSENSE
voltage is not clamped until the current largely exceeds its
permissible level in normal operation. Otherwise, the
clamping diode would prevent the RSENSE voltage from
becoming high enough to trigger the over−current
protection.
As already mentioned, the 2−phase bridgeless PFC does
not require any complex control circuitry. The NCP1653
PFC controller directly drives the two branches. The
NCP1653 is a compact 8−pin PFC controller that operates in
continuous conduction mode. As it directly adjusts the
conduction time as a function of the coil current, there is no
inner current loop to be compensated for an eased design.
Housed in a DIP8 or SO8 package and available in two
frequency versions (67 kHz or 100 kHz), the NCP1653
integrates all the features necessary for a compact and
rugged PFC stage including a current sensing technique that
allows the use of very low impedance, current sense resistors
for reduced losses and a significant improvement of the
efficiency. Compared to traditional solutions, the efficiency
increase can be as high as almost 1%.
The NCP1654 is a NCP1653 derivative that further
incorporates a brown−out detection block to disable the PFC
stage when the line magnitude is too low. Also, the voltage
regulation is made more accurate and a dynamic response
enhancer dramatically minimizes the large deviation of the
output voltage that a sharp line/load step could otherwise
produce (see [4] and [5]).
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8
C13
1 μF
Type = X2
CM1
CM2
Figure 11. Application Schematic
F1
10 A
30
N
36
R32
680k
R31
680k
R30
680k
C14
1 μF
Type = X2
40
Earth
37
42
90 to 265 Vrms
50 or 60 Hz line voltage
L
39
43
49
C18
4.7 nF
Type = Y2
C19
4.7 nF
Type = Y2
C15
1μF
Type = X2
C1
100nF / 63V
10
CS
5
R8
2.2k
R3
470k
R2
2700k
27
R1
2200k
D3
1N4007
PH1
RETURN
D1
PH2
PH2
R4
2.2k
RETURN
3
D4
1N4007
PH2
D10
1N5817
C2
100nF
C3
220nF
D2
R5
390
R6
100k
28
C5
10nF
1
3
4
D5
1N5406
12
13
15 2
24
NCP1653
Vm 5
Gnd 6
In
CS
Drv 7
Vcc 8
Vctrl
FB
U1
NCP1653
R14
390k
1
6
R15
680k
R7
56k
Vcc
R13
10
R16
680k
C4
1nF
R12
100
C25
22μF
23
R17
180k
C6
22μF
C24
220nF
51
C27
100pF
53
4
3
2
1
C28
220nF
Vcc2
NC 8
OutB 5
Vcc 7
R11
100m / 3W
R10
100m / 3W
R9
100m / 3W
MC33152
InB
Gnd
InA OutA 7
NC
RETURN
DRV2
DRV1
Vcc2
PH2
R23
10k
R22
10
L2
0.2m
R28
15k
Q3
SPP20N60
2
33
RETURN
20
R26
15k
Q1
SPP20N60
7
4
RETURN
9
C17
1μF
R19
10k
R18
10
RETURN
DRV1
C16
1μF
L1
0.2m
D8
1N4148
D9
1N4148
C11
330u / 450 V
DRV2
32
DRV1
31
CS
21
R25
10k
R24
10
R29
3
CS
R21
10k
R20
10
R27
3
Q2
SPP20N60
22
C12
330u / 450 V
Q4
SPP20N60
D7
CSD10060
26
D6
CSD10060
AND8392/D
AND8392/D
NCP1653
And
MC33152
MOSFET
driver
Bulk
converter to
generate
the Vcc
voltage
(NCP1012)
Figure 12. Photograph of the Evaluation Board
♦
Main components of the board:
♦
♦
♦
♦
Input diodes: one GSIB1580 from Vishay (15−A,
800−V). This diodes bridge has actually been
implemented to provide return−path diodes D1 and
D2 of Figure 10. Using a diodes bridge helps have
the two input diodes cooled down by the board
heatsink.
Current sense transformers (1 per branch):
WCM601−2 from West Coast Magnetics (20 A,
50 turns)
Boost diodes (1 per branch): CSD10060 (10 A,
600 V SiC diode from CREE)
Power MOSFETs (2 per branch): SPP20N60 from
Infineon (20 A, 600−V, 0.19 W)
♦
Inductors (1 per branch): 200 mH / 9.7 Arms / 16 Apk
/ 5 App coil (ferrite core)
Part Name: PFC−Choke LDU80025
Part Number: 203860
Producer: www.J−Lasslop.de, Information:
[email protected]
Controller (only 1 for the application): NCP1653:
100 kHz, 8−pin, Continuous Conduction Mode PFC
that can be easily replaced by its NCP1654
derivative that further features a brown−out
protection and a dynamic response enhancer (see [4]
and [5]).
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Performance of the 800−W board
− Typical Waveforms
90 Vrms
Iline (10 A/div)
230 Vrms
Iline (10 A/div)
Vout
Vout
CS (negative sensing)
CS (negative sensing)
Vin,1 (input voltage for branch 1)
Vin,1
Figure 13. Typical Waveforms at Low Line (Left) and High Line (Right)
”Vsense” (identical to ”CS”) is a negative representation of
the MOSFET current. The current sense transformers are
wired so that only the current drawn by the MOSFET drain
is monitored (possible current flowing in the opposite
direction cannot be sensed).
The waveforms are similar to those of a traditional CCM
PFC.
Plots Figure 13 of portray typical waveforms at full load
(Iout = 2.1 A). ”CS” is the negative voltage provided by the
current sense transformers. It is representative of the current
flowing into the MOSFETs of the two branches (”CS” is the
common output of the two current sense transformers). As
expected, the input voltage of the ”PH1 PFC stage” (”Vin,1”)
is a rectified sinusoid for one half−line cycle and null for the
other one. The line current is properly shaped.
Figure 14 provides a magnified view at the top of the line
sinusoid. The switching frequency is 100 kHz. The signal
230 Vrms
rms
90 Vrms
rms
Iline (10 A/div)
Iline (10 A/div)
Vout
Vout
Vin,1
Vsense (negative sensing)
Vsense (negative sensing)
Vin,1 (input voltage for branch 1)
Figure 14. Magnified Views of Figure 13 Plots
Thermal Measurements
Measurement Conditions:
The following results were obtained using a thermal
camera, after a 1/2 h operation. The board was operating at
a 25°C ambient temperature, without a fan. These data are
indicative.
For the bridge used to provide return−path diodes D1 and
D2 of Figure 10, the MOSFETs and the boost diodes, the
measures were actually made on the heat−sink as near as
possible of the components of interest.
Vin(rms) = 88 V
Pin(avg)) = 814 W
Vout = 381 V
I out = 2 A
PF = 0.995
THD = 9 %
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Bridge MOSFET1 Diode1
Devices
Temperature (5C)
85
95
Coil1
MOSFET2
Diode2
Coil2
Bulk Capacitor
CM EMI coil
47
86
80
48
40
45
77
Efficiency and Total Harmonic Distortion
100
20
230 Vrms
98
90 Vrms
96
THD (%)
EFFICIENCY (%)
15
120 Vrms
90 Vrms
94
90
100
200
20% Pmax
300
400
500
600
700
OUTPUT POWER (W)
0
100
800
Pmax
300
400
500
600
OUTPUT POWER (W)
700
800
900
Pmax
Figure 16 portrays the THD at 90, 120 and 230 Vrms over
the load. One can note that the total harmonic distortion
remains very low even in high line, light load (< 15%) where
the line current is small and more sensitive to all the sources
of distortion like the system inaccuracies and mainly the
EMI filter.
Conclusion
A bridgeless PFC based on the 2−phase architecture has
several merits among which one can list the ease of control
or the absence of high frequency noise injected to the line
(eased EMI).
The paper presents the performance of a prototype
controlled by the NCP1653 (100 kHz version). The
NCP1654 that further incorporates the brown−out
protection and a dynamic response enhancer could be
implemented as well.
The prototype has been tested at full load (800 W output)
without a fan (open frame, ambient temperature). In these
conditions, the full−load efficiency was measured in the
range of 94% at 90 Vrms and as high as 95% at 100 Vrms.
The THD remains very low.
A NCP1653 or NCP1654 driven 2−phase bridgeless PFC
is a solution of choice for very efficient, high power
applications. More information on this bridgeless
architecture can be found in [6].
P in(avg) + V in(rms) @ I in(rms) @ PF
Open Frame, Ambient Temperature, No Fan
To the light of Figure 15, we can note that:
♦ Like in a conventional PFC, the efficiency is higher
at high line. In fact, this is because as in any boost,
♦
200
20% Pmax
Figure 15 portrays the efficiency over the line range, from
20% to 100% of the load.
The efficiency was measured in the following conditions:
♦ The measurements were made after the board was
30 mn operated full load, low line
♦ All the measurements were made consecutively
without interruptions
♦ PF, THD, Iin(rms) were measured by a power meter
PM1200
♦ Vin(rms) was measured directly at the input of the
board by a HP 34401A multi−meter
♦ Vout was measured by a HP 34401A multi−meter
♦ The input power was computed according to:
the losses are reduced when the ratio
120 Vrms
Figure 16. Total Harmonic Distortion Over the Load
Range
Figure 15. Efficiency Performance
♦
230 Vrms
5
92
♦
10
ǒ Ǔ is
V in(t)
V out
high (close to 1).
At low line (90 Vrms), full load, the efficiency is in
the range of 94% without a fan. It was even
measured that @ 100 Vrms, we reach 95% at full
load.
We can note that the efficiency is high at light load.
For instance, at 20% of full load, efficiency is in the
range or higher than 96%.
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References
4. NCP1653 data sheet and application notes,
www.onsemi.com
5. NCP1654 data sheet and application notes,
www.onsemi.com
6. Joel Turchi, “A High−Efficiency, 300−W
Bridgeless PFC Stage”, Application Note
AND8481/D,
http://www.onsemi.com/pub_link/Collateral/AND
8481−D.PDF
1. Laszlo Huber, Yungtaek Jang and Milan M.
Jovanovic, ”Performance Evaluation of Bridgeless
PFC Boost Rectifiers”, APEC 2007
2. Pengju Kong, Shuo Wang, and Fred C. Lee,
”Common Mode EMI Noise Suppression for
Bridgeless PFC Converters”
3. Alexandre Ferrari de Souza and Ivo Barbi, ”High
Power Factor Rectifier with Reduced Conduction
and Commutation Losses”, Intelec, 1999
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