Analysis and Design of QuasiSquare Wave Resonant Converters Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 2 Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 3 What is Quasi-Square Wave Resonance ? MOSFET turns on when VDS(t) reaches its minimum value. ¾ Minimize switching losses ¾ Improves the EMI signature valley MOSFET turns on in first valley www.onsemi.com 4 MOSFET turns on in second valley Quasi-Resonant Operation In DCM, VDS must drop from (VIN + Vreflect) to VIN Because of Lp-Clump network Æ oscillations appear Oscillation half period: Vin VDS Lp Cout 1:N SW www.onsemi.com 5 Rload Vout VIN + Vreflect VIN Vin VDS t x = π LpClump Clump Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 6 A Need to Limit the Switching Frequency In a self-oscillating QR, Fsw increases as the load decreases Higher losses at light load if Fsw is not limited 2 methods to limit Fsw: – Frequency clamp with frequency foldback – Changing valley with valley lockout www.onsemi.com 7 Frequency Foldback in QR Converters QR mode Second valley First valley In light load, frequency increases and hits clamp ¾ Multiple valley jumps ¾ Jumps occur at audible range ¾ Creates signal instability www.onsemi.com 8 Changing Valley As the load decreases, the controller changes valley (1st to 4th valley in NCP1380) The controller stays locked in a valley until the output power changes significantly. ¾ No valley jumping noise ¾ Natural switching frequency limitation 80000 SWITCHING FREQUENCY (Hz) 70000 60000 50000 4th 3rd 2nd 1st 40000 30000 20000 QR operation 10000 0 0 10 VCO mode www.onsemi.com 9 20 30 OUTPUT POWER (W) 40 50 60 Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 10 Calculating the QR Inductor Calculation steps: 1. Primary to secondary turns ratio 2. Primary and secondary peak current 3. Inductance value 4. Primary and secondary rms current www.onsemi.com 11 Turns Ratio Calculation Derate maximum MOSFET BVdss: Vds , max = BVdss k D kD: derating factor For a maximum bulk voltage, select the clamping voltage: 15% derating Vos Vreflect Vos: diode overshoot Deduce turns ratio: N s kc (Vout + V f ) N ps = = Np Vclamp kc: clamping coef. kc = Vclamp / Vreflect ) www.onsemi.com Vds,max Vclamp Vclamp = Vds , max − Vin ,max − Vos 12 BVdss Vbulk,max How to Choose kc Choose kc to equilibrate MOS conduction losses and clamping resistor losses. 800 V MOSFET 600 V MOSFET 3 3 PRclamp Ptot ≈ 2.8 W 2 PMOS,on @ Vin,min Ploss (W) Ploss (W) 2 Ptot ≈ 3.8 W 1 PRclamp 1 PSW,on @ Vin,max PSW,on @ Vin,max 0 1.2 1.5 PRclamp = kleak 1.8 Pout kc 2.1 kc η kc − 1 PMOS,on @ Vin,min 2.4 2.7 0 3 PMOS ,on 1.2 4 Pout 2 = Rdson 2 3η Vin ,min 1.5 1.8 kc 2.1 2.4 ⎛ 1 kc + ⎜⎜ ⎝ Vin ,min BVdss k D − Vin ,max − Vos 2 Psw,on www.onsemi.com 13 2.7 BV k − V −V ⎞ 1⎛ = ⎜ Vin ,max + dss D in ,max os ⎟ COSS Fsw,max 2⎝ kc ⎠ 3 ⎞ ⎟⎟ ⎠ Primary Peak Current and Inductance 1 Lpri I pri , peak Fswη 2 Pout = DCM Ipri,peak ton ton 0 toff tv toff tv Tsw = I pri , peak L pri I pri , peak Vin ,min Vout + V f N ps Pout ⎛ 1 =2 + ⎜⎜ η ⎝ Vin,min Vout + V f www.onsemi.com 14 + I pri , peak L pri N ps + π L pri Clump ⎞ 2 Pout Clump Fsw ⎟⎟ + π η ⎠ Coss contribution alone. Lpri = 2 Pout I pri , peak 2 Fswη RMS Current Calculate maximum duty-cycle at maximum Pout and minimum Vin: d max = I pri , peak Lpri Vin ,min Fsw,min Deduce primary and secondary RMS current value: I pri ,rms = I pri , peak I sec ,rms = I pri , peak N ps d max 3 1 − d max 3 Ipri,rms and Isec,rms www.onsemi.com 15 Losses calculation Design Example Power supply specification: – Vout = 19 V – Pout = 60 W – Fsw,min = 45 kHz – 600 V MOSFET – Vin = 85 ~ 265 Vrms Vbulk Vout T1 . . Gnd www.onsemi.com 16 Design Example Based on equations from slides 11 to 14: ¾ Turns ratio: ¾ Peak current: N ps = kc (Vout + V f ) BVdss k D − Vin ,max − Vos I pri , peak = = ¾ Inductance: Lpri = ¾ Max. duty-cycle: d max = ¾ Secondary rms current: www.onsemi.com 1.5 × (19 + 0.8) ⇒ N ps ≈ 0.25 600 × 0.85 − 375 − 20 N ps 2 Pout ⎛ 1 + ⎜⎜ η ⎝ Vin,min Vout + V f ⎞ 2 Pout Clump Fsw ⎟⎟ + π η ⎠ 2 × 60 ⎛ 1 0.25 ⎞ 2 × 60 × 250 p × 45k + ⎜ ⎟ +π 0.85 ⎝ 100 19.8 ⎠ 0.85 2 Pout 2 × 60 = I pri , peak 2 Fswη 3.322 × 45k × 0.85 ¾ Primary rms current: 17 = I pri , peak Lpri Vin ,min Fsw,min = I pri ,rms = I pri , peak I sec ,rms = ⇒ Lpri = 285 µH 3.32 × 285µ 45k ⇒ d max = 0.43 100 d max 0.43 = 3.32 3 3 I pri , peak N ps ⇒ I pri , peak = 3.32 A ⇒ I pri ,rms = 1.26 A 1 − d max 3.32 1 − 0.43 = 3 0.25 3 ⇒ I sec ,rms = 5.8 A Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 18 MOSFET TO220 package: RθJA = 62 °C / W Ambient temperature: TA = 50 °C, MOS junction temperature: TJ = 110 °C Power dissipated by TO-220 without heatsink: MOS RDS(on) @ TJ = 110 °C: Assume we do not want a heatsink www.onsemi.com 19 RDSon120 = PTO − 220 TJ − TA = ≈ 1W RθJA PTO − 220 1 = = 0.6 Ω 2 2 I pri , RMS 1.3 ! 15 A, 600 V MOSFET MOS Heatsink We choose a 7 A, 600 V MOS: RDS(on)120 = 1.2 Ω, RDS(on)25 = 0.6 Ω MOS conduction losses: Tj Pcond = RDS ( on )120 I pri ,rms 2 = 1.2 × 1.262 = 1.9 W Thermal resistance of the heatsink: Rθ SA www.onsemi.com 20 TJ − TA 110 − 50 = − Rθ JC − Rθ CS = − 2.5 − 1.6 = 27 °C / W Pcond 1.9 Output Diode TO-220 package Æ power dissipation: 1 W I Rd MBR20200: VT0 = 0.60 V, Rd = 20 mΩ V VT0 Diode conduction losses: Pdiode = VT 0 I out + Rd I sec ,rms 2 Pdiode = 0.60 × 3.2 + 0.02 × 5.82 = 2.60 W Heatsink: Rθ SA = TJ − TA 110 − 50 − Rθ JC − Rθ CS = − 2.0 − 1.6 Pcond 2.6 Rθ SA ≈ 19 °C / W www.onsemi.com 21 Output Capacitor Selection Maximum output voltage ripple: Vripple = 2%Vout =0.38 V Maximum ESR of output capacitor: RCout ≤ Vripple I sec , peak 0.38 = ≈ 30 mΩ 13.2 RMS current circulating in Cout: I Cout , RMS = I sec ,rms 2 − I out 2 = 5.82 − 3.22 ≈ 4.83A Two 1200-µF capacitors (3.2 Arms, 13 mΩ / capacitor) Losses in Cout: PCout = RCout I Cout , RMS 2 = 6.5m × 4.832 = 0.15W www.onsemi.com 22 Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 23 Origin of Losses . OUT . GND Conduction losses in ESR of capacitor, diodes, clamp resistor, sense resistor Conduction and switching losses in MOSFET Copper and core losses in inductor www.onsemi.com 24 Switching Losses at Turn-On Traditional approach: Psw,on ⎛ Vout + V f 1 = COSS ⎜ Vin ,min − ⎜ 2 N ps ⎝ 2 ⎞ ⎟⎟ Fsw ⎠ 1 19 + 0.8 ⎞ ⎛ = 200 p ⎜100 − ⎟ 45k = 2 mW 2 0.25 ⎠ ⎝ 2 C OSS (V DS ) = Use the variable capacitor for losses calculation: Psw,on Vout + V f 2⎛ = ⎜ Vin ,min − 3 ⎜⎝ N ps 3 2 ⎞ ⎟⎟ CDO VO Fsw ⎠ C DO V 1 + DS VO CDO COSS 2⎛ 19 + 0.8 ⎞ = ⎜100 − ⎟ 3⎝ 0.25 ⎠ 3/ 2 200 p 25 45k = 3.6 mW Losses are negligible! www.onsemi.com 25 VO Bulk Capacitor Losses Power losses caused by ac current in the bulk capacitor ESR (350 mΩ) Pbulk = Rbulk I bulk ,rms 2 I bulk ,rms = I in ,mean I in,rms I in,ac 2 3Fline tc = I bulk,rms −1 Cbulk I in,mean Flyback Conduction time of diode bridge tc = 1 4 Fline ⎛ Vmin arcsin ⎜ ⎜ V peak ⎝ − 2π Fline I bulk ,rms = 0.70 ⎞ ⎟⎟ ⎠ = 3 ms 2 − 1 = 1.3 A 3 × 50 × 3m Pbulk = 350m × 1.32 = 0.59 W www.onsemi.com 26 Vpeak= 120 V Vmin= 70 V tc Diode Bridge Losses I KBU4K From datasheet curves: VT0 = 0.70 V , Rd = 70 mΩ Rd There are two diodes conducting at the same time. V Two diodes always conduct during half a cycle: Pdiodes VT0 I in ,mean ⎛ ⎞ = 2 ⎜ VT 0 + Rd I d ,rms 2 ⎟ = 2 × 0.7 × 0.35 + 70m ×1.042 = 640 mW 2 ⎝ ⎠ I in ,mean 0.70 I d ,rms = = = 1.04 A 3 Flinetc 3 × 50 × 3m ( ) As two diodes always conduct, over a cycle, the bridge power is: PKBU 4K = 2 Pdiodes = 1.28W www.onsemi.com 27 RCD Clamp Losses Power losses in clamping resistor: PRclamp = Vclamp 2 Vds,max Rclamp Vos Vclamp Rclamp can be calculated with: Rclamp Rclamp ⎛ Vout + V f 2Vclamp ⎜ Vclamp − ⎜ N ps ⎝ = Fsw Lleak I peak 2 ⎞ ⎟⎟ ⎠ 19 + 0.8 ⎞ ⎛ 2 ×120 ⎜120 − ⎟ 0.25 ⎠ ⎝ = = 7 k Ω ⇒ Rclamp = 7.3 k Ω 2 45k × 2.8µ × 3.32 PRclamp www.onsemi.com 28 Vbulk,max 1202 = ≈ 2W 7.3k Inductor Losses P Rpri = R pri ,dc I in ,mean + R pri ,ac I pri ,ac 2 PRsec = Rsec , dc I out 2 + Rsec ,ac I sec ,ac 2 2 Lleak2 Lleak1 R pri Rac Rdc . Lm Rc R sec . Core losses: Determined from data provided by the manufacturer www.onsemi.com 29 Losses Summary for the 19 V / 60 W Adapter 2.6 W 2W . 1.28 W OUT 0.15 W . 2.32 W GND 0.59 W 1.9 W 0.33 W Total losses: Ploss = 11.14 W Estimated efficiency: www.onsemi.com 30 η= Pout 60 = ≈ 84.4 % Pout + Ploss 60 + 11.14 Comparison with Real Adapter C1 10n R4 12k T1: R11 18k Nps = 1 : 0.25 Npaux = 1 : 0.18 Lp=290 µH Rx 10 - + R18 10k . D5 1N4937 X18 KBU4K IN R6 340k D6 1N4148 C5b 1.2mF C6 22p C9 220nF X2 1 8 2 7 3 6 4 5 Gnd 25V C15 2.2nF Type = Y1 T1 Gnd R5 27k R9 1k C13 100u D4 1N967 C7 100uF 35V NCP1380 18 mH 2A L1 C5a 1.2mF 35V R1 1.3MEG C18 100nF D2 MBR20200 . D1 1N4937 C14 100u . Vout L3 2.2u Vdrain D7 1N4148 M1 SPP07N60 R16 10 R29 1k R15 1k R7 39k C10 47n D3 1N4148 R2 1k C5 1n C8 220p C3 100n X6 NTC C11 4.7u Q1 BC857 R3 47k R26 0.47 R27 0.47 C20 100n Gnd Efficiency measured after the EMI filter at 85 Vrms (120 Vdc) Measured Pout = 60.1 W Pin = 70.9 W η = 84.8% Calculated Pout = 60 W Pin = 71.14 W η = 84.4% www.onsemi.com 31 X5 TL431_G R8 10k Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 32 Synchronous Rectification High rms currents in secondary side Æ increased losses in the output diode. Replace the diode with a MOSFET featuring a very low RDS(on). + - Increased efficiency Degraded standby power Vout . Cout . Q sync www.onsemi.com 33 Gnd Synchronous Rectification Basics During (t2-t1), current flows into the body diode Minimize (t2-t1) duration to reduce body diode conduction. Vout . Cout . Q sync Gnd Body diode conducts before the MOSFET is turned-on. No switching losses www.onsemi.com 34 Rload Losses in the Sync. Rect. Switch PQsync = PON + PQdiode Body diode conduction losses Body diode and MOS conduction losses for the 19 V/65 W adapter PQdiode = V f I out Fswtdelay 1 PON = RDS ( on )120 I sec ,rms 2 Ploss (W) Low if tdelay small MOSFET conduction losses PON 0.8 0.6 0.4 0.2 PQdiode 0 200 Vin (V) 300 Losses in the Sync. Rect. switch are mainly conduction losses. www.onsemi.com 35 Choosing the Sync. Rect. MOSFET Target around 1 W conduction losses in Sync. Rect. switch to avoid using an heatsink. RDSon120 = Vout = 19 V Fsw,min = 45 kHz Universal mains 1W I sec , RMS 2 RDSon120 = 70 mΩ Ploss (W) 6 MBR20200 RDSon120 = 50 mΩ 4 RDSon120 = 30 mΩ 2 0 1 2 3 4 Iout (A) www.onsemi.com 36 5 6 60 W QR Sync. Rect. Calculations Body diode losses: PQdiode = V f I out Fswtdelay = 0.7 × 3.2 × 45000 × 70n PQdiode = 7 mW MOSFET losses: PON = RDS ( on )120 I sec ,rms 2 = 30m × 5.82 PON = 1W Total Sync. Rect switch losses: PQsync = 1 + 0.007 ≈ 1W Losses into the MBR20200 diode: 2.6 W Power loss saving: 1.6 W www.onsemi.com 37 Using NCP4302 NCP4302 CS input connected to the drain of the MOSFET Trigger input for CCM. Connect it to Gnd if not used. 1 Sync Vcc 8 7 DRV 3 2 Trig 4 5 Dlyadj Adjust: - minimum on-time of the Sync. MOSFET - the minimum off-time of the Sync. MOSFET to be immune to drain ringing of the primary switch. www.onsemi.com 38 Gnd 6 TL431 Cathode TL431 VREF input Measured Efficiency with Sync. Rect. Vout L7 2.2u T1 . C5ax 1.2mF . C5bx 1.2mF 35V M2 C19 100uF 35V Gnd 25V IRFS4320 D²PAK Gnd R31 10 R24 1k R10 75 D8 1N4148 X8 DIP4302 R25 1k 1 Sync 7 DRV 3 2 Trig 4 5 Dlyadj 6 R33 15k Vcc R17 27k C3 47n 8 R30 110k R19 39k R20 10k Gnd Efficiency measured after the EMI filter at 85 Vrms Measured Pout = 60.1 W Pin = 69.25 W η = 86.8% Calculated Pout = 60 W Pin = 69.54 W η = 86.3% www.onsemi.com 39 Measured efficiency with Diode and Sync. Rect. Efficiency vs Output Power ( 1 W to 0.5 W) Efficiency vs Output Power (60 W to 6 W) 90 68 87 86 Efficiency (%) Efficiency (%) Sync.Rect. 230 V Diode 230 V Sync.Rect 110 V Diode 110 V 70 89 88 85 84 83 Sync.Rect. 230 V Diode 230 V Sync.Rect 110 V Diode 110 V 82 81 15 25 35 45 55 64 62 60 58 56 80 5 66 65 0 0.2 230 Vrms Standby power 85 Vrms www.onsemi.com 0.6 0.8 1 Pout (W) Pout (W) 40 0.4 Diode Pin = 110 mW Sync. Rect. Pin = 140 mW Diode Pin = 90 mW Sync. Rect. Pin = 122 mW 1.2 Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 41 Power Stage Borderline Conduction Mode Approximation. Neglect the high frequency Right Half Plane Zero (RHPZ) Open loop transfer function of power stage ⎛ ⎞ ηVIN Rload RESR Cout s + 1 v$ out ( s ) = H (s) = ⎜⎜ ⎟⎟ $v FB ( s ) 2α Rsense ( 2Vout + N psVIN ) ⎝ ( Req + RESR )Cout s + 1 ⎠ N ps Vout R ESR Lp Vin R load C out QR Controller www.onsemi.com 42 2Vout + N psVIN Gnd α: internal dividing ratio R sense VFB Req = Rload Vout + N psVIN between FB and CS from datasheet (typically 3 or 4) The Optocoupler Pole Parasitic capacitance of optocoupler Î opto pole Vdd sopto = 1 1 + sR pullup Copto Vout R pullup R LED a c C opto Optocoupler characterization reveals a pole at 5 kHz k e If fopto close to fc (Rpullup high) Î phase margin degradation Include the optocoupler pole in the power stage to calculate the phase shift at the crossover frequency. H (s) = www.onsemi.com 43 ηVIN Rload ( RESRCout s + 1) 2α Rsense ( 2Vout + N psVIN ) ( ( Req + RESR )Cout s + 1)( R pullup Copto s + 1) Compensating the QR with TL431 Vout R LED Vdd R upper R pullup C zero VFB TL431 R lower C pole Gnd Low frequency zero R pullup VFB ( s ) G(s) = = −CTR Vout ( s ) RLED Mid-band gain www.onsemi.com 44 ⎛ sRupper C zero + 1 ⎞ ⎛ 1 ⎜⎜ ⎟⎟ ⎜⎜ ⎝ sRupper C zero ⎠ ⎝ 1 + sR pullup C pole Pole at the origin ⎞ ⎟⎟ ⎠ High frequency pole Compensating the QR Converter Calculate fc according to specified Vout undershoot for an output step load. ΔI out fc ≈ ΔVout Cout 2π Calculate RLED to boost the gain at crossover. 60 10 − H ( fc ) 20 fc 30 |H(f)| (dB) RLED = CTR R pullup ) 0 − 30 − 60 1 10 100 3 1×10 f frequency (Hz) www.onsemi.com 45 1×10 4 1×10 5 K Factor Method Needed phase boost: Selected phase margin Power stage phase shift ⎛ Boost ⎞ k = tan⎜ + 45 ⎟ ⎝ 2 ⎠ -101° − 60 − 90 − 120 Selected cross over frequency − 150 − 180 1 10 100 1×10 3 1×10 frequency (Hz) Place the zero at frequency: fc/k C zero = Place the pole at frequency: k*fc C pole = www.onsemi.com 46 PS − 30 Arg [H(f)] (°) Boost = PM − PS − 90 0 1 2π Rupper fc k 1 2π R pullup kf c 4 1×10 5 Loop Compensation Example Specification: ΔVout = 230 mV for ΔIout = 2.8 A ΔI out 2.8 = ⇒ f c = 800 Hz ΔVout Cout 2π 230m × 2.4m × 2π RLED = CTR R pullup 10 − H ( fc ) 20 18k = 0.6 10 22 20 ≈ 1 kΩ Needed Phase Boost: Boost = PM − PS − 90 = 70 − (−101) − 90 = 81° Gain (dB) Calculated mid-band gain: 18.6 dB ) 60 200 48 160 36 120 24 80 12 0 www.onsemi.com 0 a − 12 − 40 − 24 − 80 − 36 − 120 − 48 − 160 − 60 ⎛ 81 ⎞ 10 k = tan ⎜ + 45 ⎟ 12.5 ⎝ 2 ⎠ 1 1 C zero = = = 38 nF ⇒ C zero = 47 nF fc 800 2π × 66k × 2π Rupper 12.5 k 1 1 C pole = = = 0.8 nF ⇒ C pole = 1 nF 2π R pullup kf c 2π × 18k × 12.5 × 800 47 40 PM 100 3 1×10 Frequency (Hz) 4 1×10 − 200 5 1×10 Phase (°) fc ≈ G( f ) H ( f ) Measurement versus Calculation Power stage gain and phase -- Measured gain -- Measured phase -- Calculated gain -- Calculated phase The RHPZ is around 20 kHz. www.onsemi.com 48 Measurement versus Calculation Loop gain and phase -- Measured gain PM -- Measured phase -- Calculated gain -- Calculated phase www.onsemi.com 49 Agenda 1. Quasi-Resonant (QR) Generalities 2. Limiting the free-running frequency 3. Calculating the QR inductor 4. Choosing the Power Components 5. Predicting the Losses of a QR Power Supply 6. Synchronous Rectification 7. Loop Compensation 8. NCP1380, our future QR controller www.onsemi.com 50 NCP1380 Features Operating modes: – QR current-mode with valley lockout for noise immunity – VCO mode in light load for improved efficiency HV-bulk Protections FB 2 CS 3 GND 4 www.onsemi.com NCP1380 C/D Rstart 1 Sampling date: end of January 09 Mass production: end of Feb. 09 51 Ct ZCD / OPP 8 Dovp OVP/BO 7 Vcc 6 DRV 5 Czcd Over power protection Soft-start Short circuit protection Over voltage protection Over temperature protection Brown-Out Rbou Rzcd2 – – – – – – Rzcd1 Rbol Ct Cvcc Control Topology Comparison Fixed Fsw Quasi Resonant Fixed On Time (FOT)(NCP1351) Variable (max power at max Fsw) Best Frequency Fixed Light load efficiencies Normal (with skip mode or freq foldback) Normal Variable (max power at min Fsw) Valley jumping problem (noise) Max Fsw at min Pout Variable (min Pout at min Fsw) Best Best Normal Best BCM (Borderline) CCM/DCM BCM/DCM Transformer size Normal Larger Normal Normal EMI Smaller Normal Smaller Full load efficiencies Operating mode CCM/DCM Normal QR-FOT (NCP1380) QR-FOT: your key to improve standby (FOT) and optimize both efficiency and EMI (QR) for a wide output power range !!! www.onsemi.com 52 QR Mode with Valley Lockout As the load decreases, the controller changes valley (1st to 4th valley) The controller stays locked in a valley until the output power changes significantly. ¾ No valley jumping noise ¾ Natural switching frequency limitation 80000 SWITCHING FREQUENCY (Hz) 70000 60000 50000 4th 3rd 2nd 1st 40000 30000 20000 QR operation 10000 0 0 10 VCO mode www.onsemi.com 53 20 30 OUTPUT POWER (W) 40 50 60 VCO Mode Occurs when VFB < 0.8 V (Pout decreasing) or VFB < 1.6 V (Pout increasing) Fixed peak current (17.5% of Ipk,max), variable frequency set by the FB loop. Ipk max Constant peak current (17.5% of Ipk max) Fsw1 @ Pout1 www.onsemi.com 54 Fsw2 @ Pout2 Pout1 > Pout2 OPP: How does it Work? Laux with flyback polarity swings to –NVIN during the on time. Adjust amount of OPP voltage with Ropu // Ropl. VCS,max = 0.8 V + VOPP Ropu Peak current set point CS ZCD/OPP OPP IpFlag 1 Aux Ropl ESD protection 100% V ILIMIT 60% + - Demag Vth leakage blanking DRV Tblank Non dissipative OPP ! www.onsemi.com 55 370 VIN (V) NCP1380 Versions 4 versions of NCP1380: A, B, C and D OTP OVP NCP1380 / A X X NCP1380 / B X X Auto-Recovery Latched Over current protection Over current protection X X NCP1380 / C X X NCP1380 / D X X www.onsemi.com 56 BO X X Short-Circuit Protection Internal 80-ms timer for short-circuit validation. Additional CS comparator with reduced LEB to detect winding short-circuit. VCS(stop) = 1.5 * VILIMIT S Q DRV Q R CS LEB1 Rsense + FB/4 PWMreset - Down Up OPP Reset V ILIMIT grand reset Laux LEB2 + - V CS(stop) www.onsemi.com 57 TIMER IpFlag ZCD/OPP CsStop Stop controller Short-Circuit Protection (A and C Versions) A and C versions: the fault is latched. ¾ VCC is pulled down to 5 V and waits for ac removal. S DRV Q Q Vdd R aux Vcc latch VCC management CSstop fault CS after LEB1 + FB/4 + V ILIMIT grand reset PWMreset Down Up IpFlag TIMER Reset - + V OPP S Q + CS after LEB2 CSstop Q - VCS(stop) t LEB2 < t LEB1 www.onsemi.com 58 R grand reset VCCstop SCR delatches when ICC < ICCLATCH Short Circuit Protection (B and D) Auto-recovery short circuit protection: the controller tries to restart Auto-recovery imposes a low burst in fault mode. Low average input power in fault condition S Q to DRV stage Vdd aux Q R Vcc VCC management fault CS after LEB1 + FB/4 + V ILIMIT grand reset VCCstop Down Up IpFlag TIMER VCC Reset - + V OPP grand reset + CS after LEB2 - VCS(stop) t LEB2 < t LEB1 www.onsemi.com 59 PWMreset CSstop VDS OVP / OTP (A & B Versions) OVP and OTP detection are achieved by reading the voltage on the pin 7. If the temperature increases, the NTC resistor reduces and VFault decreases. When VFault < VOTP Î the controller is latched. If VCC increases, the zener diode injects current in the clamp circuit. When VFault > VOVP Î the controller is latched. Vcc Vdd VOVP noise delay - Dz IOTP(REF) + OVPcomp Fault 7 S Q Rclamp Clamp - Vclamp VOTP R OTPcomp SS end www.onsemi.com 60 Q + NTC noise delay grand reset Latch BO / OVP (C & D Versions) ¾ If Vpin7 > BO threshold & VCC > VCCon, the controller starts pulsing. ¾ The hysteresis current source is ON when Vpin7 > BO threshold. BO HV-Bulk noise delay Vcc + S S Dz VOVP Rbou DRV Q - Q Q Latch Q OVP/BO R 7 Vdd R IBO grand reset Rbol noise delay Rclamp Clamp + BO reset Vclamp OVP VBO ¾ If VCC > BVDz, the zener diode injects current inside the clamp resistor. ¾ When Vpin7 reaches the OVP threshold, the controller is latched. www.onsemi.com 61 CS comp Conclusion Changing valley as the load decreases is a way to limit the maximum switching frequency in QR power supplies. Lots of equations to predict the efficiency of the power supply, but good matching between calculations and the measurement. Synchronous rectification increases the efficiency of the QR power supply but increases also the power consumption in standby. Friendly compensation for QR power supply (DCM: 1st order system) NCP1380 features: • QR current-mode with valley lockout for noise immunity for high load. • VCO mode in light load for improved efficiency. www.onsemi.com 62 For More Information • View the extensive portfolio of power management products from ON Semiconductor at www.onsemi.com • View reference designs, design notes, and other material supporting the design of highly efficient power supplies at www.onsemi.com/powersupplies www.onsemi.com 63