Datasheet

DATASHEET
Single-Phase R4™ Digital Hybrid PWM Controller with
Integrated Driver, PMBus/SMBus/I2C and PFM
ISL68200
Features
The ISL68200 is a single-phase synchronous-buck PWM
controller featuring Intersil’s proprietary R4™ Technology. It
supports a wide 4.5V to 24V input voltage range and a wide
0.5V to 5.5V output range. Integrated LDOs provide controller
bias voltage, allowing for single supply operation. The
ISL68200 includes a PMBus/SMBus/I2C interface for device
configuration and telemetry (VIN, VOUT, IOUT and temperature)
and fault reporting.
• Intersil’s proprietary R4™ Technology
- Linear control loop for optimal transient response
- Variable frequency and duty cycle control during load
transient for fastest possible response
- Inherent voltage feed-forward for wide range input
Intersil’s proprietary R4™ control scheme has extremely fast
transient performance, accurately regulated frequency control
and all internal compensation. An efficiency enhancing PFM
mode can be enabled to greatly improve light-load efficiency.
The ISL68200’s series bus allows for easy R4™ loop
optimization, resulting in fast transient performance over a
wide range of applications, including all ceramic output filters.
• ±0.5% DAC accuracy with remote sense
Built-in MOSFET drivers minimize external components,
significantly reducing design complexity and board space,
while also lowering BOM cost. The 4A drive strength allows for
faster switching time, improving regulator efficiency. An
integrated high-side gate-to-source resistor helps avoid Miller
coupling shoot-through and improve system reliability.
The ISL68200 has four 8-bit configuration pins, which provide
very flexible configuration options (frequency, VOUT, R4™ gain,
etc.) without the need for built-in NVM memory. This results in
a design flow that closely matches traditional analog
controllers, while still offering the design flexibility and feature
set of a digital PMBus/SMBus/I2C interface. The ISL68200
also features remote voltage sensing and completely eliminates
any potential difference between remote and local grounds. This
improves regulation and protection accuracy. A precision enable
input is available to coordinate the start-up of the ISL68200 with
other voltage rails, especially useful for power sequencing.
Applications
• Input voltage range: 4.5V to 24V
• Output voltage range: 0.5V to 5.5V
• Support all ceramic solutions
• Integrated LDOs for single input rail solution
• SMBus/PMBus/I2C compatible, up to 1.25MHz
• 256 boot-up voltage levels with a configuration pin
• Eight switching frequency options from 300kHz to 1.5MHz
• PFM operation option for improved light-load efficiency
• Start-up into precharged load
• Precision enable input to set higher input UVLO and power
sequence as well as fault reset
• Power-good monitor for soft-start and fault detection
• Comprehensive fault protection for high system reliability
- Over-temperature protection
- Output overcurrent and short-circuit protection
- Output overvoltage and undervoltage protection
- Open remote sense protection
- Integrated high-side gate-to-source resistor to prevent self
turn-on due to high input bus dv/dt
• Integrated power MOSFETs 4A drivers with adaptive
shoot-through protection and bootstrap function
• Compatible with Intersil’s PowerNavigator™ software
• High efficiency and high density POL digital power
Related Literature
• FPGA, ASIC and memory supplies
UG067, “ISL68200DEMO1Z Demonstration Board User Guide”
• Datacenter: servers, storage systems
• Wired infrastructure: routers/switches/optical networking
• Wireless infrastructure: base station
TABLE 1. SINGLE-PHASE R4™ DIGITAL HYBRID PWM CONTROLLER OPTIONS
PART
NUMBER
INTEGRATED
DRIVER
PWM
OUTPUT
PMBus/SMBus/I2C
INTERFACE
ISL68200
Yes
No
Yes
Discrete MOSFETs or Dual Channel MOSFETs
ISL68201
No
Yes
Yes
Intersil Power Stages: ISL99140
Intersil Drivers: ISL6596, ISL6609, ISL6627, ISL6622, ISL6208
March 7, 2016
FN8705.1
1
COMPATIBLE DEVICES
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2016. All Rights Reserved.
Intersil (and design), R3 Technology, R4 Technology and PowerNavigator are trademarks owned by Intersil Corporation or
one of its subsidiaries. All other trademarks mentioned are the property of their respective owners.
ISL68200
Table of Contents
Typical Applications Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Block Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Functional Pin Descriptions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Recommended Operating Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
IC Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Enable and Disable . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Resistor Reader (Patented) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Soft-Start. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Boot-Up Voltage Programming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Current Sensing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Thermal Monitoring and Compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
IOUT Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Fault Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
PGOOD Monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Adaptive Shoot-Through Protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
PFM Mode Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
SMBus, PMBus and I2C Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
R4™ Modulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
10
10
10
10
12
12
16
17
19
20
21
21
21
21
26
General Application Design Guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output Filter Design. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input Capacitor Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Design and Layout Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Voltage Regulator Design Materials . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
28
28
28
29
30
Revision History. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
About Intersil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
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ISL68200
Typical Applications Circuits
1.0µF
4.7µF
VCC
7VLDO
PVCC
4.75V TO 24V
VIN
1.0µF
0.1µF
I2C/SMBus/
PMBus
SALERT
SCL
SDA
PGOOD
PGOOD
EN
BOOT
V OUT < 7VLDO - 1.7V
UGATE
0.5V TO 5.5V
PHASE
EN
VCC
IOUT
LGATE
10k
NTC
VCC
VCC
NTC
4
1.54k
0.1µF NCP15XH103J03RC
BETA = 3380
PROG1-4
CSEN
CSRTN
VSEN
RGND
GND
FIGURE 1. WIDE RANGE INPUT AND OUTPUT APPLICATIONS
1.0µF
4.7µF
VCC
7VLDO
PVCC
VIN
1.0µF
2
I C/SMBus/
PMBus
SALERT
SCL
SDA
PGOOD
PGOOD
EN
4.5V TO 5.5V
0.1µF
BOOT
V OUT < 7VLDO - 1.7V
UGATE
0.5V TO 2.5V
PHASE
EN
VCC
IOUT
LGATE
10k
NTC
VCC
VCC
NTC
4
1.54k
0.1µF NCP15XH103J03RC
BETA = 3380
PROG1-4
CSEN
CSRTN
VSEN
RGND
GND
FIGURE 2. 5V INPUT APPLICATION
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ISL68200
Block Diagram
PROG4
SCL
SDA SALERT
PROG2 PROG3
7VLDO VINPVCC
POR
VCC
SOFT-START
AND
FAULT LOGIC
SMBUS/PMBUS/I2C
INTERFACE
BOOT
EN
OTP OCP
PGOOD
VIN VOUT IOUT TEMP
DRIVER
UGATE
PGOOD
CIRCUITRY
PHASE
DEAD TIME
GENERATION
RGND
VSEN
INTERNAL
COMPENSATION
+ AMPLIFIER
PVCC
OVERVOLTAGE/
UNDERVOLTAGE
DRIVER
LGATE
GND
5V LDO
R4™
MODULATOR
PROG1
VIN
7V LDO
REFERENCE
VOLTAGE
CIRCUITRY
OVERCURRENT (OCP) AND
OVER-TEMPERATURE (OTP)
CURRENT SENSE
AND TEMPERATURE
COMPENSATION
SWITCHING
FREQUENCY
7VLDO
CSEN
CSRTN
NTC
GND
IOUT
FIGURE 3. SIMPLIFIED FUNCTIONAL BLOCK DIAGRAM OF ISL68200
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ISL68200
Pin Configuration
PVCC
BOOT
UGATE
PHASE
LGATE
GND
ISL68200
(24 LD 4x4 QFN)
TOP VIEW
24
23
22
21
20
19
EN
1
18 PROG1
VIN
2
17 PROG2
7VLDO
3
VCC
4
SCL
5
14 IOUT
SALERT
6
13 NTC
9
10
11
12
CSEN
8
CSRTN
7
VSEN
15 PROG4
RGND
GND PAD
PGOOD
16 PROG3
SDA
25
Functional Pin Descriptions
PIN NUMBER
SYMBOL
DESCRIPTION
1
EN
Precision Enable input. Pulling EN above the rising threshold voltage initiates the soft-start sequence, while pulling EN below the
failing threshold voltage suspends the Voltage Regulator (VR) operation.
2
VIN
Input voltage pin for R4™ loop and LDOs (5V and 7V). Place a high quality low ESR ceramic capacitor (1.0μF, X7R) in close
proximity to the pin. External series resistor is not advised.
3
7VLDO
7V LDO from VIN is used to bias current sensing amplifier. Place a high quality low ESR ceramic capacitor (1.0μF, X7R, 10V+)
in close proximity to the pin.
4
VCC
Logic bias supply that should be connected to PVCC rail externally. Place a high quality low ESR ceramic capacitor (1.0μF,
X7R) from this pin to GND.
5
SCL
Synchronous clock signal input of SMBus/PMBus/I2C.
6
SALERT
7
SDA
8
PGOOD
9
RGND
This pin monitors the negative rail of regulator output. Connect to ground at point of regulation.
10
VSEN
This pin monitors the positive rail of regulator output. Connect to point of regulation
11
CSRTN
This pin monitors the negative flow of output current for overcurrent protection and telemetry.
12
CSEN
This pin monitors the positive flow of output current with a series resistor and for overcurrent protection and telemetry. The
series resistor sets the current gain and should be within 40Ωand 3.5kΩ.
13
NTC
Input pin for the temperature measurement. Connect this pin through an NTC thermistor (10kΩ,  ~ 3380) and a decoupling
capacitor (~0.1μF) to GND and a resistor (1.54kΩ)to VCC of the controller. The voltage at this pin is inversely proportional to
the VR temperature.
14
IOUT
Output current monitor pin. An external resistor sets the gain and an external capacitor provides the averaging function; an
external pull-up resistor to VCC is recommended to calibrate the no load offset. See “IOUT Calibration” on page 19.
15
PROG4
Programming pin for Modulator (R4™) RR impedance and output slew rate during Soft-Start (SS) and Dynamic VID (DVID).
It also sets AV gain multiplier to 1x or 2x and determines the AV gain on PROG3.
16
PROG3
Programming pin for ultrasonic PFM operation, fault behavior, switching frequency and R4™ (AV) control loop gain.
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Output pin for transferring the active low signal driven asynchronously from the VR controller to SMBus/PMBus.
I/O pin for transferring data signals between SMBus/PMBus/I2C host and VR controller.
Open-drain indicator output.
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Functional Pin Descriptions (Continued)
PIN NUMBER
SYMBOL
17
PROG2
Programming pin for PWM/PFM mode, temperature compensation and serial bus (SMBus/PMBus/I2C) address.
18
PROG1
Programming pin for boot-up voltage.
19
GND
20
LGATE
Low-side MOSFET gate driver output. Connect to the gate terminal of the low-side MOSFET of the converter.
21
PHASE
Return path for the UGATE high-side MOSFET driver, and zero inductor current detector input for diode emulation.
22
UGATE
High-side MOSFET gate driver output. Connect to the gate terminal of the high-side MOSFET of the converter.
23
BOOT
Positive input supply for the UGATE high-side MOSFET gate driver. Connect an MLCC (0.22µF, X7R) between BOOT and
PHASE pins.
24
PVCC
Output of the 5V LDO and input for the LGATE and UGATE MOSFET driver circuits. Place a high quality low ESR ceramic
capacitor (4.7μF, X7R) in close proximity to the pin.
25
DESCRIPTION
Return current path for the LGATE MOSFET driver. Connect directly to system ground plane.
GND PAD Return of logic bias supply VCC. Connect directly to system ground plane with at least 5 vias.
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
ISL68200IRZ
ISL68200DEMO1Z
PART
MARKING
TEMP RANGE
(°C)
PACKAGE
(RoHS Compliant)
ISL 68200I
-40 to +85
24 Ld 4x4 QFN
PKG.
DWG. #
L24.4x4C
20A Demonstration Board with on-board transient
NOTES:
1. Add “-T” suffix for 6k units, “-T7A” = suffix for 250 units and “-TK” for 1k units. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see product information page for ISL68200. For more information on MSL please see techbrief TB363.
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ISL68200
Absolute Maximum Ratings
Thermal Information
VCC, PVCC, VSEN. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7.0V
Input Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +27V
7VLDO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to GND, 7.75V
BOOT Voltage (VBOOT-GND) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 33V
BOOT to PHASE Voltage (VBOOT-PHASE) . . . . . . . . . . . . . . . . -0.3V to 7V (DC)
-0.3V to 9V (<10ns)
PHASE Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . (GND - 0.3V) to 28V
(GND - 9V) (<20ns Pulse Width, 10µJ)
UGATE Voltage. . . . . . . . . . . . . . . . . . . . . . . . . (VPHASE - 0.3V) (DC) to VBOOT
(VPHASE - 5V) (<20ns Pulse Width, 10µJ) to VBOOT
LGATE Voltage . . . . . . . . . . . . . . . . . . . . . . . (GND - 0.3V) (DC) to VCC + 0.3V
(GND - 2.5V) (<20ns Pulse Width, 5µJ) to VCC + 0.3V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to GND, VCC + 0.3V
ESD Ratings
Machine Model (Tested per JESD22-A115C) . . . . . . . . . . . . . . . . . . 200V
Charged Device Model (Tested per JS-002-2014) . . . . . . . . . . . . . . . 1kV
Human Body Model (Tested per JS-001-2010) . . . . . . . . . . . . . . . . .2.5kV
Latch-Up (Tested per JESD78D, Class 2, Level A) . . . . ±100mA at +125°C
Thermal Resistance (Typical)
JA (°C/W) JC (°C/W)
24 Ld QFN (Notes 4, 5) . . . . . . . . . . . . . . . .
39
2.5
Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-55°C to +150°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see TB493
Recommended Operating Conditions
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C
Wide Range Input Voltage, VIN, Figure 1 . . . . . . . . . . . . . . . . . 4.75V to 24V
5V Application Input Voltage, VIN, Figure 2 . . . . . . . . . . . . . . . . 4.5V to 5.5V
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
-40°C to +85°C, unless otherwise stated.
All typical specifications TA = +25°C, VCC = 5V. Boldface limits apply across the operating temperature range,
PARAMETER
SYMBOL
MIN
(Note 6)
TYP
MAX
(Note 6)
UNIT
EN = 5V, VCC = 5V, fSW = 500kHz, DAC = 1V
14
16.5
mA
EN = 0V, VCC = 5V
14
16.5
EN = 5V, VCC = 5V, fSW = 500kHz, DAC = 1V
2
TEST CONDITIONS
VCC AND PVCC
VCC Input Bias Current
IVCC
PVCC Input Bias Current
IPVCC
EN = 0V, VCC = 5V
mA
mA
1.0
mA
4.20
4.35
V
3.95
4.15
V
VCC AND VIN POR THRESHOLD
VCC, PVCC Rising POR Threshold Voltage
VCC, PVCC Falling POR Threshold Voltage
3.80
VIN, 7VLDO Rising POR Threshold Voltage
VIN, 7VLDO Falling O POR Threshold Voltage
4.20
4.35
V
3.80
3.95
4.15
V
ENABLE INPUT
EN High Threshold Voltage
VENTHR
0.81
0.84
0.87
V
EN Low Threshold Voltage
VENTHF
0.71
0.76
0.81
V
DAC ACCURACY
DAC Accuracy
(TA = 0°C to +85°C)
DAC Accuracy
(TA = -45°C to +85°C)
Submit Document Feedback
7
2.5V < DAC ≤ 5.5V
-0.5
0.5
%
1.6V < DAC ≤ 2.5V
-0.75
0.75
%
1.2V < DAC ≤ 1.6V
-10
10
mV
0.5V ≤ DAC ≤ 1.2V
-8
8
mV
2.5V < DAC ≤ 5.5V
-0.75
0.75
%
1.6V < DAC ≤ 2.5V
-1.0
1.0
%
1.2V < DAC ≤ 1.6V
-11
11
mV
0.5V ≤ DAC ≤ 1.2V
-9
9
mV
FN8705.1
March 7, 2016
ISL68200
Electrical Specifications
All typical specifications TA = +25°C, VCC = 5V. Boldface limits apply across the operating temperature range,
-40°C to +85°C, unless otherwise stated. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNIT
CHANNEL FREQUENCY
300kHz Configuration
PWM mode
260
300
335
kHz
400kHz Configuration
PWM mode
345
400
450
kHz
500kHz Configuration
PWM mode
435
500
562
kHz
600kHz Configuration
PWM mode
510
600
670
kHz
700kHz Configuration
PWM mode
610
700
790
kHz
850kHz Configuration
PWM mode
730
850
950
kHz
1000kHz Configuration
PWM mode
865
1000
1120
kHz
1500kHz Configuration
PWM mode
1320
1500
1660
kHz
0.0616
0.078
0.096
mV/µs
0.13
0.157
0.18
mV/µs
0.25
0.315
0.37
mV/µs
0.53
0.625
0.70
mV/µs
1.05
1.25
1.40
mV/µs
2.10
2.50
2.80
mV/µs
4.20
5.00
5.60
mV/µs
8.60
10.0
10.9
mV/µs
140
200
260
µs
250
µA
SOFT-START AND DYNAMIC VID
Soft-Start and DVID Slew Rate
Soft-Start Delay from Enable High
Excluding 5.5ms POR timeout, See Figures 22
and 23 on page 22
REMOTE SENSE
Bias Current of VSEN and RGND Pins
Maximum Differential Input Voltage
6.0
V
POWER-GOOD
PGOOD Pull-Down Impedance
RPG
PGOOD = 5mA sink
PGOOD Leakage Current
IPG
PGOOD = 5V
10
50
Ω
1.0
µA
5.15
V
LDOs
5V LDO Regulation
VIN = 12V, load = 50mA
4.85
5V LDO Regulation
VIN = 4.75V, load = 50mA
4.45
V
125
mA
5V LDO Current Capability
7V LDO Regulation
250µA load
7V Dropout
VIN = 4.75V, 250µA load
7V LDO Current Capability
Not recommended for external use
7.2
5.00
7.4
7.5
V
4.50
V
2
mA
CURRENT SENSE
Average OCP Trip Level
82
IOC_TRIP
Short-Circuit Protection Threshold
100
123
130
µA
% IOCP
Sensed Current Tolerance
74
78
83
µA
Sensed Current Tolerance
35
38
42
µA
Maximum Common-Mode Input Voltage
Submit Document Feedback
8
7VLDO = 7.4V
5.7
V
VCC = PVCC = 7VLDO = 4.5V
2.8
V
FN8705.1
March 7, 2016
ISL68200
Electrical Specifications
All typical specifications TA = +25°C, VCC = 5V. Boldface limits apply across the operating temperature range,
-40°C to +85°C, unless otherwise stated. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNIT
FAULT PROTECTION
UVP Threshold Voltage
Latch
68
74
80
% DAC
Start-Up OVP Threshold Voltage
0V ≤ VBOOT ≤ 1.08V
1.10
1.15
1.25
V
1.08V < VBOOT ≤ 1.55V
1.58
1.65
1.75
V
1.55V < VBOOT ≤ 1.85V
1.88
1.95
2.05
V
1.85V < VBOOT ≤ 2.08V
2.09
2.15
2.25
V
2.08V < VBOOT ≤ 2.53V
2.56
2.65
2.75
V
2.53V < VBOOT ≤ 3.33V
3.36
3.45
3.6
V
3.33V < VBOOT ≤ 5.5V
5.52
5.65
5.85
Start-Up OVP Hysteresis
100
V
mV
OVP Rising Threshold Voltage
VOVRTH
0.5 ≤ DAC ≤ 5.5
114
120
127
% DAC
OVP Falling Threshold Voltage
VOVFTH
0.5 ≤ DAC ≤ 5.5
96
100
108
% DAC
Over-Temperature Shutdown Threshold
READ_TEMP = 72h
20
22.31
26
% VCC
Over-Temperature Shutdown Reset Threshold
READ_TEMP = 8Eh
25
27.79
30
% VCC
1
V
SMBus/PMBus/I2C
Signal Input Low Voltage
Signal Input High Voltage
1.6
Signal Output Low Voltage
V
4mA pull-up current
DATE, ALERT # Pull-Down Impedance
11
CLOCK Maximum Speed
0.4
V
50
Ω
1.25
MHz
CLOCK Minimum Speed
0.05
Telemetry Update Rate
108
Timeout
25
PMBus Accessible Timeout from All Rails’ POR
MHz
µs
30
35
ms
See Figure 22 on page 22
5.5
6.5
ms
1.0
GATE DRIVER
UGATE Pull-Up Resistance
RUGPU
200mA source current
Ω
UGATE Source Current
IUGSRC
UGATE - PHASE = 2.5V
2.0
A
UGATE Sink Resistance
RUGPD
250mA sink current
1.0
Ω
UGATE Sink Current
IUGSNK
UGATE - PHASE = 2.5V
2.0
A
LGATE Pull-Up Resistance
RLGPU
250mA source current
1.0
Ω
LGATE Source Current
ILGSRC
LGATE - GND = 2.5V
2.0
A
LGATE Sink Resistance
RLGPD
250mA sink current
0.5
Ω
LGATE Sink Current
ILGSNK
LGATE - GND = 2.5V
4.0
A
UGATE to LGATE Dead Time
tUGFLGR
UGATE falling to LGATE rising, no load
10
ns
LGATE to UGATE Dead Time
tLGFUGR
LGATE falling to UGATE rising, no load
18
ns
BOOTSTRAP DIODE
ON-Resistance
RF
16
30
Ω
NOTE:
6. Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design.
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9
FN8705.1
March 7, 2016
ISL68200
Operation
The following sections will provide a detailed description of the
ISL68200 operation.
IC Supplies
The ISL68200 has 4 bias pins: VIN, 7VLDO, PVCC and VCC. The
PVCC and 7VLDO voltage rails are 5V LDO and 7.4V LDO supplied
by VIN, respectively, while the VCC pin needs to connect to PVCC
rail externally to be biased. For 5V input applications, all these
pins should be tied together and biased by a 5V supply. Since the
VIN pin voltage information is used by the R4™ Modulator loop,
the user CANNOT bias VIN with a series resistor. In addition, the
VIN pin CANNOT be biased independently from other rails.
Enable and Disable
The IC is disabled until the 7VLDO, PVCC, VCC, VIN and EN pins
increase above their respective rising threshold voltages and the
typical 5.5ms timeout (worst case = 6.5ms) expires, as shown in
Figures 22 and 23 on page 22. The controller will become
disabled when the 7VLDO, PVCC, VCC, VIN or EN pins drop below
their respective falling POR threshold voltages.
The precision threshold EN pin allows the user to set a precision
input UVLO level with an external resistor divider, as shown in
Figure 4. For 5V input applications or wide range input
applications, the EN pin can directly connect to VCC, as shown in
Figure 5. If an external enable control signal is available and is an
open-drain signal, a pull-up impedance (100k or higher) can be
used.
EXTERNAL CIRCUIT
ISL68200
VIN
100k
SOFTSTART
In addition, based upon ON_OFF_CONFIG [02h] setting, the IC be
enabled or disabled by series bus command “OPERATION [01h]”
and/or EN pin. See Table 11 on page 25 for more details.
Resistor Reader (Patented)
The ISL68200 offers four programming pins to customize their
regulator specifications. The details of these pins are summarized
in Table 2, followed by the detailed description of resistor reader
operation.
TABLE 2. DEFINITION OF PROG PINS
PIN
BIT
NAME
DESCRIPTION
PROG1 [7:0]
BOOT-UP
VOLTAGE
Set output boot-up voltage, 256 different
options: 0, 0.5V to 5.5V (see Table 7)
PROG2 [7:7]
PWM/PFM
[6:5]
Enables PFM mode or forced PWM.
Temperature Adjust NTC temperature compensation:
Compensation OFF, +5, +15, +30°C.
[4:0]
ADDR
Set serial bus 32 different addresses
(see Table 10).
PROG3 [7:7]
uSPFM
Ultrasonic (25kHz clamp) PFM enable
[6:6] Fault Behavior OCP fault behavior:
Latch, Infinite 9ms retry
[5:3]
FSW
[2:0]
R4™ Gain
PROG4 [7:5]
RAMP_RATE
Set switching frequency (fSW).
Set error amplifier gain (AV).
Set soft-start and DVID ramp rate.
[4:3]
RR
Select RR impedance for R4™ loop.
[2:2]
AVMLTI
Select AV Gain Multiplier (1x or 2x)
[1:0]
Not Used
Intersil has developed a high resolution ADC using a patented
technique with a simple 1%, 100ppm/K or better temperature
coefficient resistor divider. The same type of resistors are
preferred so that it has similar change over-temperature. In
addition, the divider is compared to the internal divider off VCC
and GND nodes and therefore must refer to VCC and GND pins,
not through any RC decoupling network.
EN
9.09k
VIN UVLO = 10.2V/9.24V
FIGURE 4. INPUT UVP CONFIGURATION
ISL68200
EXTERNAL CIRCUIT
ISL68200
VCC
REGISTER
TABLE
REN
SOFTSTART
EXTERNAL CIRCUIT
VCC
RUP
ADC
OPTIONAL
EN
VIN UVLO = 4.20/3.95V
REN is ONLY needed when the user wants to
control the IC with an external enable signal
RDW
FIGURE 6. SIMPLIFIED RESISTOR DIVIDER ADC
FIGURE 5. 5V INPUT OR WIDE RANGE INPUT CONFIGURATION
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10
FN8705.1
March 7, 2016
ISL68200
Tables 3 through 6 show the RUP and RDW values of each pin for
a specific system design with some tie-high and tie-low options,
which are for easy programming with reduced resistors and can
be used to validate the regulator operation during In-Circuit Test
(ICT) for 0V boot-up voltage option. Additional options are
available using Intersil’s PowerNavigator™ or Resistor Reader
calculator, please contact Intersil Application support at
www.intersil.com/en/support. DATA for corresponding registers
can be read out via series bus command (DC to DF). Note that
more options are in PowerNavigator™ GUI or Resistor Reader
calculator and the case of 10kΩ tie-high or tie-low is equivalent
0Ω tie-high or tie-low.
TABLE 3. PROG 1 RESISTOR READER EXAMPLE
TABLE 5. PROG 3 RESISTOR READER EXAMPLE
PROG3
(DE)
RUP
(kΩ)
00h
Open
0
Disabled
20h
Open
21.5
40h
Open
60h
RDW ULTRASONIC
FAULT
(kΩ)
PFM
BEHAVIOR
R4 GAIN
fSW
(kHz)
1x
2x
Retry
300
42
84
Disabled
Retry
700
42
84
34.8
Disabled
Latch
300
42
84
Open
52.3
Disabled
Latch
700
42
84
80h
Open
75
Enabled
Retry
300
42
84
A0h
Open
105
Enabled
Retry
700
42
84
C0h
Open
147
Enabled
Latch
300
42
84
E0h
Open
499
Enabled
Latch
700
42
84
PROG1 (DC)
RUP
(kΩ)
RDW
(kΩ)
VOUT
(V)
00h
Open
0
0.797
1Fh
0
Open
Disabled
Retry
600
1
2
20h
Open
20
0.852
3Fh
21.5
Open
Disabled
Retry
1500
1
2
40h
Open
34.8
0.898
60h
Open
52.3
0.953
5Fh
34.8
Open
Disabled
Latch
600
1
2
80h
Open
75
1.000
7Fh
52.3
Open
Disabled
Latch
1500
1
2
A0h
Open
105
1.047
9Fh
75
Open
Enabled
Retry
600
1
2
C0h
Open
147
1.102
BFh
105
Open
Enabled
Retry
1500
1
2
E0h
Open
499
1.203
DFh
147
Open
Enabled
Latch
600
1
2
1Fh
0
Open
1.352
FFh
499
Open
Enabled
Latch
1500
1
2
3Fh
20
Open
1.500
5Fh
34.8
Open
1.797
7Fh
52.3
Open
2.500
9Fh
75
Open
3.000
BFh
105
Open
DFh
147
FFh
499
TABLE 6. PROG 4 RESISTOR READER EXAMPLE
PROG4
(DF
RUP
(kΩ)
RDW
(kΩ)
SS RATE
(mV/µs)
(kΩ
AVMLTI
3.297
00h
Open
0
1.25
200
1
Open
5.000
20h
Open
20
2.5
200
1
Open
0.000
40h
Open
34.8
5
200
1
60h
Open
52.3
10
200
1
TABLE 4. PROG 2 RESISTOR READER EXAMPLE
PROG2
(DD)
RUP
(kΩ)
RDW
(kΩ)
PWM/PFM
TEMP
COMP
RR
PM_ADDR
(7-BIT)
80h
Open
75
0.078
200
1
A0h
Open
105
0.157
200
1
00h
Open
0
Enabled
30
60h
C0h
Open
147
0.315
200
1
20h
Open
20
Enabled
15
60h
E0h
Open
499
0.625
200
1
40h
Open
34.8
Enabled
5
60h
1Fh
0
Open
1.25
800
2
60h
Open
52.3
Enabled
OFF
60h
80h
Open
75
Disabled
30
60h
3Fh
20
Open
2.5
800
2
A0h
Open
105
Disabled
15
60h
5Fh
34.8
Open
5
800
2
C0h
Open
147
Disabled
5
60h
7Fh
52.3
Open
10
800
2
E0h
Open
499
Disabled
OFF
60h
9Fh
75
Open
0.078
800
2
1Fh
0
Open
Enabled
30
7F
BFh
105
Open
0.157
800
2
3Fh
20
Open
Enabled
15
7F
DFh
147
Open
0.315
800
2
5Fh
34.8
Open
Enabled
5
7F
FFh
499
Open
0.625
800
2
7Fh
52.3
Open
Enabled
OFF
7F
9Fh
75
Open
Disabled
30
7F
BFh
105
Open
Disabled
15
7F
DFh
147
Open
Disabled
5
7F
FFh
499
Open
Disabled
OFF
7F
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FN8705.1
March 7, 2016
ISL68200
Soft-Start
The ISL68200 based regulator has 4 periods during soft-start, as
shown in Figure 7 on page 12. After a 5.5ms timeout (worst
case = 6.5ms) of bias supplies, as shown in Figures 22 and 23 on
page 22, once the EN pin reaches above its enable threshold, the
controller begins the first soft-start ramp after a fixed soft-start
delay period of tD1. The output voltage reaches the boot-up voltage
(VBOOT) at a fixed slew rate in period tD2. Then, the controller will
regulate the output voltage at VBOOT for another period tD3 until the
SMBus/PMBus/ I2C sends a new VOUT command. If the VOUT
command is valid, the ISL68200 will initiate the ramp until the
voltage reaches the new VOUT command voltage in period tD4. The
soft-start time is the sum of the 4 periods, as shown in
Equation 1.
t SS = t D1 + t D2 + t D3 + t D4
(EQ. 1)
tD1 is a fixed delay with the typical value as 200µs. tD3 is
determined by the time to obtain a new valid VOUT command
voltage from the SMBus/PMBus/I2C bus. If the VOUT command is
valid before the output reaches the boot-up voltage, the output will
turn around to respond to the new VOUT command code.
VBOOT < PRECHARGED < OVP
VOUT
ISL68200 supports precharged load start-up up to the maximum
VOUT of 5.5V with sufficient boot capacitor charge. For an
extended precharged load, the boot capacitor will be discharged
to “PVCC - VOUT - VD” by high-side drive circuits’ standby current.
For instance, an extended 4V precharged load, the boot capacitor
will reduce to a less than 1V boot capacitor voltage, which is
insufficient to power-up the VR. In this case, it is recommended
to let the output drop below 2.5V with an external bleed resistor
before issuing another soft-start command.
Boot-Up Voltage Programming
An 8-bit pin PROG1 is dedicated for the boot-up voltage
programmability, which offers 256 options 0V and 0.5V to 5.5V,
as in Table 7. The most popular boot-up voltage levels are placed
on the tie-low spots (0h, 20h, 40h, 60h, 80h, A0h, C0h, E0h) and
the tie-high spots (1Fh, 3Fh, 5Fh, 7Fh, 9Fh, BFh, DFh, FFh) for
easy programming, as summarized in Table 3. 0V boot-up
voltage is considered as “OFF,” the driver will be in tri-state and
the internal DAC will set to 0V.
In addition, if the VOUT_COMMAND (21h) is executed
successfully 5.5ms (typically, worst 6.5ms) after VCC POR and
prior to Enable, it will override the boot-up voltage set by the
PROG1 pin.
TABLE 7. PROG1 8-BIT (BOOT-UP VOLTAGE)
VBOOT
PRECHARGED < VBOOT
0V
tD2
tD1
tD3 tD4
EN
PGOOD
FIGURE 7. SOFT-START WAVEFORMS
During tD2 and tD4, ISL68200 digitally controls the DAC voltage
change. The ramp time tD2 and tD4 can be calculated based on
Equations 2 and 3, once the slew rate is set by the PROG4 pin.
V BOOT
t D2 = --------------------------------------  s 
RAMP_RATE
(EQ. 2)
V OUT – V BOOT
t D4 = ------------------------------------------  s 
RAMP_RATE
(EQ. 3)
The ISL68200 supports precharged start-up, it initiates the first
PWM pulse until the internal reference (DAC) reaches the
pre-charged level at RAMP_RATE, programmed by PROG4 or
D5[2:0]. When the precharged level is below VBOOT, the output
walks up to the VBOOT at RAMP_RATE and releases PGOOD at
tD1 + tD2, when the precharged output is above VBOOT but below
OVP, it walks down to VBOOT at RAMP_RATE and then releases
PGOOD at tD1 +tD2, in which tD2 is defined in Equation 4 and
longer than a normal start-up.
V PRECHARGED V PRECHARGED – V BOOT
t D2 = -------------------------------------------- + -----------------------------------------------------------------------  s 
RAMP_RATE
RAMP_RATE
Submit Document Feedback
12
(EQ. 4)
DELTA FROM
PREVIOUS
CODE (mV)
HEX CODE
VBOOT
(V)
VOUT
COMMAND
CODE (HEX)
00000000
0
0.7969
66
00000001
1
0.5000
40
00000010
2
0.5078
41
7.8125
00000011
3
0.5156
42
7.8125
00000100
4
0.5234
43
7.8125
00000101
5
0.5313
44
7.8125
00000110
6
0.5391
45
7.8125
00000111
7
0.5469
46
7.8125
00001000
8
0.5547
47
7.8125
00001001
9
0.5625
48
7.8125
00001010
A
0.5703
49
7.8125
00001011
B
0.5781
4A
7.8125
00001100
C
0.5859
4B
7.8125
00001101
D
0.5938
4C
7.8125
00001110
E
0.6016
4D
7.8125
00001111
F
0.6094
4E
7.8125
00010000
10
0.6172
4F
7.8125
00010001
11
0.6250
50
7.8125
00010010
12
0.6328
51
7.8125
00010011
13
0.6406
52
7.8125
00010100
14
0.6484
53
7.8125
00010101
15
0.6563
54
7.8125
BINARY
CODE
FN8705.1
March 7, 2016
ISL68200
TABLE 7. PROG1 8-BIT (BOOT-UP VOLTAGE) (Continued)
TABLE 7. PROG1 8-BIT (BOOT-UP VOLTAGE) (Continued)
HEX CODE
VBOOT
(V)
VOUT
COMMAND
CODE (HEX)
DELTA FROM
PREVIOUS
CODE (mV)
BINARY
CODE
HEX CODE
VBOOT
(V)
VOUT
COMMAND
CODE (HEX)
DELTA FROM
PREVIOUS
CODE (mV)
00010110
16
0.6641
55
7.8125
00111101
3D
0.9531
7A
7.8125
00010111
17
0.6719
56
7.8125
00111110
3E
0.9609
7B
7.8125
00011000
18
0.6797
57
7.8125
00111111
3F
1.5000
C0
00011001
19
0.6875
58
7.8125
01000000
40
0.8984
73
00011010
1A
0.6953
59
7.8125
01000001
41
0.9688
7C
7.8125
00011011
1B
0.7031
5A
7.8125
01000010
42
0.9766
7D
7.8125
00011100
1C
0.7109
5B
7.8125
01000011
43
0.9844
7E
7.8125
00011101
1D
0.7188
5C
7.8125
01000100
44
0.9922
7F
7.8125
00011110
1E
0.7266
5D
7.8125
01000101
45
1.0000
80
7.8125
00011111
1F
1.3516
AD
01000110
46
1.0078
81
7.8125
00100000
20
0.8516
6D
01000111
47
1.0156
82
7.8125
00100001
21
0.7344
5E
7.8125
01001000
48
1.0234
83
7.8125
00100010
22
0.7422
5F
7.8125
01001001
49
1.0313
84
7.8125
00100011
23
0.7500
60
7.8125
01001010
4A
1.0391
85
7.8125
00100100
24
0.7578
61
7.8125
01001011
4B
1.0469
86
7.8125
00100101
25
0.7656
62
7.8125
01001100
4C
1.0547
87
7.8125
00100110
26
0.7734
63
7.8125
01001101
4D
1.0625
88
7.8125
00100111
27
0.7813
64
7.8125
01001110
4E
1.0703
89
7.8125
00101000
28
0.7891
65
7.8125
01001111
4F
1.0781
8A
7.8125
00101001
29
0.7969
66
7.8125
01010000
50
1.0859
8B
7.8125
00101010
2A
0.8047
67
7.8125
01010001
51
1.0938
8C
7.8125
00101011
2B
0.8125
68
7.8125
01010010
52
1.1016
8D
7.8125
00101100
2C
0.8203
69
7.8125
01010011
53
1.1094
8E
7.8125
00101101
2D
0.8281
6A
7.8125
01010100
54
1.1172
8F
7.8125
00101110
2E
0.8359
6B
7.8125
01010101
55
1.1250
90
7.8125
00101111
2F
0.8438
6C
7.8125
01010110
56
1.1328
91
7.8125
00110000
30
0.8516
6D
7.8125
01010111
57
1.1406
92
7.8125
00110001
31
0.8594
6E
7.8125
01011000
58
1.1484
93
7.8125
00110010
32
0.8672
6F
7.8125
01011001
59
1.1563
94
7.8125
00110011
33
0.8750
70
7.8125
01011010
5A
1.1641
95
7.8125
00110100
34
0.8828
71
7.8125
01011011
5B
1.1719
96
7.8125
00110101
35
0.8906
72
7.8125
01011100
5C
1.1797
97
7.8125
00110110
36
0.8984
73
7.8125
01011101
5D
1.1875
98
7.8125
00110111
37
0.9063
74
7.8125
01011110
5E
1.1953
99
7.8125
00111000
38
0.9141
75
7.8125
01011111
5F
1.7969
E6
00111001
39
0.9219
76
7.8125
01100000
60
0.9531
7A
00111010
3A
0.9297
77
7.8125
01100001
61
1.2031
9A
7.8125
00111011
3B
0.9375
78
7.8125
01100010
62
1.2109
9B
7.8125
00111100
3C
0.9453
79
7.8125
01100011
63
1.2188
9C
7.8125
BINARY
CODE
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13
FN8705.1
March 7, 2016
ISL68200
TABLE 7. PROG1 8-BIT (BOOT-UP VOLTAGE) (Continued)
TABLE 7. PROG1 8-BIT (BOOT-UP VOLTAGE) (Continued)
HEX CODE
VBOOT
(V)
VOUT
COMMAND
CODE (HEX)
DELTA FROM
PREVIOUS
CODE (mV)
BINARY
CODE
HEX CODE
VBOOT
(V)
VOUT
COMMAND
CODE (HEX)
DELTA FROM
PREVIOUS
CODE (mV)
01100100
64
1.2266
9D
7.8125
10001011
8B
1.5156
C2
7.8125
01100101
65
1.2344
9E
7.8125
10001100
8C
1.5234
C3
7.8125
01100110
66
1.2422
9F
7.8125
10001101
8D
1.5313
C4
7.8125
01100111
67
1.2500
A0
7.8125
10001110
8E
1.5391
C5
7.8125
01101000
68
1.2578
A1
7.8125
10001111
8F
1.5469
C6
7.8125
01101001
69
1.2656
A2
7.8125
10010000
90
1.5547
C7
7.8125
01101010
6A
1.2734
A3
7.8125
10010001
91
1.5625
C8
7.8125
01101011
6B
1.2813
A4
7.8125
10010010
92
1.5703
C9
7.8125
01101100
6C
1.2891
A5
7.8125
10010011
93
1.5781
CA
7.8125
01101101
6D
1.2969
A6
7.8125
10010100
94
1.5859
CB
7.8125
01101110
6E
1.3047
A7
7.8125
10010101
95
1.5938
CC
7.8125
01101111
6F
1.3125
A8
7.8125
10010110
96
1.6016
CD
7.8125
01110000
70
1.3203
A9
7.8125
10010111
97
1.6094
CE
7.8125
01110001
71
1.3281
AA
7.8125
10011000
98
1.6172
CF
7.8125
01110010
72
1.3359
AB
7.8125
10011001
99
1.6250
D0
7.8125
01110011
73
1.3438
AC
7.8125
10011010
9A
1.6328
D1
7.8125
01110100
74
1.3516
AD
7.8125
10011011
9B
1.6406
D2
7.8125
01110101
75
1.3594
AE
7.8125
10011100
9C
1.6484
D3
7.8125
01110110
76
1.3672
AF
7.8125
10011101
9D
1.6563
D4
7.8125
01110111
77
1.3750
B0
7.8125
10011110
9E
1.6641
D5
7.8125
01111000
78
1.3828
B1
7.8125
10011111
9F
3.0000
180
01111001
79
1.3906
B2
7.8125
10100000
A0
1.0469
86
01111010
7A
1.3984
B3
7.8125
10100001
A1
1.6719
D6
7.8125
01111011
7B
1.4063
B4
7.8125
10100010
A2
1.6797
D7
7.8125
01111100
7C
1.4141
B5
7.8125
10100011
A3
1.6875
D8
7.8125
01111101
7D
1.4219
B6
7.8125
10100100
A4
1.6953
D9
7.8125
01111110
7E
1.4297
B7
7.8125
10100101
A5
1.7031
DA
7.8125
01111111
7F
2.5000
140
10100110
A6
1.7109
DB
7.8125
10000000
80
1.0000
80
10100111
A7
1.7188
DC
7.8125
10000001
81
1.4375
B8
7.8125
10101000
A8
1.7266
DD
7.8125
10000010
82
1.4453
B9
7.8125
10101001
A9
1.7344
DE
7.8125
10000011
83
1.4531
BA
7.8125
10101010
AA
1.7422
DF
7.8125
10000100
84
1.4609
BB
7.8125
10101011
AB
1.7500
E0
7.8125
10000101
85
1.4688
BC
7.8125
10101100
AC
1.7578
E1
7.8125
10000110
86
1.4766
BD
7.8125
10101101
AD
1.7656
E2
7.8125
10000111
87
1.4844
BE
7.8125
10101110
AE
1.7734
E3
7.8125
10001000
88
1.4922
BF
7.8125
10101111
AF
1.7813
E4
7.8125
10001001
89
1.5000
C0
7.8125
10110000
B0
1.7891
E5
7.8125
10001010
8A
1.5078
C1
7.8125
10110001
B1
1.7969
E6
7.8125
BINARY
CODE
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14
FN8705.1
March 7, 2016
ISL68200
TABLE 7. PROG1 8-BIT (BOOT-UP VOLTAGE) (Continued)
TABLE 7. PROG1 8-BIT (BOOT-UP VOLTAGE) (Continued)
HEX CODE
VBOOT
(V)
VOUT
COMMAND
CODE (HEX)
DELTA FROM
PREVIOUS
CODE (mV)
BINARY
CODE
HEX CODE
VBOOT
(V)
VOUT
COMMAND
CODE (HEX)
DELTA FROM
PREVIOUS
CODE (mV)
10110010
B2
1.8047
E7
7.8125
11011001
D9
3.4063
1B4
78.125
10110011
B3
1.8125
E8
7.8125
11011010
DA
3.4844
1BE
78.125
10110100
B4
1.8203
E9
7.8125
11011011
DB
3.5625
1C8
78.125
10110101
B5
1.8281
EA
7.8125
11011100
DC
3.6406
1D2
78.125
10110110
B6
1.8359
EB
7.8125
11011101
DD
3.7188
1DC
78.125
10110111
B7
1.9141
F5
78.125
11011110
DE
3.7969
1E6
78.125
10111000
B8
1.9922
FF
78.125
11011111
DF
5.0000
280
10111001
B9
2.0703
109
78.125
11100000
E0
1.2031
9A
10111010
BA
2.1484
113
78.125
11100001
E1
3.8750
1F0
78.125
10111011
BB
2.2266
11D
78.125
11100010
E2
3.9531
1FA
78.125
10111100
BC
2.3047
127
78.125
11100011
E3
4.0313
204
78.125
10111101
BD
2.3828
131
78.125
11100100
E4
4.1094
20E
78.125
10111110
BE
2.4609
13B
78.125
11100101
E5
4.1875
218
78.125
10111111
BF
3.2969
1A6
11100110
E6
4.2656
222
78.125
11000000
C0
1.1016
8D
11100111
E7
4.3438
22C
78.125
11000001
C1
2.4688
13C
7.8125
11101000
E8
4.4219
236
78.125
11000010
C2
2.4766
13D
7.8125
11101001
E9
4.5000
240
78.125
11000011
C3
2.4844
13E
7.8125
11101010
EA
4.5781
24A
78.125
11000100
C4
2.4922
13F
7.8125
11101011
EB
4.6563
254
78.125
11000101
C5
2.5000
140
7.8125
11101100
EC
4.7344
25E
78.125
11000110
C6
2.5078
141
7.8125
11101101
ED
4.8125
268
78.125
11000111
C7
2.5156
142
7.8125
11101110
EE
4.8906
272
78.125
11001000
C8
2.5234
143
7.8125
11101111
EF
4.9688
27C
78.125
11001001
C9
2.6016
14D
78.125
11110000
F0
4.9766
27D
7.8125
11001010
CA
2.6797
157
78.125
11110001
F1
4.9844
27E
7.8125
11001011
CB
2.7578
161
78.125
11110010
F2
4.9922
27F
7.8125
11001100
CC
2.8359
16B
78.125
11110011
F3
5.0000
280
7.8125
11001101
CD
2.9141
175
78.125
11110100
F4
5.0078
281
7.8125
11001110
CE
2.9922
17F
78.125
11110101
F5
5.0156
282
7.8125
11001111
CF
3.0703
189
78.125
11110110
F6
5.0234
283
7.8125
11010000
D0
3.1484
193
78.125
11110111
F7
5.0313
284
7.8125
11010001
D1
3.2266
19D
78.125
11111000
F8
5.1094
28E
78.125
11010010
D2
3.2813
1A4
54.6875
11111001
F9
5.1875
298
78.125
11010011
D3
3.2891
1A5
7.8125
11111010
FA
5.2656
2A2
78.125
11010100
D4
3.2969
1A6
7.8125
11111011
FB
5.3438
2AC
78.125
11010101
D5
3.3047
1A7
7.8125
11111100
FC
5.4219
2B6
78.125
11010110
D6
3.3125
1A8
7.8125
11111101
FD
5.4922
2BF
70.3125
11010111
D7
3.3203
1A9
7.8125
11111110
FE
5.5000
2C0
7.8125
11011000
D8
3.3281
1AA
7.8125
11111111
FF
0
0
BINARY
CODE
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15
FN8705.1
March 7, 2016
ISL68200
The inductor DCR value will increase as the temperature
increases. Therefore, the sensed current will increase as the
temperature of the current sense element increases. In order to
compensate the temperature effect on the sensed current signal,
the integrated temperature compensation function of ISL68200
should be utilized. The integrated temperature compensation
function is described in “Thermal Monitoring and Compensation”
on page 17.
VCC
-
VSEN
L
+
DRIVER
DCR
INDUCTOR
VL
+
ISL68200
INTERNAL CIRCUIT
RGND
+
VC(s)
IOUT
FIGURE 8. EXTERNAL PROGRAMMABLE REGULATION
R
Current Sensing
The ISL68200 supports inductor DCR sensing, or resistive
sensing techniques, and senses current continuously for fast
response. The current sense amplifier uses the CSEN and CSRTN
inputs to reproduce a signal proportional to the inductor current,
IL. The sense current, ISEN, is proportional to the inductor current
and is used for current reporting and overcurrent protection.
The input bias current of the current sensing amplifier is typically
10s of nA; less than 15kΩ input impedance connected to CSEN
pin is preferred to minimize the offset error, i.e., use a larger C
value (select 0.22µF to 1µF instead of 0.1µF when needed). In
addition, the current sensing gain resistor connected to CSRTN
pin should be within 40Ωto 3.5kΩ.
INDUCTOR DCR SENSING
An inductor’s winding is characteristic of a distributed resistance,
as measured by the DCR (Direct Current Resistance) parameter.
A simple R-C network across the inductor extracts the DCR
voltage, as shown in Figure 9.
CSRTN
-
DCR
I SEN = I ------------------LR
ISEN
FIGURE 9. DCR SENSING CONFIGURATION
RESISTIVE SENSING
For accurate current sense, a dedicated current-sense resistor
RSENSE, in series with each output inductor can serve as the current
sense element (see Figure 10). This technique, however, reduces
overall converter efficiency due to the additional power loss on the
current sense element RSENSE.
I
DCR
I SEN = I L  -----------------R ISEN
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(EQ. 6)
16
L
L
RSEN
ESL
RSENSE
VR
+
ISL68200
INTERNAL CIRCUIT
Equation 6 shows that the ratio of the inductor current to the
sensed current, ISEN, is driven by the value of the sense resistor
and the DCR of the inductor.
C
CSEN
+
(EQ. 5)
VOUT
COUT
PLACE THESE IN CLOSE
PROXIMITY TO ISL68200
R V (s)
C
OPTIONAL
IOUT
+
If the R-C network components are selected such that the RC
time constant (= R*C) matches the inductor time constant
(= L/DCR), the voltage across the capacitor VC is equal to the
voltage drop across the DCR. With the internal low-offset current
amplifier, the capacitor voltage VC is replicated across the sense
resistor RISEN. Therefore, the current out of the CSRTN pin, ISEN,
is proportional to the inductor current.
OPTIONAL
RISEN
CURRENT
SENSE
The voltage on the capacitor VC, can be shown to be proportional
to the inductor current IL , as in Equation 5.
L
 s  ------------+ 1   DCR  I L 
 DCR

V C  s  = -------------------------------------------------------------------- s  RC + 1 
COUT
PLACE THESE IN CLOSE
PROXIMITY TO ISL68200
B. VOUT LOWER THAN DAC
A. VOUT HIGHER THAN DAC
VOUT
-
VOUT
+
-
VSEN
I s
L
-
VOUT
VIN
-
As shown in Table 7, 1 step is 2-7 = 7.8125mV; some selections
are higher than 1 step from adjacent codes. However, the
resolution is ±7.8125mV around the popular voltage regulation
points, as in Table 3 on page 11, for fine tune purpose. For finer
than 7.8125mV tuning, a large ratio resistor divider can be
placed on the VSEN pin between the output (VOUT) and RGND for
positive offset or VCC for negative offset, as in Figure 8.
CURRENT
SENSE
CSEN
+
C
RISEN
CSRTN
R SEN
I SEN = I ----------------LR
ISEN
FIGURE 10. SENSE RESISTOR IN SERIES WITH INDUCTORS
FN8705.1
March 7, 2016
ISL68200
A current sensing resistor has a distributed parasitic inductance,
known as ESL (equivalent series inductance, typically less than
1nH) parameter. A simple R-C network across the current sense
resistor extracts the RSEN voltage, as shown in Figure 10 on
page 16.
The voltage on the capacitor VC, can be shown to be proportional
to the inductor current IL, see Equation 7.
ESL
 s  --------------- + 1   R SEN  I L 
 R

SEN
V C  s  = ------------------------------------------------------------------------ s  RC + 1 
(EQ. 7)
If the R-C network components are selected such that the RC
time constant matches the ESL-RSEN time constant
(R*C = ESL/RSEN), the voltage across the capacitor VC is equal
to the voltage drop across the RSEN, i.e., proportional to the
inductor current. As an example, a typical 1mΩ sense resistor
can use R = 348 and C = 820pF for the matching. Figures 11 and
12 show the sensed waveforms without and with matching RC
when using resistive sense.
on current sensing will not provide a fast OCP response and hurt
system reliability.
LOAD
VIOUT
FIGURE 13. DESIRED LOAD TRANSIENT RESPONSE WAVEFORMS
LOAD
VIOUT
FIGURE 14. LOAD TRANSIENT RESPONSE WHEN R-C TIME
CONSTANT IS TOO SMALL
LOAD
FIGURE 11. VOLTAGE ACROSS R WITHOUT RC
VIOUT
FIGURE 15. LOAD TRANSIENT RESPONSE WHEN R-C TIME
CONSTANT IS TOO LARGE
FIGURE 12. VOLTAGE ACROSS C WITH MATCHING RC
Equation 8 shows that the ratio of the inductor current to the
sensed current, ISEN, is driven by the value of the sense resistor
and the RISEN.
R SEN
I SEN = I L  -----------------R ISEN
(EQ. 8)
Thermal Monitoring and Compensation
L/DCR OR ESL/RSEN MATCHING
Figure 13 shows the expected load transient response waveforms
if L/DCR or ESL/RSEN is matching the R-C time constant. When
the load current has a square change, the IOUT pin voltage (VIOUT)
without a decoupling capacitor also has a square response.
However, there is always some PCB contact impedance of current
sensing components between the two current sensing points; it
hardly accounts into the L/DCR or ESL/RSEN matching calculation.
Fine tuning the matching is necessarily done at the board level to
improve overall transient performance and system reliability.
If the R-C timing constant is too large or too small, VC(s) will not
accurately represent real-time output current and will worsen the
overcurrent fault response. Figure 14 shows the IOUT pin
transient voltage response when the R-C timing constant is too
small. VIOUT will sag excessively upon load insertion and may
create a system failure or early overcurrent trip. Figure 15 shows
the transient response when the R-C timing constant is too large.
VIOUT is sluggish in reaching its final value. The excessive delay
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Note that the integrated thermal compensation applies to the DC
current, but not the AC current; therefore, the peak current seen
by the controller will increase as the temperature decreases and
can potentially trigger an OCP event. To overcome this issue, the
RC should be over-matching L/DCR at room temperature by
(-40°C +25°C) * 0.385%/°C = +25% for -40°C operation.
17
The block diagram of thermal monitoring function is shown in
Figure 16 on page 18. One NTC resistor should be placed close to
the respective power stage of the voltage regulator VR to sense
the operational temperature and pull-up resistors are needed to
form the voltage dividers for the NTC pin. As the temperature of
the power stage increases, the resistance of the NTC will reduce,
resulting in the reduced voltage at the NTC pin. Figure 18 on
page 18 shows the TM voltage over the temperature for a typical
design with a recommended 10kΩ NTC (P/N:
NCP15XH103J03RC from Murata,  = 3380) and 1.54kΩ resistor
RTM. It is recommended to use those resistors for the accurate
temperature compensation since the internal thermal digital
code is developed based upon these two components. If a
different value is used, the temperature coefficient must be close
to 3380 and RTM must be scaled accordingly. For instance, say
NTC = 20kΩ ( = 3380), then RTM should be
20kΩ/10kΩ*1.54kΩ = 3.08kΩ.
FN8705.1
March 7, 2016
ISL68200
to the NTC temperature of the practical design should be similar
to that in Figure 18.
VCC
THERMAL TRIP
+136°C/+122ºC
100
90
80
NTC
+
OTP
-
RNTC
ºC
VTM/VCC (%)
RTM
ISL68200
70
60
50
40
BETA~ 3380
30
20
FIGURE 16. BLOCK DIAGRAM OF THERMAL MONITORING AND
PROTECTION
The ISL68200 supports inductor DCR sensing, or resistive
sensing techniques. The inductor DCR has a positive temperature
coefficient, which is about +0.385%/°C. Since the voltage across
the inductor is sensed for the output current information, the
sensed current has the same positive temperature coefficient as
the inductor DCR. In order to obtain the correct current
information, the ISL68200 utilizes the voltage at the NTC pin and
“TCOMP” register to compensate the temperature impact on the
sensed current. The block diagram of this function is shown in
Figure 17.
0
20
40
60
80
100
120
140
TEMPERATURE (°C)
FIGURE 18. THE RATIO OF TM VOLTAGE TO NTC TEMPERATURE
WITH RECOMMENDED PART
Since the NTC attaches to the PCB, but not directly to the current
sensing component, it inherits high thermal impedance between
the NTC and the current sensing element. The “TCOMP” register
values can be utilized to correct the temperature difference
between NTC and the current sense component. As shown in
Figure 19, the NTC should be placed in proximity to the output
rail; DON’T place it close to the MOSFET side, which generates
much more heat.
VCC
ISL68200
RTM
NTC
o
c
NON-LINEAR
A/D
RNTC
PLACE NTC
CLOSE TO
INDUCTOR
D/A
CHANNEL
CURRENT
SENSE
CSSEN
CSRTN
OUTPUT
INDUCTOR
NTC
VOUT
POWER STAGE
IPH
ki
FIGURE 19. RECOMMENDED PLACEMENT OF NTC
IOUT MONITOR AND
OVERCURRENT
PROTECTION
A/D
TCOMP
FIGURE 17. BLOCK DIAGRAM OF INTEGRATED TEMPERATURE
COMPENSATION
When the NTC is placed close to the current sense component
(inductor), the temperature of the NTC will track the temperature
of the current sense component. Therefore, the NTC pin voltage
can be utilized to obtain the temperature of the current sense
component. Since the NTC could pick up noise from the phase
node, a 0.1µF ceramic decoupling capacitor is recommended on
the NTC pin in close proximity to the controller.
The ISL68200 multiplexes the “TCOMP” value with the NTC
digital signal to obtain the adjustment gain to compensate the
temperature impact on the sensed channel current. The
compensated current signal is used for IOUT and overcurrent
protection functions. The TCOMP “OFF” code is to disable thermal
compensation when the current sensing element is the resistor
or smart power stage (internally thermal compensated) that has
little thermal drifting.
TABLE 8. “TCOMP” VALUES
D1h
TCOMP (°C)
D1h
TCOMP (°C)
0h
30
2h
5
1h
15
3h
OFF
Based on the VCC voltage, the ISL68200 converts the NTC pin
voltage to a digital signal for temperature compensation. With
the nonlinear A/D converter of the ISL68200, the NTC digital
signal is linearly proportional to the NTC temperature. For
accurate temperature compensation, the ratio of the NTC voltage
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FN8705.1
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ISL68200
Thermal compensation design procedure for inductor current
sensing is summarized as follows:
1. Properly choose the voltage divider for the NTC pin to match
the NTC voltage vs temperature curve with the recommended
curve in Figure 18 on page 18.
63.875A load current. The IOUT voltage is linearly digitized every
108µs and stored in the READ_IOUT register (8Ch).
VCC
2. Run the actual board under the full load and the desired
airflow condition.
3. After the board reaches the thermal steady state (often takes
15 minutes), record the temperature (TCSC) of the current
sense component (inductor) and the voltage at NTC and VCC
pins.
4. Use Equation 9 to calculate the resistance of the NTC, and
find out the corresponding NTC temperature TNTC from the
NTC datasheet or using Equation 10, where is equal to 3380
for recommended NTC.
V TM xR
TM
R NTC  @T NTC  = ----------------------------V CC – V
(EQ. 9)
TM

T NTC = ------------------------------------------------------------------------------- – 273.15
 R NTC  @25 0 C  

ln  ------------------------------------------- + ----------------- R NTC  @T NTC  298.15
(EQ. 10)
5. Choose a number close to the result as in Equation 11 for the
“TCOMP” register.
T COMP = T CSC – T NTC
(EQ. 11)
6. Run the actual board under full load again.
7. Record the IOUT pin voltage as V1 immediately after the
output voltage is stable with the full load. Record the IOUT pin
voltage as V2 after the VR reaches the thermal steady state.
8. If the IOUT pin voltage increases over 10mV as the
temperature increases, i.e., V2 - V1 > 10mV, reduce “TCOMP”
value. If the IOUT pin voltage decreases over 10mV as the
temperature increases, i.e., V1 - V2 > 10mV, increase
“TCOMP” value. “TCOMP” value can be adjusted via the series
bus for easy thermal compensation optimization.
IOUT Calibration
The current flowing out of the IOUT pin is equal to the sensed
average current inside ISL68200. A resistor is placed from the
IOUT pin to GND to generate a voltage, which is proportional to
the load current and the resistor value, as shown in Equation 12:
R x xI OCP
2.5Vx  -------------------------
2.5VxR ISEN
 100A 
R IOUT = ---------------------------------- = ----------------------------------------------63.875AxR x
63.875AxR x
EXTERNAL CIRCUIT
ISL68200
RIOUT_UP
DIGITIZED
IOUT (8Ch)
IOUT
RIOUT_DW
FIGURE 20. IOUT NO LOAD OFFSET CALIBRATION
A small capacitor can be placed between IOUT and GND to
reduce the noise impact and provide averaging, > 200µs
(typically).
To deal with layout and design variation of different platforms,
ISL68200 is intentionally trimmed to negative at no load, thus,
an offset can easily be added to calibrate the digitized IOUT
reading (8Ch). Hence, the analog vs digitized current slope is set
by the equivalent impedance of RIOUT_UP//RIOUT_DW =RIOUT
(as in Figure 20); the slope of the ideal curve should set to 1 A/A
with 0A offset.
For a precision digital IOUT, follow the fine-tune procedure below
step-by-step; steps 1 to 5 must be completed before step 6.
1. Properly tune L/DCR or ESL/RSEN matching as shown on
page 17 over the range of temperature operation. +25% overmatching L/DCR at room temperature is needed for -40°C
operation.
2. Properly complete thermal compensation as shown on
“Thermal Monitoring and Compensation” on page 17.
3. Finalize RISEN resistor to set OCP for overall operating
conditions and board variations as shown in “Overcurrent and
Short-Circuit Protection” on page 20.
4. Collect no load IOUT current with sufficient prototypes and
determine the mean of no load IOUT current.
5. The pull-up impedance on IOUT pin should be
“VCC/IOUT_NO_LOAD”; for instance, a mean of -2.5µA IOUT at
0A load, it will need RIOUT_UP = 2MΩ.
6. Start with the value below and then fine tune the RIOUT_DW
value until the average slope of various boards equals 1A/A.
(EQ. 12)
2.5VxI OCP
25VxI OCP
= --------------------------------------------- = ----------------------------- k
63.875Ax100A
63.875A
R IOUT_UP xR IOUT
R IOUT_DW = -------------------------------------------------R IOUT_UP – R IOUT
(EQ. 13)
Where VIOUT is the voltage at the IOUT pin, RIOUT is the resistor
between the IOUT pin and GND, ILOAD is the total output current
of the converter, RISEN is the sense resistor connected to the
CSRTN pin and RX is the DC resistance of the current sense
element, either the DCR of the inductor or RSENSE depending on
the sensing method. The RIOUT resistor should be scaled to
ensure that the voltage at the IOUT pin is typically 2.5V at
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ISL68200
Fault Protection
The ISL68200 provides high system reliability with many fault
protections, as summarized in Table 9.
TABLE 9. FAULT PROTECTION SUMMARY
FAULT
Input UVLO
DESCRIPTION
FAULT ACTION
VIN pin UVLO; or set by EN pin with Shutdown and recover
when VIN > UVLO
an external divider for higher
level. See Figures 4 and 5.
Bias UVLO
VCC, PVCC, 7VLDO UVLO
Start-Up
OVP
Higher than VBOOT. See Electrical Latch OFF, reset by VCC
Specifications on page 7.
or toggling Enable
(including EN pin and/
Rising = 116%; Falling = 100%
or OPERATION
74% of VOUT, Latch OFF
command based upon
ON_OFF_CONFIG
setting)
Output OVP
Output UVP
Output OCP Average OCP = 100µA with
128µs blanking time.
Short-Circuit Peak OCP = 130% of Average
Protection
OCP with 50ns filter.
OTP
Rising = 22.31%VCC (~+136°C);
Falling =27.79%VCC (~+122°C).
Shutdown and recover
when Bias > UVLO
Latch OFF (reset by
VCC or toggling enable
including EN pin and/
or OPERATION
command based upon
ON_OFF_CONFIG
setting), or retry every
9ms; option is
programmable by
PROG3 or D3[0]
Shut down above
+136°C and recover
when temperature
drops below +122°C
Input UVLO and OTP faults will respond to the current state with
hysteresis, while output OVP and output UVP faults are latch
events, while output OCP and output short-circuit faults can be
latch or retry events depending upon PROG3 or D3[0] setting. All
fault latch events can be reset by VCC cycling, toggling the Enable
pin and/or series bus OPERATION command based upon
ON_OFF_CONFIG setting, while the OCP retry event has a hiccup
time of 9ms and the regulator can be recovered when the fault is
removed.
OVERVOLTAGE PROTECTION
The OVP fault detection circuit triggers after the voltage between
VSEN+ and VSEN- is above the rising overvoltage threshold. When
an OVP fault is declared, the controller will be latched off and the
PGOOD pin will be asserted low. The fault will remain latched and
can be reset by VCC cycling or toggling EN pin and/or series bus
OPERATION command based upon ON_OFF_CONFIG setting.
Although the controller has latched-off in response to an OVP
fault, the LGATE gate-driver output will retain the ability to toggle
the low-side MOSFET on and off, in response to the output
voltage transversing the OVP rising and falling thresholds. The
LGATE gate-driver will turn on the low-side MOSFET to discharge
the output voltage, protecting the load. The LGATE gate driver will
turn off the low-side MOSFET once the sensed output voltage is
lower than the falling overvoltage threshold (typically 100%). If
the output voltage rises again, the LGATE driver will again turn on
the low-side MOSFET when the output voltage is above the rising
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overvoltage threshold (typically 120%). By doing so, the IC protects
the load when there is a consistent overvoltage condition.
In addition to normal operation OVP, 5.5ms (typically, worst
6.5ms) after all rails (VCC, PVCC, 7VLDO, VIN) POR and prior to
the end of soft-start, the start-up OVP circuits are enabled to
protect against OVP event, while the OVP level is set higher than
VBOOT. See Electrical Specifications on page 7.
UNDERVOLTAGE PROTECTION
The UVP fault detection circuit triggers after the output voltage is
below the undervoltage threshold (typically 74% of DAC). When an
UVP fault is declared, the controller will be latched off, forcing the
LGATE and UGATE gate-driver outputs low, and the PGOOD pin will
be asserted low. The fault will remain latched and can be reset by
VCC cycling or toggling EN pin and/or series bus OPERATION
command based upon ON_OFF_CONFIG setting.
OVERCURRENT AND SHORT-CIRCUIT PROTECTION
The average Overcurrent Protection (OCP) is triggered when the
internal current out of the IOUT pin goes above the fault
threshold (typically 100µA) with 128µs blanking time. It also has
a fast (50ns filter) secondary overcurrent protection whose
threshold is +30% above average OCP; this protects inductor
saturation from a short-circuit event and provides a more robust
power train and system protection. When an OCP or short-circuit
fault is declared, the controller will be latched off, forcing the
LGATE and UGATE gate-driver outputs low, or retry with a hiccup
time of 9ms; the fault response is programmable by PROG3 or
D3[0]. The latched off event however can be reset by VCC cycling
or toggling EN pin and/or series bus OPERATION command
based upon ON_OFF_CONFIG setting.
Equation 14 provides a starting point to set a preliminary OCP trip
point, where IOCP is the targeted OCP trip point and I (as in
Equation 15 on page 28) is the peak-to-peak inductor ripple
current.
R x xI OCP
R ISEN1 = -----------------------100A
I
R x x  ----- + I OCP
2

R ISEN2 = ------------------------------------------------------------100Ax  100% + 30% 
(EQ. 14)
R ISEN = MAX (RISEN1, R ISEN2 
To deal with layout and PCB contact impedance variation, follow
the fine tune procedure below step-by-step for a more precision
OCP; steps 1 to 3 must be completed before step 4.
1. Properly tune L/DCR or ESL/RSEN matching as shown on
page 17 over the range of temperature operation. +25%
over-matching L/DCR at room temperature is needed for
-40°C operation.
2. Properly complete thermal compensation as shown on
“Thermal Monitoring and Compensation” on page 17.
3. Collect OCP trip points (IOCP_MEASURED) with sufficient
prototypes and determine the means for overall operating
conditions and board variations.
4. Change RISEN by IOCP_TARGETED/IOCP_MEASURED
percentage to meet the targeted OCP.
FN8705.1
March 7, 2016
ISL68200
Note that if the inductor peak-to-peak current is higher or closer
to 30%, the +30% threshold could be triggered instead of the
average OCP threshold. However, the fine tune procedure still can
be used.
OVER-TEMPERATURE PROTECTION
As shown in Figure 16, there is a comparator with hysteresis to
compare the NTC pin voltage to the threshold set. When the NTC
pin voltage is lower than 22.31% of VCC voltage (typically
+136°C), it triggers Over-Temperature Protection (OTP) and shuts
down ISL68200 operation, when the NTC pin voltage is above
27.79% of VCC voltage (typically +122.4°C), it will resume
normal operation. When an OTP fault is declared, the controller
will force the LGATE and UGATE gate-driver outputs low.
PGOOD Monitor
The PGOOD pin indicates when the converter is capable of
supplying regulated voltage. If there is a fault condition of a rail’s
(VCC, PVCC, 7VLDO, or VIN) UVLO, output Overcurrent (OCP),
Overvoltage (OVP), Undervoltage (UVP), or Over-Temperature (OTP),
PGOOD is asserted low. Note that the PGOOD pin is an undefined
impedance with insufficient VCC (typically <2.5V).
Adaptive Shoot-Through Protection
The LGATE and UGATE pins are MOSFET driver outputs. The
LGATE pin drives the low-side MOSFET of the converter while the
UGATE pin drives the high-side MOSFET of the converter. Adaptive
shoot-through protection prevents a gate-driver output from
turning on until the opposite gate-driver output has fallen below
approximately 1V. The dead time shown in Figure 21 is extended
by the additional period that the falling gate voltage remains
above the 1V threshold. The high-side gate-driver output voltage
is measured across the UGATE and PHASE pins while the low-side
gate-driver output voltage is measured across the LGATE and
GND pins.
PFM Mode Operation
In PFM mode, programmable by PROG2 or series bus D0[0:0],
the switching frequency is dramatically reduced to minimize the
switching loss and significantly improve light-load efficiency. The
ISL68200 can enter and exit PFM mode seamlessly as load
changes. For high VOUT applications implemented with high Qg
MOSFETs, the LGATE might not turn on long enough to charge the
boot capacitor in PFM mode with 0A load. It is recommended to
enable ISL68200’s ultrasonic PFM feature (by PROG3 or series
bus D2[0:0]), which maintains LGATE switching frequency above
20kHz and keeps the boot capacitor charged for immediate load
apply event. Alternatively, an external Schottky diode or
maintaining a minimum load can enhance the boot capacitor
charge.
SMBus, PMBus and I2C Operation
The ISL68200 features SMBus, PMBus and I2C with 32
programmable addresses via PROG2 pin, while SMBus/PMBus
includes an Alert# line (SALERT) and Packet Error Check (PEC) to
ensure data properly transmitted. The telemetry update rate is
108µs (Typically). The supported SMBus/PMBus/I2C addresses
are summarized in Table 10. The 7-bit format address does not
include the last bit (write and read): 40-47h, 60-67h and 70-7Fh.
SMBus/PMBus/I2C allows to program the registers as in
Table 11, except for SMBus/PMBus/I2C addresses, 5.5ms
(typically, worst 6.6ms) after all rails (VCC, PVCC, 7VLDO and VIN)
above POR. Figures 22 and 23 on page 22 show the initialization
timing diagram for the series bus with different state of EN
(enable) pin.
For proper operation, users should follow the SMBus, PMBus and
I2C protocol, as shown Figure 24 on page 23. Note that STOP (P)
bit is NOT allowed before the repeated START condition when
“reading” contents of register.
When the device’s series bus is not used, simply ground the
device’s SCL, SDA and SALERT pins and do not connect them to
the bus.
TABLE 10. SMBus/PMBus/I2C 7-BIT FORMAT ADDRESS (HEX)
UGATE-PHASE
1V
1V
tUGFLGR
tLGFUGR
1V
1V
LGATE-GND
FIGURE 21. GATE DRIVE ADAPTIVE SHOOT-THROUGH PROTECTION
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7-BIT ADDRESS
7-BIT ADDRESS
7-BIT ADDRESS
40
63
76
41
64
77
42
65
78
43
66
79
44
67
7A
45
70
7B
46
71
7C
47
72
7D
60
73
7E
61
74
7F
62
75
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March 7, 2016
ISL68200
VIN, PVCC,
7VLDO,
VCC
VCC POR
TIM EOU T
READER
DONE
W RITE AND READ
CONFIG URATION
5m s
0.5m s
0m s TO INFINITY
200µs
SS DELAY
W RITE AND READ
CONFIG URATION
W RITE AND READ
CONFIG URATIO N
W RITE AND READ
CONFIG URATIO N
0m s TO INFINITY
ENABLE
PM Bus COM M UNICATION
NOT ACTIVATED
PM Bus
COM M AND
PM Bus
COM M AN D
PM Bus
COM M AND
PM Bus
COM M AND
V BOOT
0V
V OUT
FIGURE 22. SIMPLIFIED SMBus/PMBus/I2C INITIALIZATION TIMING DIAGRAM WITH ENABLE LOW
VIN, PVCC,
7VLDO,
VCC
V CC POR
TIMEOUT
5ms
200 µs
SS DELAY
READER
DONE
0.5m s
W RITE AND READ
CONFIGURATION
W RITE AND READ
CONFIGURATION
W RITE AND READ
CONFIGURATION
0ms TO INFINITY
ENABLE
PMBus COM MUNICATION
NOT ACTIVATED
PMBus
COM MAND
PMBus
COMMAND
PMBus
COMMAND
V BOOT
V OUT
0V
FIGURE 23. SIMPLIFIED SMBus/PMBus/I2C INITIALIZATION TIMING DIAGRAM WITH ENABLE HIGH
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ISL68200
S: Start Condition
1. Send Byte Protocol
A: Acknowledge (“0”)
1
S
7+1
Slave Address_0
1
8
1
8
1
1
A
Command Code
A
PEC
A
P
N: Not Acknowledge (“1”)
W: Write (“0”)
RS: Repeated Start Condition
R: Read (“1”)
Optional 9 Bits for SMBus/PMBus
NOT used in I2C
PEC: Packet Error Checking
P: Stop Condition
Example command: 03h Clear Faults
(This will clear all of the bits in Status Byte for the selected Rail)
Acknowledge or DATA from Slave,
ISL68200
Not Used for One Byte Word
2. Write Byte/Word Protocol
1
S
7+1
Slave Address_0
1
8
1
8
1
8
1
8
1
1
A
Command Code
A
Low Data Byte
A
High Data Byte
A
PEC
A
P
Optional 9 Bits for SMBus/PMBus
NOT used in I2C
Example command: D0h ENABLE_PFM (one word, High Data Byte and ACK are not used)
3. Read Byte/Word Protocol
1
7+1
S
Slave Address_0
1
8
1
A
Command Code
A
1
7+1
RS
Slave Address_1
Not Used for One Byte Word Read
1
8
1
8
1
8
1
1
A
Low Data Byte
A
High Data Byte
A
PEC
N
P
Optional 9 Bits for SMBus/PMBus
NOT used in I2C
Example command: 8B READ_VOUT (Two words, read voltage of the selected rail).
NOTE: That all Writable commands are read with one byte word protocol.
STOP (P) bit is NOT allowed before the repeated START condition when “reading” contents of a register.
4. Block Write Protocol
1
S
7+1
Slave Address_0
1
8
1
8
1
A
Command Code
A
Byte Count = N
A
8
Lowest Data Byte
1
8
1
8
1
1
A
Data Byte N
A
PEC
A
P
1
A
8
Data Byte 2
1
A
Optional 9 Bits for SMBus/PMBus
NOT used in I2C
Example command: ADh IC_DEVICE_ID (2 Data Byte)
FIGURE 24. SMBus/PMBus/I2C COMMAND PROTOCOL
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March 7, 2016
ISL68200
5. Block Read Protocol
1
7+1
1
8
1
1
7+1
1
8
1
8
S
Slave Address_0
A
Command Code
A
RS
Slave Address_1
A
Byte Count = N
A
Lowest Data Byte
1
8
1
8
1
8
1
1
A
Data Byte 2
A
Data Byte N
A
PEC
N
P
Optional 9 Bits for SMBus/PMBus
NOT used in I2C
Example command: 8B READ_VOUT (Two words, read voltage of the selected rail).
NOTE: That all Writable commands are read with one byte word protocol.
STOP (P) bit is NOT allowed before the repeated START condition when “reading” contents of a register.
6. Group Command Protocol - No more than one command can be sent to the same Address
1
7+1
1
8
1
8
1
8
1
8
1
S
Slave ADDR1_0
A
Command Code
A
Low Data Byte
A
High Data Byte
A
PEC
A
1
7+1
1
8
1
8
1
8
1
RS
Slave ADDR2_0
A
Command Code
A
Data Byte
A
PEC
A
1
7+1
1
8
1
8
1
8
1
8
1
1
RS
Slave ADDR3_0
A
Command Code
A
Low Data Byte
A
High Data Byte
A
PEC
A
P
Optional 9 Bits for SMBus/PMBus
NOT used in I2C
FIGURE 25. SMBus/PMBus/I2C COMMAND PROTOCOL
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ISL68200
TABLE 11. SMBus, PMBus, AND I2C SUPPORTED COMMANDS
ACCESS
WORD
LENGTH
(BYTE)
DEFAULT
VALUE
COMMAND NAME
01h[7:0]
R/W
ONE
80h
OPERATION
02h[7:0]
R/W
ONE
1Fh
ON_OFF_CONFIG
03h
SEND BYTE
N/A
20h[7:0]
R
ONE
19h
VOUT_MODE
21h[2:0]
R/W
TWO
PROG1[7:0]
VOUT_COMMAND
24h[15:0]
R/W
TWO
VBOOT+500mV
VOUT_MAX
33h[15:0]
R/W
TWO
PROG3[5:3]
78h[8:0]
R
ONE
STATUS_BYTE
88h[15:0]
R
TWO
READ_VIN
8Bh[15:0]
R
TWO
READ_VOUT
VR Output Voltage, Resolution = 7.8125mV = 2-7
VOUT (V) = HEX2DEC(8B hex data) * 2-7
8Ch[15:0]
R
TWO
READ_IOUT
VR Output Current (N = -3, IMAX = 63.875A)
IOUT (A) = HEX2DEC(8C hex data-E800) * 0.125A when IOUT pin
voltage = 2.5V at 63.875A load.
COMMAND
CODE
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CLEAR_FAULTS
DESCRIPTION
VR Enable (depending upon ON_OFF_CONFIG configuration):
Bit[7]: 0 = OFF (0-F); 1 = ON (80-8Fh)
Bit[6:4] = 0
Bit[3:0] = Don’t care
Configure VR Enabled by OPERATION and/or EN pin:
Bit[7:5] = 0
Bit[4] = 1
Bit[3] = OPERATION command Enable
0h = OPERATION command has no control on VR
1h = OPERATION command can turn ON/OFF VR
Bit[2] = CONTROL pin Enable
0h = EN Pin has no control on VR
1h = EN pin can turn ON/OFF VR
Bit[1] = 1
Bit[0] = 1
Bit[3:2] = 00b = 13h (ALWAYS ON)
Bit[3:2] = 01b = 17h (EN controls VR)
Bit[3:2] = 10b = 1Bh (OPERATION controls VR)
Bit[3:2] = 11b = 1Fh (EN and OPERATION control VR)
Clear faults in status registers
Set host format of VOUT command.
Always Linear Format: N = -7
Set output voltage
HEX Code = DEC2HEX [ROUND(VOUT/2-7)]
Set maximum output voltage that VR can command
(DAC ≤ VOUT_MAX). Linear Format. N = -7
HEX Code = DEC2HEX(ROUNDUP(VOUT_MAX/ 2-7)
FREQUENCY_SWITCH Set VR Switching Frequency (In Linear Format)
Support 8 options (N = 0):
12Ch = 300kHz; 190h = 400kHz; 1F4h = 500kHz
258h = 600kHz; 2BCh = 700kHz; 352h = 850kHz
3E8h = 1MHz; 5DCh = 1.5MHz*
* Very high frequency is not recommended for very high duty cycle
applications as the boot capacitor will not has enough time to be
charged due to low LGATE ON time.
Fault Reporting;
Bit7 = Busy
Bit6 = OFF (Reflect current state of operation and ON_OFF_CONFIG
registers as well as VR Operation)
Bit5 = OVP
Bit4 = OCP
Bit3 = 0
Bit2 = OTP
Bit1 = Bus communication error
Bit0 = NONE OF ABOVE (OUTPUT UVP, VOUT_COMAND >
VOUT_MAX, or VOUT OPEN SENSE)
Input Voltage (N = - 4, Max = 31.9375V)
VIN (V) = HEX2DEC(88 hex data - E000h) * 0.0625V
FN8705.1
March 7, 2016
ISL68200
TABLE 11. SMBus, PMBus, AND I2C SUPPORTED COMMANDS (Continued)
COMMAND
CODE
ACCESS
WORD
LENGTH
(BYTE)
8Dh[15:0]
R
TWO
98h[7:0]
R
ONE
02h
PMBUS_REVISION
AD[15:0]
BLOCK R
TWO
8200h
IC_DEVICE_ID
AE[15:0]
BLOCK R
TWO
0003h
D0[0:0]
R/W
ONE
PROG2[7:7]
ENABLE_PFM
PFM OPERATION
0h = PFM Enabled (DCM at light load)
1h = PFM Disabled (always CCM mode)
D1[1:0]
R/W
ONE
PROG2[6:5]
TEMP_COMP
Thermal Compensation:
0h = 30°C; 01h = 15°C; 02h = 5°C; 03h = OFF
D2[0:0]
R/W
ONE
PROG3[7:7]
D3[0:0]
R/W
ONE
PROG3[6:6]
OCP_BEHAVIOR
D4[2:0]
R/W
ONE
PROG3[2:0]
AV_GAIN
D5{2:0]
R/W
ONE
PROG4[7:5]
RAMP_RATE
D6[1:0]
R/W
ONE
PROG4[4:3]
SET_RR
DC[7:0]
R
ONE
READ_PROG1
Read PROG1
DD{7:0]
R
ONE
READ_PROG2
Read PROG2
DE[7:0]
R
ONE
READ_PROG3
Read PROG3
DF[7:0]
R
ONE
READ_PROG4
Read PROG4
DEFAULT
VALUE
COMMAND NAME
READ_TEMP
DESCRIPTION
VR Temperature
TEMP (°C) = 1/{ln[Rup*HEX2DEC(8D hex
data)/(511 - HEX2DEC(8D hex data)/RNTC(at +25°C)]/Beta +
1/298.15} -273.15
Indicates PMBus Revision 1.2
ISL68200 Device ID
IC_DEVICE_REVISION ISL68200 Device Revision
ENABLE_ULTRASONIC Ultrasonic PFM Enable
0h = 25kHz Clamp Disabled
1h = 25kHz Clamp Enabled
Set latch or infinite retry for OCP fault:
0h = Retry every 9ms; 01 = Latch-OFF
R4 AV GAIN (PROG4, AV Gain Multiplier = 2x)
0h = 84; 1h = 73; 2h = 61; 3h = 49
4h = 38; 5h = 26; 6h = 14; 7h = 2
R4 AV GAIN (PROG4, AV Gain Multiplier = 1x)
0h = 42; 1h = 36.5; 2h = 30.5; 3h = 29.5
4h = 19; 5h = 13; 6h = 7; 7h = 1
Soft-Start and Margining DVID Rate (mV/µs)
0h = 1.25; 1h = 2.5; 2h = 5; 3h = 10; 4h = 0.078; 5h = 0.157
6h = 0.315; 7h = 0.625;
Set RR
0h = 200k; 01h = 400k; 02h = 600k; 03h = 800k
NOTE: Series bus communication is valid 5.5m (typically, worst 6.5ms) after VCC, VIN, 7VLDO and PVCC above POR. The telemetry update rate is 108µs.
R4™ Modulator
STABILITY
The R4™ modulator is an evolutionary step in R3™ technology.
Like R3™, the R4™ modulator is a linear control loop and
variable frequency control during load transients to eliminate
beat frequency oscillation at the switching frequency and
maintains the benefits of current-mode hysteretic controllers.
However, in addition, the R4™ modulator reduces regulator
output impedance and uses accurate referencing to eliminate
the need for a high-gain voltage amplifier in the compensation
loop. The result is a topology that can be tuned to voltage-mode
hysteretic transient speed while maintaining a linear control
model and removes the need for any compensation. This greatly
simplifies the regulator design for customers and reduces
external component cost.
The removal of compensation derives from the R4™ modulator’s
lack of need for high DC gain. In traditional architectures, high DC
gain is achieved with an integrator in the voltage loop. The
integrator introduces a pole in the open-loop transfer function at
low frequencies. That, combined with the double-pole from the
output L/C filter, creates a three pole system that must be
compensated to maintain stability.
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Classic control theory requires a single-pole transition through
unity gain to ensure a stable system. Current-mode architectures
(includes peak, peak-valley, current-mode hysteric, R3™ and
R4™) generate a zero at or near the L/C resonant point,
effectively canceling one of the system’s poles. The system still
contains two poles, one of which must be canceled with a zero
before unity gain crossover to achieve stability.
FN8705.1
March 7, 2016
ISL68200
COMPENSATION TO COUNTER
INTEGRATOR POLE
INTEGRATOR
FOR HIGH DC GAIN
Figure 28 shows the R4™ error-amplifier that does not require an
integrator for high DC gain to achieve accurate regulation. The
result to the open-loop response can be seen in Figure 29.
R4™ LOOP GAIN (dB)
VOUT
L/C DOUBLE-POLE
VCOMP
VDAC
p1
FIGURE 26. CLASSICAL INTEGRATOR ERROR-AMPLIFIER
CONFIGURATION
f (Hz)
FIGURE 29. UNCOMPENSATED R4™ OPEN-LOOP RESPONSE
TRANSIENT RESPONSE
In addition to requiring a compensation zero, the integrator in
traditional architectures also slows system response to transient
conditions. The change in COMP voltage is slow in response to a
rapid change in output voltage. If the integrating capacitor is
removed, COMP moves as quickly as VOUT, and the modulator
immediately increases or decreases switching frequency to
recover the output voltage.
R3™ LOOP GAIN (dB)
IOUT
INTEGRATOR POLE
p1
NO COMPENSATOR IS
NEEDED
ec
/d
B
0d dec
-2
/
B
c
0d
/ de
dB
-40
Because R4™ does not require a high-gain voltage loop, the
integrator can be removed, reducing the number of inherent
poles in the loop to two. The current-mode zero continues to
cancel one of the poles, ensuring a single-pole crossover for a
wide range of output filter choices. The result is a stable system
with no need for compensation components or complex
equations to properly tune the stability.
CURRENT-MODE
ZERO
z1
-2
Figure 26 illustrates the classic integrator configuration for a
voltage loop error amplifier. While the integrator provides the
high DC gain required for accurate regulation in traditional
technologies, it also introduces a low-frequency pole into the
control loop. Figure 27 shows the open-loop response that results
from the addition of an integrating capacitor in the voltage loop.
The compensation components found in Figure 26 are necessary
to achieve stability.
SYSTEM HAS 2 POLES
AND 1 ZERO
p2
t
R4™
L/C DOUBLE-POLE
R3™
VCOMP
p2
-20dB CROSSOVER
REQUIRED FOR STABILITY
p3
VOUT
COMPENSATOR TO
ADD z2 IS NEEDED
CURRENT-MODE
ZERO
z1
FIGURE 30. R3™ vs R4™ IDEALIZED TRANSIENT RESPONSE
ec
c
ec
/de
/d
dB
dB
0
-2
-40
-60dB/d
t
f (Hz)
FIGURE 27. UNCOMPENSATED INTEGRATOR OPEN-LOOP RESPONSE
R2
VOUT
VCOMP
R1
VDAC
FIGURE 28. NON-INTEGRATED R4™ ERROR-AMPLIFIER
CONFIGURATION
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27
The dotted red and blue lines in Figure 30 represent the time
delayed behavior of VOUT and VCOMP in response to a load
transient when an integrator is used. The solid red and blue lines
illustrate the increased response of R4™ in the absence of the
integrator capacitor.
To optimize transient response and improve phase margin for
very wide range applications, ISL68200 integrates a couple of
selectable AV and RR options that move DC gain and z1 point, as
shown in Figure 27. The defaulted AV gain of 42 and RR of
200kΩ however, can cover many cases and provides sufficient
gain and phase margin. For some extreme cases, lower AV gain
and bigger RR values are needed to provide a better phase
margin and improve transient ringback. The optimal choice AV
and RR can be obtained, by simple monitoring transient
response when playing with AV and RR values via the series bus.
FN8705.1
March 7, 2016
ISL68200
General Application Design
Guide
Thus, once the output capacitors are selected, the maximum
allowable ripple voltage, VP-P(MAX), determines the lower limit on
the inductance, as shown in Equation 16.
This design guide is intended to provide a high-level explanation of
the steps necessary to design a single-phase buck converter. It is
assumed that the reader is familiar with many of the basic skills
and techniques referenced in the following. In addition to this
guide, Intersil provides complete reference designs that include
schematics, bills of materials and example board layouts.
Output Filter Design
The output inductors and the output capacitor bank together to
form a low-pass filter responsible for smoothing the pulsating
voltage at the phase nodes. The output filter also must provide
the transient energy until the regulator can respond. Because it
has a low bandwidth compared to the switching frequency, the
output filter necessarily limits the system transient response. The
output capacitor must supply or sink load current while the
current in the output inductors increases or decreases to meet
the demand.
In high-speed converters, the output capacitor bank is usually the
most costly (and often the largest) part of the circuit. Output filter
design begins with minimizing the cost of this part of the circuit.
The critical load parameters in choosing the output capacitors are
the maximum size of the load step, I; the load current slew rate,
di/dt; and the maximum allowable output voltage deviation under
transient loading, VMAX. Capacitors are characterized according
to their capacitance, ESR and ESL (equivalent series inductance).
At the beginning of the load transient, the output capacitors
supply all of the transient current. The output voltage will initially
deviate by an amount approximated by the voltage drop across
the ESL. As the load current increases, the voltage drop across
the ESR increases linearly until the load current reaches its final
value. The capacitors selected must have sufficiently low ESL and
ESR so that the total output voltage deviation is less than the
allowable maximum. Neglecting the contribution of inductor
current and regulator response, the output voltage initially
deviates by an amount, as shown in Equation 15:
I
ESL
1
V  I  ESR + ----------------  V IN + -----------------  ----------------------------L OUT
C OUT 8  N  f
(EQ. 15)
SW
V OUT   1 – D 
I = ---------------------------------------L OUT  f
SW
The filter capacitor must have sufficiently low ESL and ESR so
that V < VMAX.
Most capacitor solutions rely on a mixture of high-frequency
capacitors with relatively low capacitance in combination with bulk
capacitors having high capacitance but limited high-frequency
performance. Minimizing the ESL of the high-frequency capacitors
allows them to support the output voltage as the current increases.
Minimizing the ESR of the bulk capacitors allows them to supply the
increased current with less output voltage deviation. The ESR of the
bulk capacitors also creates the majority of the output voltage
ripple. As the bulk capacitors sink and source the inductor AC
ripple current, a voltage develops across the bulk-capacitor ESR
equal to IC(P-P) (ESR).
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V OUT   V IN – V OUT 
L OUT  ESR  -------------------------------------------------------------f SW  V IN  V P – P MAX 
(EQ. 16)
Since the capacitors are supplying a decreasing portion of the
load current while the regulator recovers from the transient, the
capacitor voltage becomes slightly depleted. The output
inductors must be capable of assuming the entire load current
before the output voltage decreases more than VMAX. This
places an upper limit on inductance.
Equation 17 gives the upper limit on L for cases when the trailing
edge of the current transient causes a greater output-to-voltage
deviation than the leading edge. Equation 18 addresses the
leading edge. Normally, the trailing edge dictates the selection of
L because duty cycles are usually less than 50%. Nevertheless,
both inequalities should be evaluated, and L should be selected
based on the lower of the two results. In each equation, L is the
per-channel inductance, C is the total output capacitance.
2  C  V OUT
L OUT  ---------------------------------- V MAX – I  ESR
 I  2
1.25  C- V


L OUT  -------------------MAX – I  ESR  V IN – V OUT
 I  2
(EQ. 17)
(EQ. 18)
Input Capacitor Selection
The input capacitors are responsible for sourcing the AC
component of the input current flowing into the upper MOSFETs.
Their RMS current capacity must be sufficient to handle the AC
component of the current drawn by the upper MOSFETs, which is
related to duty cycle and the number of active phases. The input
RMS current can be calculated with Equation 19.
I IN RMS =
D
 D – D 2   Io 2 + ------   I  2
12
(EQ. 19)
Use Figure 31 to determine the input capacitor RMS current
requirement given the duty cycle, maximum sustained output
current (IO), and the ratio of the per-phase peak-to-peak inductor
current (IL(P-P) to IO. Select a bulk capacitor with a ripple current
rating, which will minimize the total number of input capacitors
required to support the RMS current calculated. The voltage rating of
the capacitors should also be at least 1.25x greater than the
maximum input voltage.
Low capacitance, high-frequency ceramic capacitors are needed
in addition to the bulk capacitors to suppress leading and falling
edge voltage spikes. The result from the high current slew rates
produced by the upper MOSFETs turn on and off. Select low ESL
ceramic capacitors and place one as close as possible to each
upper MOSFET drain to minimize board parasitic impedances
and maximize noise suppression.
FN8705.1
March 7, 2016
ISL68200
TABLE 12. DESIGN AND LAYOUT CHECKLIST
INPUT-CAPACITOR CURRENT (IRMS/IO)
0.6
IL(P-P) = 0.75 IO
0.4
PIN
NAME
NOISE
SENSITIVITY
EN
Yes
There is an internal 1µs filter. Decoupling the
capacitor is NOT needed, but if needed, use a
low time constant one to avoid too large a
shutdown delay.
VIN
Yes
Place 16V+ X7R 1µF in close proximity to VIN
pin and the system ground plane.
7VLDO
Yes
Place 10V+ X7R 1µF in close proximity to
7VLDO pin and the system ground plane.
VCC
Yes
Place X7R 1µF in close proximity to VCC pin
and the system ground plane.
SCL, SDA
Yes
50kHz to 1.25MHz signal when the SMBus,
PMBus, or I2C is sending commands. Pairing
up with SALERT and routing carefully back to
SMBus, PMBus or I2C master. 20 mils spacing
within SDA, SALERT, and SCL; and more than
30 mils to all other signals. Refer to the
SMBus, PMBus or I2C design guidelines and
place proper terminated (pull-up) resistance
for impedance matching. Tie them to GND
when not used.
SALERT
No
Open drain and high dv/dt pin during
transitions. Route it in the middle of SDA and
SCL. Tie it to GND when not used.
PGOOD
No
Open-drain pin. Tie it to ground when not used.
RGND,
VSEN
Yes
Differential pair routed to the remote sensing
points with sufficient decoupling ceramics
capacitors and not across or go above/under
any switching nodes (BOOT, PHASE, UGATE,
LGATE) or planes (VIN, PHASE, VOUT) even
though they are not in the same layer. At least
20 mils spacing from other traces. DO NOT
share the same trace with CSRTN.
CSRTN
Yes
Connect to the output rail side of the output
inductor or current sensing resistor pin with a
series resistor in close proximity to the pin. The
series resistor sets the current gain and should
be within 40Ωand 3.5kΩ. Decoupling
(~0.1µF/X7R) on the output end (not the pin)
is optional and might be required for long
sense trace and a poor layout (see Figures 9
and 10 on page 16).
CSEN
Yes
Connect to the phase node side of the output
inductor or current sensing resistor pin with
L/DCR or ESL/RSEN matching network in close
proximity to CSEN and CSRTN pins.
Differentially routing back to the controller
with at least 20 mils spacing from other
traces. Should NOT cross or go above/under
the switching nodes [BOOT, PHASE, UGATE,
LGATE] and power planes (VIN, PHASE, VOUT)
even though they are not in the same layer.
IL(P-P) = 0
IL(P-P) = 0.5 IO
0.2
0
0
0.2
0.4
0.6
DUTY CYCLE (VOUT/VIN)
0.8
1.0
FIGURE 31. NORMALIZED INPUT-CAPACITOR RMS CURRENT vs
DUTY CYCLE FOR SINGLE-PHASE CONVERTER
Design and Layout Considerations
To ensure a first pass design, the schematics design must be
done right and the board must be carefully laid out.
As a general rule, power layers should be close together, either
on the top or bottom of the board, with the weak analog or logic
signal layers on the opposite side of the board or internal layers.
The ground-plane layer should be in between power layers and
the signal layers to provide shielding, often the layer below the
top and the layer above the bottom should be the ground layers.
There are two sets of components in a DC/DC converter, the
power components and the small signal components. The power
components are the most critical because they switch large
amount of energy. The small signal components connect to
sensitive nodes or supply critical bypassing current and signal
coupling.
The power components should be placed first and these include
MOSFETs, input and output capacitors and the inductor. Keeping
the distance between the power train and the control IC short
helps keep the gate drive traces short. These drive signals
include the LGATE, UGATE, GND, PHASE and BOOT.
When placing MOSFETs, try to keep the source of the upper
MOSFETs and the drain of the lower MOSFETs as close as
thermally possible. Input high frequency capacitors should be
placed close to the drain of the upper MOSFETs and the source of
the lower MOSFETs. Place the output inductor and output
capacitors between the MOSFETs and the load. High frequency
output decoupling capacitors (ceramic) should be placed as
close as possible to the decoupling target, making use of the
shortest connection paths to any internal planes. Place the
components in such a way that the area under the IC has less
noise traces with high dV/dt and di/dt, such as gate signals,
phase node signals and VIN plane.
Tables 12 and 13 provide design and layout checklists that
designer must pay attention to.
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DESCRIPTION
FN8705.1
March 7, 2016
ISL68200
TABLE 12. DESIGN AND LAYOUT CHECKLIST (Continued)
PIN
NAME
NOISE
SENSITIVITY
NTC
Yes
IOUT
Yes
DESCRIPTION
Place NTC 10k (Murata, NCP15XH103J03RC,
 = 3380) in close proximity to the output
inductor’s output rail, not close to MOSFET side
(see Figure 19); the return trace should be 20
mils away from other traces. Place 1.54kΩ
pull-up and decoupling capacitor (typically
0.1µF) in close proximity to the controller. The
pull-up resistor should be exactly tied to the
same point as VCC pin, not through an RC filter.
If not used, connect this pin to VCC.
Scale R such that IOUT pin voltage is 2.5V at
63.875A load. Place R and C in general
proximity to the controller. The time constant
of RC should be sufficient as an averaging
function for the digital IOUT. An external pull-up
resistor to VCC is recommended cancel IOUT
offset at 0A load. See “IOUT Calibration” on
page 19
PROG1-4
No
Resistor divider must be referenced to VCC pin
and the system ground; they can be placed
anywhere. DO NOT use decoupling capacitors
on these pins.
GND
Yes
Directly connect to low noise area of the
system ground. The GND PAD should use at
least 4 vias. Separate analog ground and
power ground with a 0Ω resistor is highly NOT
recommended.
LGATE
No
Low-side driver output and short and wide
trace in between this pin and MOSFET gate pin
as possible. High dV/dt signals should not be
close to any sensitive signals.
UGATE
No
BOOT,
PHASE
Yes
PVCC
Yes
High-side driver output and short and wide
trace in between this pin and MOSFET gate pin
as possible. High dV/dt signals should not be
close to any sensitive signals.
Place X7R 0.1µF or 0.22µF in proximity to
BOOT and PHASE pins. High dV/dt signals
should not be close to any sensitive signals.
Place X7R 4.7µF in proximity to PVCC pin and
the system ground plane.
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TABLE 13. TOP LAYOUT TIPS
#
DESCRIPTION
1
The layer next to controller (top or bottom) should be a ground
layer. Separate analog ground and power ground with a 0Ω
resistor is highly NOT recommended. Directly connect GND
PAD to low noise area of the system ground with at least 4
vias.
2
Never place controller and its external components above or
under VIN plane or any switching nodes.
3
Never share CSRTN and VSEN on the same trace.
4
Place the input rail decoupling ceramic capacitors close to
the high-side FET on the same layer as possible. Never use
only one via and a trace to connect the input rail decoupling
ceramics capacitors; must connect to VIN and GND planes.
5
Place all decoupling capacitors in close proximity to the
controller and the system ground plane.
6
Connect remote sense (VSEN and RGND) to the load and
ceramic decoupling capacitors nodes; never run this pair
below or above switching noise plane.
7
Always double check critical component pinout and their
respective footprints.
Voltage Regulator Design Materials
To support VR design and layout, Intersil also developed a set of
tools and evaluation boards, as listed in Tables 14 and 15,
respectively. Contact Intersil’s local office or field support at
www.intersil.com/ask for the latest available information.
TABLE 14. AVAILABLE DESIGN ASSISTANCE MATERIALS
ITEM
DESCRIPTION
SMBus/PMBus/I2C communication tool with
1
PowerNavigator GUI
2
Evaluation board schematics in OrCAD format and
layout in allegro format. See Table 15 for details.
TABLE 15. AVAILABLE DEMO BOARDS
DEMO BOARD
DESCRIPTION
ISL68200DEMO1Z
17x17mm2 1-phase, 20A solution,
400kHz, with Dual FET
ISL68201_99140DEMO1Z
17x17mm2 1-phase, 35A solution,
400kHz, with ISL99140
FN8705.1
March 7, 2016
ISL68200
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted.
Please go to the web to make sure that you have the latest revision.
DATE
REVISION
CHANGE
March 7, 2016
FN8705.1
Removed unreleased parts from Tables 1 and 15
March 2, 2016
FN8705.0
Initial Release
About Intersil
Intersil Corporation is a leading provider of innovative power management and precision analog solutions. The company's products
address some of the largest markets within the industrial and infrastructure, mobile computing and high-end consumer markets.
For the most updated datasheet, application notes, related documentation and related parts, please see the respective product
information page found at www.intersil.com.
You may report errors or suggestions for improving this datasheet by visiting www.intersil.com/ask.
Reliability reports are also available from our website at www.intersil.com/support.
For additional products, see www.intersil.com/en/products.html
Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted
in the quality certifications found at www.intersil.com/en/support/qualandreliability.html
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time
without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be
accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
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FN8705.1
March 7, 2016
ISL68200
Package Outline Drawing
L24.4x4C
24 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 2, 10/06
4.00
4X 2.5
A
20X 0.50
B
PIN 1
INDEX AREA
PIN #1 CORNER
(C 0 . 25)
24
19
1
18
4.00
2 . 50 ± 0 . 15
13
0.15
(4X)
12
7
0.10 M C A B
0 . 07
24X 0 . 23 +- 0
. 05 4
24X 0 . 4 ± 0 . 1
TOP VIEW
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
C
0 . 90 ± 0 . 1
BASE PLANE
( 3 . 8 TYP )
SEATING PLANE
0.08 C
SIDE VIEW
(
2 . 50 )
( 20X 0 . 5 )
C
0 . 2 REF
5
( 24X 0 . 25 )
0 . 00 MIN.
0 . 05 MAX.
( 24X 0 . 6 )
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
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FN8705.1
March 7, 2016
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