EL4451 Datasheet

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October 1994, Rev A
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8-INT
1-88
EL4451
®
Wideband Variable-Gain Amplifier,
Gain of 2
The EL4451 is a complete variable
gain circuit. It offers wide bandwidth
and excellent linearity while including a
powerful output voltage amplifier, drawing modest supply
current.
FN7169
Features
• Complete variable-gain amplifier with output amplifier,
requires no extra components
• Excellent linearity of 0.2%
• 70MHz signal bandwidth
• Operates on ±5V to ±15V supplies
The EL4451 operates on ±5V to ±15V supplies and has an
analog input range of ±2V, making it ideal for video signal
processing. AC characteristics do not change appreciably
over the ±5V to ±15V supply range.
• All inputs are differential
The circuit has an operational temperature range of -40°C to
+85°C and is packaged in plastic 14-pin DIP and 14-pin SO.
Applications
• 400V/µs slew rate
• > 70dB attenuation @ 4MHz
• Leveling of varying inputs
The EL4451 is fabricated with Elantec’s proprietary
complementary bipolar process which provides excellent
signal symmetry and is free from latch up.
• Variable filters
Pinout
• Text insertion into video
• Fading
Ordering Information
EL4451
(14-PIN PDIP, SO)
TOP VIEW
1
PART
NUMBER
TEMP. RANGE
PACKAGE
PKG. NO.
EL4451CN
-40°C to +85°C
14-Pin PDIP
MDP0031
EL4451CS
-40°C to +85°C
14-Pin SO
MDP0027
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2003. All Rights Reserved. Elantec is a registered trademark of Elantec Semiconductor, Inc.
All other trademarks mentioned are the property of their respective owners.
EL4451
Absolute Maximum Ratings (TA = 25°C)
V+
VS
VIN
∆VIN
IIN
Positive Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . 16.5V
V+ to V- Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . .33V
Voltage at any Input or Feedback . . . . . . . . . . . . . . . V+ to VDifference between Pairs of Inputs or Feedback. . . . . . . . .6V
Current into any Input, or Feedback Pin . . . . . . . . . . . . . 4mA
IOUT
PD
TA
TS
Continuous Output Current . . . . . . . . . . . . . . . . . . . . . . 30mA
Maximum Power Dissipation . . . . . . . . . . . . . . . . See Curves
Operating Temperature Range . . . . . . . . . . . .-40°C to +85°C
Storage Temperature Range. . . . . . . . . . . . .-60°C to +150°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
IMPORTANT NOTE: All parameters having Min/Max specifications are guaranteed. Typical values are for information purposes only. Unless otherwise noted, all tests
are at the specified temperature and are pulsed tests, therefore: TJ = TC = TA
Open-Loop DC Electrical Specifications
PARAMETER
VDIFF
Power Supplies at ±5V, TA = 25°C, RL = 500Ω.
DESCRIPTION
Signal input differential input voltage
Clipping
MIN
TYP
1.8
2.0
V
1.3
V
±2.8
V
±12.8
V
0.2% nonlinearity
VCM
Common-mode range of VIN; VDIFF = 0,
VS = ±5V
±2.0
VS = ±15V
MAX
UNITS
VOS
Input offset voltage
7
25
mV
VOS, FB
Output offset voltage
8
25
mV
VG, 100%
Extrapolated voltage for 100% gain
1.9
2.1
2.2
V
VG, 0%
Extrapolated voltage for 0% gain
-0.16
-0.06
0.06
V
VG, 1V
Gain at VGAIN = 1V
0.95
1.05
1.15
V/V
IB
Input bias current (all inputs)
-20
-9
0
µA
IOS
Input offset current between VIN +and VIN-, Gain+ and Gain-, FB and Ref
0.2
4
µA
NL
Nonlinearity, VIN between -1V and +1V, VG = 1V
0.2
0.5
%
Ft
Signal feedthrough, VG = -1V
-100
-70
dB
RIN, VIN
Input resistance, VIN
100
230
kΩ
RIN, FB
Input resistance, FB
200
460
kΩ
RIN, RGAIN
Input resistance, gain input
50
100
kΩ
CMRR
Common-mode rejection ratio of VIN
70
90
dB
PSRR
Power supply rejection ratio of VOS, FB, VS = ±5V to ±15V
50
60
dB
VO
Output voltage swing (VIN = 0, VREF varied)
VS = ±5V
±2.5
±2.8
V
VS = ±15V
±12.5
±12.8
40
85
ISC
Output short-circuit current
IS
Supply current, VS = ±15V
2
15.5
mA
18
mA
EL4451
Closed-Loop AC Electrical Specifications
PARAMETER
Power supplies at ±12V, TA = 25°C. RL = 500Ω, CL = 15pF, VG = 1V
DESCRIPTION
MIN
TYP
MAX
UNITS
BW, -3dB
-3dB small-signal bandwidth, signal input
70
MHz
BW, ±0.1dB
0.1dB flatness bandwidth, signal input
10
MHz
Peaking
Frequency response peaking
0.6
dB
BW, gain
-3dB small-signal bandwidth, gain input
70
MHz
SR
Slew rate, VOUT between -2V and +2V, RF = RG = 500Ω
400
V/µs
VN
Input referred noise voltage density
110
nV/√Hz
dG
Differential gain error, Voffset between -0.7V and +0.7V
0.9
%
dθ
Differential phase error, Voffset between -0.7V and +0.7V
0. 2
°
Test Circuit
Note: For typical performance curves, RF = 0, RG = ∞, VGAIN = 1V, RL = 500Ω, and CL = 15pF unless otherwise noted.
3
EL4451
Typical Performance Curves
Frequency Response
for Various Feedback
Divider Ratios
Gain, -3dB Bandwidth,
and Peaking
vs Load Resistance
Frequency Response for
Various Gain Settings
4
Frequency Response
for Various RL, CL
VS = ±5V
-3dB Bandwidth and Peaking
vs Supply Voltage
Slew Rate
vs Supply Voltage
Frequency Response
for Various RL, CL
VS = ±15V
-3dB Bandwidth and Peaking
vs Die Temperature
Slew Rate
vs Die Temperature
EL4451
Typical Performance Curves
Common-Mode
Rejection Ratio
vs Frequency
Differential Gain Error
vs Input Offset Voltage
VS = ±5V or ±12V
Differential Gain
and Phase Errors
vs Gain Setting
5
(Continued)
Input Voltage Noise
vs Frequency
Nonlinearity vs
Input Signal Differential Gain
Differential Phase Error
vs Input Offset Voltage
VS = ±5V
Differential Phase Error
vs Input Offset Voltage
VS = ±12V
Differential Gain
and Phase Errors
vs Load Resistance
EL4451
Typical Performance Curves
(Continued)
Gain vs VGAIN
Change in
VG, 100% and VG, 0%
vs Die Temperature
Offset Voltage
vs Die Temperature
Bias Current
vs Die Temperature
Supply Current
vs Die Temperature
Supply Current
vs Supply Voltage
Applications Information
The EL4451 is a complete two-quadrant multiplier/gain
control with 70MHz bandwidth. It has three sets of inputs; a
differential signal input VIN, a differential gain-controlling
input VGAIN, and another differential input which is used to
6
VG, 0% and VG, 100%
vs Supply Voltage
Common Mode
Input Range
vs Supply Voltage
14-Pin Package
Power Dissipation vs
Ambient Temperature
complete a feedback loop with the output. Here is a typical
connection:
EL4451
The gain of the feedback divider is:
H = RG/(RG + RF)
The transfer function of the part is:
VOUT = AO × (((VIN+) - (VIN-)) × ((VGAIN+) - (VGAIN-)) +
(VREF - VFB))
VFB is connected to VOUT through a feedback network, so
VFB = H × VOUT. AO is the open-loop gain of the amplifier,
and is approximately 600. The large value of AO drives:
((VIN+)-(VIN-))×((VGAIN+)-(VGAIN-))+(VREF - VFB) → 0
Rearranging and substituting for VFB:
VOUT = (((VIN+) - (VIN-)) × ((VGAIN+) - (VGAIN)) +
VREF)/H
or
VOUT = (VIN × VGAIN + VREF)/H
Thus the output is equal to the difference of the VIN’s times
the difference of VGAIN’S and offset by VREF, all gained up
by the feedback divider ratio. The EL4451 is stable for a
direct connection between VOUT and FB, and the divider
may be used for higher output gain, although with the
traditional loss of bandwidth.
It is important to keep the feedback divider’s impedance at
the FB terminal low so that stray capacitance does not
diminish the loop’s phase margin. The pole caused by the
parallel impedance of the feedback resistors and stray
capacitance should be at least 150MHz; typical strays of 3pF
thus require a feedback impedance of 360Ω or less.
Alternatively, a small capacitor across RF can be used to
create more of a frequency-compensated divider. The value
of the capacitor should scale with the parasitic capacitance
at the FB input. It is also practical to place small capacitors
across both the feedback and the gain resistors (whose
values maintain the desired gain) to swamp out parasitics.
For instance, two 10pF capacitors across equal divider
resistors for a maximum gain of 4 will dominate parasitic
effects and allow a higher divider resistance.
The REF pin can be used as the output’s ground reference,
for DC offsetting of the output, or it can be used to sum in
another signal.
Gain-Control Characteristics
The quantity VGAIN in the above equations is bounded as
0 ≤ VGAIN ≤ 2, even though the externally applied voltages
exceed this range. Actually, the gain transfer function around
0 and 2V is “soft”, that is, the gain does not clip abruptly
below the 0%-VGAIN voltage nor above the 100%-VGAIN
level. An overdrive of 0.3V must be applied to VGAIN to
obtain truly 0% or 100%. Because the 0%- or 100%- VGAIN
levels cannot be precisely determined, they are extrapolated
from two points measured inside the slope of the gain
7
transfer curve. Generally, an applied VGAIN range of -0.5V to
+2.5V will assure the full numerical span of 0 ≤ VGAIN ≤ 2.
The gain control has a small-signal bandwidth equal to the
VIN channel bandwidth, and overload recovery resolves in
about 20nsec.
Input Connections
The input transistors can be driven from resistive and
capacitive sources, but are capable of oscillation when
presented with an inductive input. It takes about 80nH of
series inductance to make the inputs actually oscillate,
equivalent to four inches of unshielded wiring or 6 of
unterminated input transmission line. The oscillation has a
characteristic frequency of 500MHz. Often placing one’s
finger (via a metal probe) or an oscilloscope probe on the
input will kill the oscillation. Normal high-frequency
construction obviates any such problems, where the input
source is reasonably close to the input. If this is not possible,
one can insert series resistors of around 51Ω to de-Q the
inputs.
Signal Amplitudes
Signal input common-mode voltage must be between
(V-)+3V and (V+)-3V to ensure linearity. Additionally, the
differential voltage on any input stage must be limited to ±6V
to prevent damage. The differential signal range is ±2V in the
EL4451. The input range is substantially constant with
temperature.
The Ground Pin
The ground pin draws only 6µA maximum DC current, and
may be biased anywhere between(V-)+2.5V and (V+)-3.5V.
The ground pin is connected to the IC’s substrate and
frequency compensation components. It serves as a shield
within the IC and enhances input stage CMRR and
feedthrough over frequency, and if connected to a potential
other than ground, it must be bypassed.
Power Supplies
The EL4451 works with any supplies from ±3V to ±15V. The
supplies may be of different voltages as long as the
requirements of the ground pin are observed (see the
Ground Pin section). The supplies should be bypassed close
to the device with short leads. 4.7µF tantalum capacitors are
very good, and no smaller bypasses need be placed in
parallel. Capacitors as small as 0.01µF can be used if small
load currents flow.
Single-polarity supplies, such as +12V with +5V can be
used, where the ground pin is connected to +5V and V- to
ground. The inputs and outputs will have to have their levels
shifted above ground to accommodate the lack of negative
supply.
The power dissipation of the EL4451 increases with power
supply voltage, and this must be compatible with the
EL4451
package chosen. This is a close estimate for the dissipation
of a circuit:
PD = 2 × VS × IS, max + (VS - VO) × VO/RPAR
where
IS, max is the maximum supply current
VS is the ± supply voltage (assumed equal)
Leveling Circuits
Often a variable-gain control is used to normalize an input
signal to a standard amplitude from a modest range of
possible input amplitude. A good example is in video
systems, where an unterminated cable will yield a twicesized standard video amplitude, and an erroneously twiceterminated cable gives a 2/3-sized input.
Here is a ±6dB range preamplifier:
VO is the output voltage
RPAR is the parallel of all resistors loading the output
For instance, the EL4451 draws a maximum of 18mA. With
light loading, RPAR →∞ and the dissipation with ±5V
supplies is 180mW. The maximum supply voltage that the
device can run on for a given PD and other parameters is:
VS, max = (PD + VO2/RPAR) / (2IS + VO/RPAR)
The maximum dissipation a package can offer is:
PD, max = (TJ, max-TA, max) / θJA
Where
TJ,max is the maximum die temperature, 150°C for
reliability, less to retain optimum electrical performance
TA,max is the ambient temperature, 70°C for commercial
and 85°C for industrial range
θJAis the thermal resistance of the mounted package,
obtained from data sheet dissipation curves
The more difficult case is the SO-14 package. With a
maximum die temperature of 150°C and a maximum
ambient temperature of 85°C, the 65°C temperature rise and
package thermal resistance of 120°C/W gives a dissipation
of 542mW at 85°C. This allows the full maximum operating
supply voltage unloaded, but reduced if loaded.
Output Loading
The output stage of the EL4451 is very powerful. It typically
can source 80mA and sink 120mA. Of course, this is too
much current to sustain and the part will eventually be
destroyed by excessive dissipation or by metal traces on the
die opening. The metal traces are completely reliable while
delivering the 30mA continuous output given in the Absolute
Maximum Ratings table in this data sheet, or higher purely
transient currents.
Gain changes only 0.2% from no load to 100Ω load. Heavy
resistive loading will degrade frequency response and video
distortion for loads < 100Ω.
Capacitive loads will cause peaking in the frequency
response. If capacitive loads must be driven, a small-valued
series resistor can be used to isolate it. 12Ω to 51Ω should
suffice. A 22Ω series resistor will limit peaking to 2.5dB with
even a 220pF load.
8
FIGURE 1. LINEARIZED LEVELING AMPLIFIER
In this arrangement, the EL4451 outputs a mixture of the
signal routed through the multiplier and the REF terminal.
The multiplier port produces the most distortion and needs
to handle a fraction of an oversized video input, whereas the
REF port is just like an op-amp input summing into the
output. Thus, for oversized inputs the gain will be decreased
and the majority of the signal is routed through the linear
REF terminal. For undersized inputs, the gain is increased
and the multiplier’s contribution added to the output.
Here are some component values for two designs:
ATTENUATION
RATIO
RF
RG
R1
R2
R3
-3dB
BANDWIDTH
1.5
200Ω 400Ω 300Ω 100Ω 200Ω
47MHz
2
400Ω 400Ω 500Ω 100Ω 200Ω
28MHz
EL4451
EL4451 Leveler Circuit
Attenuation Ratio = 1.5
Here is the EL4451 used as an oscillator with simple AGC:
EL4451 Leveler Circuit
Attenuation Ratio = 2
FIGURE 2. LOW-DISTORTION SINEWAVE OSCILLATOR
With the higher attenuation ratio, the multiplier sees a
smaller input amplitude and distorts less, however the higher
output gain reduces circuit bandwidth. As seen in the next
curves, the peak differential gain error is 0.47% for the
attenuation ratio of 1.5, but only 0.27% with the gain of 2
constants. To maintain bandwidth, an external op amp can
be used instead of the RF - RG divider to boost the EL4451’s
output by the attenuation ratio.
Sinewave Oscillators
Generating a stable, low distortion sinewave has long been a
difficult task. Because a linear oscillator’s output tends to
grow or diminish continuously, either a clipping circuit or
automatic gain control (AGC) is needed. Clipping circuits
generate severe distortion which needs subsequent filtering,
and AGCs can be complicated.
The oscillation frequency is set by the resonance of a seriestuned circuit, which may be an L-C combination or a crystal.
At resonance, the series impedance of the tuned circuit
drops and its phase lag is 0°, so the EL4451 needs a gain
just over unity to sustain oscillation. The VGAIN- terminal is
initially at -0.7V and the VGAIN+ terminal at about +2.1V,
setting the maximum gain in the EL4451. At such high gain,
the loop oscillates and output amplitude grows until D1
rectifies more positive voltage at VGAIN-, ultimately reducing
gain until a stable 0.5Vrms output is produced.
Using a 2MHz crystal, output distortion was -53dBc, or
0.22%. Sideband modulation was only 14Hz wide at -90dBc,
limited by the filter of the spectrum analyzer used.
The circuit works up to 30MHz. A parallel-tuned circuit can
replace the 510Ω resistor and the 510Ω resistor moved in
place of the series-tuned element to allow grounding of the
tuned components.
Filters
The EL4451 can be connected to act as a voltage-variable
integrator as shown:
EL4451 CONNECTED AS VARIABLE INTEGRATOR
9
EL4451
The input RC cancels a zero produced by the output op-amp
feedback connection at = 1/RC. With the input RC connected
VOUT/VIN = 1/sRC; without it VOUT/VIN = (1 + sRC)/sRC.
This variable integrator may be used in networks such as the
Bi-quad. In some applications the input RC may be omitted.
If a negative gain is required, the VIN+ and VIN- terminals
can be exchanged.
A voltage-controlled equalizer and cable driver can be
constructed so:
EQUALIZATION AND LINE DRIVER AMPLIFIER
The main signal path is via the REF pin. This ensures
maximum signal linearity, while the multiplier input is used to
allow a variable amount of frequency-shaped input from R1,
R2, and C. For optimum linearity, the multiplier input is
attenuated by R1 and R2. This may not be necessary,
depending on input signal amplitude, and R1 might be set to
0. R1and R2 should be set to provide sufficient peaking,
depending on cable high-frequency losses, at maximum
gain. RF and RG are chosen to provide the desired circuit
gain, including backmatch resistor loss.
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Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
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10
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