Electronic Ballasts Using the Cost-Saving IR215X Drivers

Application
Notes
AN-995A
Electronic Ballasts Using the Cost-Saving
IR215X Drivers
Introduction
Electronic ballast circuits have recently undergone a
revolution in sophistication from the early bipolar designs of ten years ago. This has been brought about
partly by the advent of power MOSFET switches with
their inherent advantages in efficiency, but mainly by incentives and utility rebate programs sponsored by domestic and foreign governments. New IEC requirements
have also spurred the design of high power factor ballasts and are starting to impose further restrictions on
crest factor, ballast factor and life expectancy (see IEC
555 Standard.)
Until power semiconductors allowed for today’s innovations in ballast design, coil and core fluorescent ballasts were manufactured in large quanitities by a few key
suppliers.
Now there are hundreds of electronics companies that
are “in the ballast business” and more are joining their
ranks all the time.
Most electronic ballasts use two power switches in a
totem pole (half-bridge) topology and the tube circuits
consist of L-C series resonant circuits with the lamp(s)
across one of the reactances. Figure 1 shows this basic
topology.
In this circuit the switches are power MOSFETs driven
to conduct alternately by windings on a current trans-
former. The primary of this transformer is driven by the
current in the lamp circuit and operates at the resonant
frequency of L-C.
Unfortunately, the circuit is not self starting and must
be pulsed by the DIAC connected to the gate of the
lower MOSFET.
After the initial turn-on of the lower switch, oscillation
sustains and a high frequency square wave (30 - 80 kHz)
excites the L-C resonant circuit. The sinusoidal voltage
across C is magnified by the Q at resonance and develops sufficient amplitude to strike the lamp, which then
provides flicker-free illumination.
This basic circuit has been the standard for electronic
ballasts for many years, but has the following inherent
shortcomings:
1) not self starting
2) poor switch times
3) labor intensive torroidal current transformer
4) not amenable to dimming
5) expensive to manufacture in large quantity.
Next Generation Ballast
These criticisms have all been resolved in the new,
cost-saving International Rectifier IR215X Control IC
series.
+
L
C
-
Figure 1. Electronic ballast using transformer drive
C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ MANUAL
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Application
Notes
International Rectifier Control ICs are monolithic
power integrated circuits capable of driving low-side and
high-side MOSFETs or IGBTs from logic level, ground
referenced inputs. They provide offset voltage capabilities up to 600 VDC and, unlike driver transformers, can
provide super-clean waveforms of any duty cycle 0 99%.
The IR215X series is a recent addition to the Control
IC family and, in addition to the above features, these
devices have a front end similar in function to the
CMOS 555 timer IC.
These drivers provide the designer with self-oscillating
or synchronized oscillation functions merely with the
addition of external RT and CT components (Figure 2).
They also have internal circuitry which provides a nominal 1.2 µs dead time between outputs and alternating
high side and low side outputs for driving half-bridge
power switches.
When used in the self oscillating mode the frequency
of oscillation is given by:
f =
1
1.4 × (RT + 75Ω) × C T
(1)
The three available self-oscillating drivers are IR2151,
IR2152 and IR2155.
IR2155 has larger output buffers that switch a 1000 pF
capacitive load with tr = 80 ns and tf = 40 ns. It has micro power start-up and 150 ohm RT source.
IR2151 has tr and tf of 100 ns and 50 ns and functions
similarly to IR2155.
IR2152 is identical to IR2151 but with phase inversion
from RT to LO.
IR2151 and 2152 have 75 ohm RT source (Equation 1.)
These drivers are intended to be supplied from the rectified AC input voltage and for that reason they are designed for minimum quiescent current and have a 15V
internal shunt regulator so that a single dropping resistor
can be used from the DC rectified bus voltage.
Referring again to Figure 2, note the synchronizing capability of the driver. The two back-to-back diodes in series with the lamp circuit are effectively a zero crossing
detector for the lamp current. Before the lamp strikes, the
resonant circuit consists of L, C1 and C2 all in series.
C1 is a DC blocking capacitor with a low reactance, so
that the resonant circuit is effectively L and C2. The voltage across C2 is magnified by the Q factor of L and C2 at
resonance and strikes the lamp.
After the lamp strikes, C 2 is effectively shorted by the
lamp voltage drop and the frequency of the resonant circuit now depends upon L and C1.
This causes a shift to a lower resonant frequency during
normal operation, again synchronized by sensing the zero
crossing of the AC current and using the resultant voltage to
control the driver oscillator.
In addition to the driver quiescent current, there are two
other components of DC supply current that are a function of the actual application circuit:
1) current due to charging the input capacitance of the
power switches
2) current due to charging and discharging the junction
isolation capacitance of the International Rectifier gate
driver.
Both components of current are charge-related and
therefore follow the rules:
Q = CV
(2)
+
VCC
VB
RT
HO
IR2151
CT
VS
RT
COM
C1
LO
L
CT
C2
SYNC
Figure 2. Electronic ballast using IR2151 driver
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C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ M ANUAL
Application
The low power factor circuit shown in figure 3 accepts
115 VAC or 230 VAC 50/60/400 Hz inputs to produce a
nominal DC bus of 320 VDC. Since the input rectifiers
conduct only near the peaks of the AC input voltage, the
input power factor is approximately 0.6 lagging with a
non-sinusoidal current waveform. This type of rectifier
is not recommended for anything other than an evaluation circuit or low power compact fluorescents and indeed may become unacceptable as harmonic currents in
power distribution systems are further reduced by power
quality regulations.
Note that the International Rectifier IR2151 Control
IC operates directly off the DC bus through a dropping
resistor and oscillates at around 45 kHz in compliance
with the following relationship:
It can readily be seen, therefore, that to charge and
discharge the power switch input capacitances, the required charge is a product of the gate drive voltage and
the actual input capacitances and the input power required is directly proportional to the product of charge
and frequency and voltage squared:
Power =
QV 2
×f
2
(3)
The above relationships suggest the following considerations when designing an actual ballast circuit:
1) select the lowest operating frequency consistent
with minimizing inductor size;
2) select the smallest die size for the power switches
consistent with low conduction losses (this reduces the
charge requirements);
3) DC bus voltage is usually specified, but if there is a
choice, use the lowest voltage.
NOTE: Charge is not a function of switching speed.
The charge transferred is the same for 10 ns or 10 µs
switch times.
Let us now consider some practical ballast circuits
which are possible with the self-oscillating drivers. By
far the most popular fluorescent fixture is the so-called
‘Double 40’ type which uses two standard T12 or T8
lamps in a common reflector.
Two suggested ballast circuits are shown in figures 3
and 4. One is a low power factor circuit, and the other
uses a novel diode/capacitor configuration to achieve a
power factor > 0.95.
3 x 0.22
µF
250 VAC
L1
L1
L2
f =
1
1.4 × (RT + 75Ω) × C T
Power for the high side switch gate drive comes from a
bootstrap capacitor of 0.1 µF which is charged to approximately 14V whenever VS (lead 6) is pulled low
during the low side power switch conduction. The bootstrap diode 11DF4 blocks the DC bus voltage when the
high side switch conducts. A fast recovery diode (<100
ns) is required to ensure that the bootstrap capacitor is
not partially discharged as the diode recovers and blocks
the high voltage bus.
The high frequency output from the half-bridge is a
square wave with very fast transition times (approximately 50 ns). In order to avoid excessive radiated noise
4 x 1N4007
+320V
100 µF
200v
+
100 µF
20V
0.01µF
600V
+
fOSC = 45 kHz
P.F. = 0.6 LAG
120 VAC INPUT
USE L1-N
230 VAC INPUT
USE L1-L2
+
1
47 µF
20V
2
15K
3
0.01µF
PTC
V CC VB 8
RT HO
VS
CT
LO
COM
4
+
40W
91K 11DF4
1/2W
IR2151
N
Notes
IRF
720
7
22
6
0.1 µF
1µF
400V
0.01µF
600V
+
L2
IRF 720
5
22
40W
PTC
10Ω
1/2W
0.001µF
600v
L3
*Polyproplylene Capacitor
L1
Core: Micorometals T106-26 L2-L3 Core: TDK PC30EE302 Bobbin: TDK BE30-1110CP
P.T.C. CERA MITE #307C1260BHEAB
18T Bifilar #18 HAPT
64T #22 HATP
Inductance 2 x 30
µH
Inductance 1.35 mH: Gap spacer 0.01 inch
or XFMRS Inc. part #XFO213EE30
Figure 3. ‘Double 40’ ballast using IR2151 oscillator/driver
C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ MANUAL
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Application
Notes
from the fast wave fronts, a 0.5W snubber of 10Ω and
0.001 µF is used to reduce the switch times to approximately 0.5 µs. Note that there is a built-in dead time of
1.2 µs in the IR2151 driver to prevent shoot-through currents in the half-bridge.
The fluorescent lamps are operated in parallel, each
with its own L-C resonant circuit. Up to four tube circuits can be driven from a single pair of MOSFETs sized
to suit the power level.
The reactance values for the lamp circuit are selected
from L-C reactance tables or from the equation for series
resonance:
f=
1
2π LC
the striking voltage with hot filaments is reached and the
lamp strikes.
High Power Factor
The circuit shown in figure 4 is a passive power factor
improvement (no active boost circuit) and is applicable
to low power ballasts such as compact fluorescent. It
suffers from the disadvantage of low DC rectified output
voltage and results in a crest factor of about 2.
Note that a crest factor standard not exceeding 1.7 is
recommended by fluorescent lamp manufacturers to realize the maximum life projections of 20,000 hours for
these lamps.
Peak Current
(4)
Crest Factor = RMS Current
The Q of the lamp circuits is rather low because of the
need for operation from a fixed frequency which, of
course, can vary because of RT and CT tolerances. Fluorescent lamps do not normally require very high striking
voltages so a Q of 2 or 3 is sufficient. ‘Flat Q’ curves
tend to result from larger inductors and small capacitor
ratios where:
Q=
2π fL
R
If the ballast delivers a pure sine wave of voltage and
current to the lamp, the crest factor would be 2 . In an
electronic ballast, a DC bus voltage is derived from a
mains frequency rectifier and is filtered by means of an
electrolytic capacitor. The 2x line frequency ripple voltage on the DC bus gives rise to additional ripple currents
in the lamp. Even if the lamp current is sinusoidal (crest
factor 1.414) the mains-related ripple adds to the peak
current value and causes the crest factor to increase. Referring to the waveforms of figure 5, it is clear that the
ripple voltage amplitude is VP/2 which results in a crest
factor of approximately 2.
What is needed, therefore, is a power factor correction
using active control to minimize current ripple and stabilize the DC bus voltage. Boost regulator correction cir-
(5)
and R tends to be larger as more turns are used.
Soft-starting with tube filament pre-heating can be
cheaply incorporated by using P.T.C. thermistors across
each lamp. In this way, the voltage across the lamp
gradually increases as the P.T.C. self-heats until finally
0.22µF
250 VAC
4 x 1N4007
230 VAC
+ 40W Output
0.22µF
250 VAC
P.F. > 0.96 LAG
1N4007
1N
4007
10µF
200v
10µF
200v
47Ω
1W
Ω resistor
Note: The addition of 47
improves P.F. from 0.94 to 0.96
1N4007
-
Figure 4. Passive high power factor rectifier/filter
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C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ M ANUAL
Application
cuits have become popular for off-line power supplies
and several semiconductor manufacturers supply control
ICs for this topology.
For electronic ballasts, however, the sophistication of
these control chips may not be necessary and it is relatively simple to provide power factors exceeding 0.95 by a
simple boost topology operating at a fixed 50% duty
cycle. Using the IR2151 driver it is also possible to provide dimming merely by changing the duty cycle and,
hence, the boost ratio.
Figures 6* and 7 illustrate how this can be accomplished.
Dimming Control
The IR2151 has a ‘front end’ oscillator circuit akin to
the 555 IC and is amenable to the same type of circuitry
to control the duty cycle of the output waveforms.
Dimming control to 50% of power input is easily
achieved by this control. When R T (lead 2) switches high
the charging path for CT (lead 3) is through the forward
biased diode and the left side of the duty cycle control
pot. When C T charges to two-thirds VCC, R T switches
low and CT discharges through the right side of the control pot. Until the one-third VCC voltage is reached, the
cycle then repeats. Note that although the charge and
discharge times of CT can be varied, the sum of them
remains constant and hence the oscillation frequency is
also constant. This allows sufficient lamp striking voltage even under dimmed conditions.
In actual operation, the ‘on’ time of the boost MOSFET is reduced as RT(CHG) becomes smaller than
AC Voltage and current
PF = 0.96 LAG
200 V/Div., 0.5 A/Div., 5 msec/Div.
Notes
RT(DISCH). Obviously, if the “on” time of the boost
MOSFET is reduced the boost voltage ratio is also reduced proportionately:
boost voltage ratio = VIN ×
1
1−D
e.g., at 50% duty cycle: VIN ×
(6)
1
= 2VIN
0.5
where VIN = instantaneous input voltage and D is the
‘on’ time ratio of the boost MOSFET.
A variation of this circuit, shown in Figure 8, allows
dimming to be controlled remotely by a variable resistor.
The circuits of Figures 7 and 8 both suffer from a basic
flaw; namely that if the lamps are removed or broken the
open circuit DC bus voltage rises until the power
MOSFETs avalanche and fail or the filter capacitor overheats and fails due to overvoltage.
To prevent this, the duty cycle of the boost transistor
can be reduced so that the DC bus is regulated to a constant level, as shown in Figure 9.
In operation, the duty cycle of the boost regulator is
determined by comparing a fraction of the DC bus voltage with a reference triangle wave appearing on the timing capacitor CT. The switching levels of the IR2151
Control IC timing circuit occur at one-third VCC and
two-thirds VCC. Since VCC is regulated by an internal
voltage regulator, the amplitude of the CT waveform is
also regulated.
*U.S. Patent No. 5001400 Nilssen, March 1991
DC Bus voltage showing 50% Vp ripple
100V/Div., 2msec/Div.
Figure 5. Waveforms of Figure 4
C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ MANUAL
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Application
Notes
The LM311 comparator produces a positive output
whenever the instantaneous voltage on CT exceeds a
fraction of the DC bus voltage. This output is ‘OR’ed
with the 50% LO waveform and impedance matched to
drive the boost MOSFET by a 2N2222A emitter follower. The DC bus regulation resulting from this technique is 210-225 VDC with an input AC range of 90
VAC to 130 VAC and dimming from 50% to 100% (225
VDC maximum with bulb removed). Dimming is performed by raising the operating frequency to approximately double for a 50% reduction in power output.
Reliable striking of the lamp is always assured at any
dimming setting because the circuit is synchronized to
the natural resonance of the lamp circuit. Note the backto-back diodes which form a zero current crossing detector for the lamp current and the connection of CT to this
synchronizing voltage (see also figure 2.)
After the lamp strikes, the synchronization circuit is no
longer able to control the frequency which then reverts
to whatever is selected by CT and the variable RT.
In addition to the popular fluorescent ballast applications, there is a growing interest in High Intensity Discharge (HID) ballasts for outdoor lighting. These too can
be simply designed using the International Rectifier
IR2151. A 70 watt high-pressure sodium (HPS) ballast is
illustrated in Figure 10.
0.22 µF
250 VAC
HPS ballasts have some unique requirements not
found in fluorescent ballasts. They must:
• not be damaged when operating into open circuit
• supply sufficient energy at 3 - 4 kV to start the
•
•
•
The circuit shown in figure 10 provides an input power
factor >0.9 and has DC bus control limiting the voltage
to 225 VDC whether or not the lamp is energized. L3
performs two functions:
1) current limiting for the negative resistance
characteristics of the lamp
2) a pulse voltage step-up function to strike the
HPS lamp.
The 3 kV pulse voltage is derived from a 135V SIDAC
which discharges a 1 µF capacitor into the 2 turn winding of L3. The 30:1 step-up ratio of L3 supplies the
starting pulse to the lamp. After the lamp strikes, there is
insufficient charge voltage on the 1 µF capacitor in the 2
turn winding circuit to avalanche the SIDAC and no fur-
L4
0.22 µF
4x
250 VAC 1N4007
10KF6
-
120 VAC
L1
lamp
accomodate large variations in lamp voltage
not cause arc instability in the lamp
be matched to lamp characteristics to maximize
lamp life
91K, 1/2W
10KF6
0.01µF
600V
+
100 µF
200V
10KF6
+
1 V
CC
2
47µF
20V
15K
3
VB 8
HO 7
IR2151
6
CT
VS
RT
4 COM
IRF830
LO 5
22
0.001 µF
+
40
W
PTC
IRF820
1µF
400V
22
01µF
0.01µF
600V
+
10Ω
L2
PTC
0.01 µF
600V
L3
Figure 6. ‘Double 40’ IR2151 ballast with active power factor correction
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C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ M ANUAL
40
W
Application
ther start pulses are supplied. The hot re-strike time of
this ballast is approximately 75 seconds.
The circuits described above have illustrated some of
the ways in which the IR2151 Control IC may be used in
synchronized and non-synchronized ballasts.
Some applications require higher lamp voltages which
may be too high for the simple half-bridge topology. By
using four power MOSFETs in a full bridge circuit, the
output voltage may be doubled without increasing the
MOSFET current. A full bridge circuit automatically
doubles output power and this topology can be implemented with the IR2151 low-cost master oscillator driv-
Notes
ing an IR2111 slave circuit. Figure 12 illustrates this topology which is described in the following text.
This ballast is intended to drive two 80W fluorescent
lamps such as F96-T12 type. These lamps are operated
at the same current as their 48 inch counterparts but require twice the voltage both for striking and normal operation. These slim line lamps have single pin contacts
and are designed to be instantly started from suitable
ballasts. Since the lamps start with cold electrodes, the
ballast must provide in excess of 800V RMS for reliable
starting of any lamp at low ambient temperatures.
The circuit shows a full bridge with each leg driven
from a separate Control IC. U1 is a self-oscillating
driver (IR2151) and U2 is a slave driver (IR2111). The
L1
120
VAC
L1
40
W
VCC
4.7K
VB
RT
HO
IR2151
CT
VS
4.7K
COM
20K
40
W
LO
L2
0.001 µF
IRF820
L3
POWER FACTOR >0.95
FOR COMPONENT VALUES SEE FIGURE 6.
Figure 7. Local dimming by bus voltage control
AC
+15V
51K
1N914
1N914
20K
V CC
RT
CT
51K
VB
IR2151
COM
1 µF
HO
TO LAMP
CIRCUIT
VS
LO
0.001 µF
50K
POWER FACTOR > 0.95
DIMMING
Figure 8. Remote dimming control by variable resistor
C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ MANUAL
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Application
Notes
operation of the IR2151 is the same as previously described for the ‘Simple Double 40 Fluorescent Ballast.’
The full bridge circuit essentially doubles the available
AC output voltage compared with the half-bridge design.
The slave driver U2 is driven from lead 2 of U1 and
provides an inversion of its input signal at lead 2 to the
LO drive waveform at lead 4. U1 does not have this inversion feature so its LO waveform is in phase with pin
2. When driven in this fashion, it is apparent that Q1 and
Q4 conduct together and on the other half cycle Q2 and
Q3 conduct together. The resultant output square wave
has the same RMS value as the DC bus voltage (400
VDC). The lamp circuits are resonant at the self-oscillating frequency of U1 determined from equation (1).
The low-Q lamp circuits have a broad resonance curve
so that tolerance buildups of the timing components R1
and C3 do not seriously compromise the available striking voltage for each lamp. Even with a Q of only 2, the
RMS lamp striking voltage exceeds 800V — more than
sufficient to strike the F96T12 lamps.
Also shown on the schematic is a power factor correction circuit following the AC input rectifier. These circuits
use a boost topology to achieve in-phase AC sinusoidal
current waveforms with low harmonic content, and are
becoming nearly universally required, particularly at
higher power levels. Many papers have been presented on
the subject and a few semiconductor manufacturers provide control chips and application information on their
use.
The ballast circuit will operate with or without a P.F.C.
rectifier; the simplest approach being a configuration
similar to the ‘Simple Double 40 Ballast’ circuit. If this
option is used, the DC bus voltage is around 320 VDC
and the values of L2 and L3 should be reduced by 25%
to around 1 mH (by increasing the core gap.) The R1
value should also be reduced to provide the slightly
higher resonant frequency now required.
Summary
This application note has described a few ballast circuits which are easily implemented with International
Rectifier’s IR215X Control IC family. Additional possibilities are limited only by the imagination of the designer.
This PC board (shown actual size) is designed to drive a 13W to 40W fluorescent lamp using the IR2155, IR2151 or IR2152.)
Input is 115 or 230 VAC. (Schematic, parts list and board available on request. Ask for Design Tips DT 94-3.)
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Application
Notes
L
4 X 1N4007
0.22 µ
250 VAC
N
12K
600 µH
1 µF
DIM
20K
11DF4
4 7µ F
16V
11DF4
15K, 1/2W
+15V
1
IN 914
IRFD
224
V CC
2
10K
2N2222A
160k
3
76 2
LM311 1 +
4
3
CT
4
6.2K
4.7K
RT
VB
IR2151
COM
HO
VS
LO
IRFD224
8
7
0.1µ F
6
4 7µF
250V
0.22 µF
400V
IRFD224
5
0.01µF
26W
0.001 µF
2X1N4001
630 VAC
SYNC
POWER FACTOR >0.95
4.3 mH
Figure 9. Compact fluorescent with dimming and bus voltage control
0.22 µF
250 VAC
L1
0.22 µF
250 VAC
4 X 1N4007
120 VAC
L2
1 µF 400
15K
1/2W
47µF 16V
+225V
11DF4
160k
1
2
10K
1N914
- 3
2
+
1
2N2222A
4
7
IRF
624
NOTE
1 Select value for 70 watt lamp power.
2 Polypropylene capacitor
3 Adjust gap for L = 400µH
L1
6
6.2K
4.7K
1µF 200V
11DF4
L2
(15K)
1
V CC
VB
7
HO
IR2151
3
6
CT
VS
4
RT
8
COM
LO
5
0.001µ F
Core: TDK #EE-30Z
Bobbin: TDK #BE-30-1110CP
Wind: 64T #22 AWG HAPT
L = 720 mH with approximately
0.035 inch gap spacer
Core: Micrometais #T106-26
Wind: 18T Bilifar #18 AWG HAPT
L = 2 x 30µH
IRF720
220K
+ 22 µF
250V
LU
70
22
0.1µF
IRF720
22
L3
2
63T
1µ F
200V
2T
L3
3
Core: Phillips EC-35-3C81
Bobbin: Special 3-Slot bobbin (see Figure 11)
Wind: 63T #22 AWG HAPT (21T/slot)
2T #20 AWG HAPT wind over
Low voltage (near ground) slot
Connect 2-turn winding in series
with 63-turn winding.
Figure 10: 70 watt HPS ballast
Figure 11: Section A-A view of L3 bobbin
C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ MANUAL
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Application
Notes
Figure 12. Full bridge 160 watt fluorescent ballast
Component List
U1
IR2151
U2
IR2111
Q1, Q2, Q3, Q4
IRF820
CR1, CR2, CR3, CR4
1N4007
CR5, CR6
10DF6
R1
15K, ¼W
R2, R3, R4, R5
22Ω, ¼W
C1, C2
0.22 µF 250VAC
C3
0.001 µF, 50V
C4, C7
0.1 µF, 50V
C5, C6
0.01 µF, 1600V, Polypropylene
C8
47 µF, 16V, Aluminum Electrolytic
L1
Core: Micrometals # T106-26
Wind: 18T BIFILAR # 18 AWG HAPT
L = 2 x 30 µH
L2, L3
Core: TDK # EE-30Z
Bobbin: TDK # BE-30-1110CP
Wind: 64T # 22 AWG HAPT
L = 1.35mH with 0.01 inch Gap Spacer
P.F.C.
Motorola MC34262 Data Sheet
Figure 20 Schematic or equivalent from Unitrode, Micro Linear,
SGS Thompson, Cherry, etc.
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C ONTROL I NTEGRATED C IRCUIT D ESIGNERS’ M ANUAL
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