Design Guidelines ICL8001G/ICLS8082G

LED D ri v er IC s
ICL 8001 G / I CLS 808 2G
Des i g n G ui de l i nes
Phase-Cut-Dimmable Single-Stage LED Driver with PFC
using Quasi-Resonant Primary Power Control
Version 2.0
App l i c ati on No te
Version 2.0, 2011-04-14
Indus t ri al & M u l ti m ark e t
Edition 2011-04-14
Published by
Infineon Technologies AG
81726 Munich, Germany
© 2011 Infineon Technologies AG
All Rights Reserved.
Legal Disclaimer
The information given in this document shall in no event be regarded as a guarantee of conditions or
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and liabilities of any kind, including without limitation, warranties of non-infringement of intellectual property rights
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ICL8001G / ICLS8082G
Design Guidelines
ICL8001G / ICLS8082G Design Guidelines
[email protected]
Revision History: 2011-04-14, Version 2.0
Previous Revision:
Page
Subjects (major changes since last revision)
First edition
Application Note
3
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Table of Contents
Table of Contents
Table of Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2
2.1
2.2
2.3
Pin Configuration and Functionality . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Pin Configuration with PG-DSO-8 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Package PG-DSO-8 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Pin Functionality . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
3
Control Principle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
4
4.1
4.2
4.3
4.4
4.5
4.6
4.7
Design Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Transformer Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Power Switch . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Primary Peak Current Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Foldback Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Switch-on Determination for Quasi-resonant Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
VCC for Dimming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
5
5.1
5.1.1
5.1.2
5.1.3
5.1.4
5.2
5.2.1
5.2.2
5.3
Design Optimizations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Dimming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Dimming Principle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Dimmer Compatibility . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Dimmer List . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Dimmer Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Power Stabilization for Line Voltage Variations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Stabilization using IC Foldback Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Discrete Power Stabilization Circuit for ICL8001G . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Optimized Power Factor Correction Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
15
15
15
17
18
19
19
19
20
22
6
6.1
6.2
6.3
Protection Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output OVP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output Short Circuit Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input Overvoltage Protection (OVP) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
23
23
23
23
7
7.1
7.2
Explicit Questions and Answers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Control Principle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8
References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Application Note
4
6
6
7
7
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Introduction
1
Introduction
Objective
The objective of this application note is to describe how a dimmable, highly efficient single-stage LED driver based
on the ICL8001G / ICLS8082G primary control developed by Infineon Technologies AG can be designed and how
different design targets can be considered. For this purpose, quantitative design tools for dimensioning of the
flyback transformer for QR (quasi-resonant) operation and further discrete components relevant to the power
factor correction (PFC) and dimming control functions are provided. The design process refers to a concrete
application example of a phase cut dimmable LED driver. Explicit questions and answers are treated in conclusion.
Features of ICL8001G / ICLS8082G control
•
•
•
•
•
•
•
•
•
High, stable efficiency over a wide operating range
Optimized for trailing and leading-edge dimmers
Precise PWM for primary PFC and dimming control
HV power cell for VCC precharging with a constant current
Built-in digital soft start
Foldback correction and cycle-by-cycle peak current limitation
VCC over/undervoltage lockout
Auto restart mode for short circuit protection
Adjustable latch-off mode for output overvoltage protection (OVP)
Description
The ICL8001G / ICLS8082G employs a quasi-resonant (QR) operation mode and, due to the availability of
outstanding PFC performance, is optimized for off-line LED lighting applications such as dimmable LED bulbs for
incandescent lamp replacement, LED downlights and LED tubes in a power range from typically 5 W to 50 W.
Precise PWM generation enables primary control for phase cut dimming and potential for high power factors of
PF > 99 %. Depending on the power class, significantly improved driver efficiency of up to 91 % is feasible. The
product has a wide operation range of IC voltage supply and low power consumption. Multiple safety functions
ensure full system protection in failure situations. With its full feature set and simple application, the ICL8001G /
ICLS8082G represents an outstanding choice for QR flyback designs combining feature set and performance at
a minimum BOM cost.
Application Note
5
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Pin Configuration and Functionality
Application circuit
Figure 1 below shows the application circuit for the ICL8001G.
T1
Rn
Vin
Dn
Do
np
Cn
Ro
Co
Vo
ns
Rzc1
Dvcc
Cin
L1
Rvcc
DR
Ru
Cvcc
Rc
Vac
Rv
BR
Cx
na
Rzc2 Czc
CR
Cc
L2
VR
ZCV
VCC
U1
HV
Start-Up
Cell
ICL8001G
PWM-Control
Protection
Gate Driver
Vm( Vs )
NC
GD
GND
Q1
CS
Rs
Figure 1
ICL8001G application circuit
2
Pin Configuration and Functionality
2.1
Pin Configuration with PG-DSO-8
Table 1
Pin Configuration for PG-DSO-8
Pin
Symbol
Function
1
ZCV
Zero Crossing
2
VR
Voltage Sense
3
CS
Current Sense
4
GD
Gate Drive Output
5
HV
High Voltage Input
6
n.c.
Not connected
7
VCC
Controller Supply Voltage
8
GND
Controller Ground
Application Note
6
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Pin Configuration and Functionality
2.2
Package PG-DSO-8
ZCV
1
8
GND
VR
2
7
VCC
CS
3
6
NC
GD
4
5
HV
Figure 2
Pin Configuration of PG-DSO-8 (top view)
2.3
Pin Functionality
ZCV (Zero Crossing)
The voltage from the auxiliary winding after a time delay circuit is applied at this pin. Internally, the pin is connected
to the zero-crossing detector for switch-on determination. Additionally, the output overvoltage detection is realized
by comparing the voltage Vzc with an internal preset threshold.
VR (Voltage Sense)
The rectified input mains voltage is sensed at this pin. The signal is used to set the peak current of the peak-current
control and therefore enable the PFC and phase-cut dimming functionality.
CS (Current Sense)
This pin is connected to the shunt resistor for the primary current sensing, externally, and to the PWM signal
generator for switch-off determination (together with the feedback voltage), internally. Moreover, short-winding
protection is realized by monitoring the voltage Vcs during on-time of the main power switch.
GD (Gate Drive Output)
This output signal drives the external main power switch, which is a power MOSFET in most cases.
HV (High Voltage)
The HV pin is connected to the bus voltage, externally, and to the power cell, internally. The current through this
pin precharges the VCC capacitor with a constant current once the supply bus voltage is applied.
VCC (Power supply)
The VCC pin is the positive supply of the IC. The operating range is between VVCCoff and VVCCOVP.
GND (Ground)
This is the common ground of the controller.
Application Note
7
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Control Principle
3
Control Principle
VCC
Zero Crossing
Power Managment
Startup Cell
Over / UnderVoltage Lockout
Zero
Current
Detection
ZCV
HV
Voltage
Reference
& Biasing
Over
Voltage
Protection
Restart /
Latchup
Control
OTP
Foldback
Correction
Depl. CoolMOS 
GND
Gate Drive
Gate
Control
GD
Leading
Edge
Blanking
CS
Short
Winding
Detection
Protection
PWM Control
PM / PFC
Control
PWM
Comparator
VFB
25k
VREF
2pF
VR
Figure 3
VG ( Vs )
Block diagram of the ICL8001G
An inspection of the ICL8001G block diagram shows that the voltage measured at the shunt resistor Rs (see also
the application circuit in Figure 1), which varies according to the instant transformer primary current Ip(t),
determines the gain voltage VG as
VG ( Ip(t ))  GPWM Ip(t ) Rs  VPWM
(1)
With the PWM OP gain GPWM and the offset voltage ram VPWM VG is compared to the voltage VR, which is derived
from the input voltage divider (Ro, Ru). This means that the peak current through the power switch Q1 and the
primary inductance L varies according to the instant input voltage Vin(t) as expressed by
Ip(t ) 
Application Note
Ru
Vin(t )
Ro  Ru GPWM Rs
8
(2)
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Parameters
It can be seen in the non-dimming case that a sinusoidal waveform of the primary peak current is dominant, which
defines the almost sinusoidal shape of the input current Iin(t). Beside the PFC function Equation (2) above shows
that during phase cut dimming the input current will follow the phase cut-modified input voltage, which assures the
dependence of the driver output power on the RMS input voltage Vrms according to the law described by
Equation (24) on page 15.
4
Design Parameters
This chapter presents the essential design process and illustrates it in parallel with concrete values which are
considered for a sample design with the main target magnitudes VLED = 27 V, ILED = 360 mA, PF = 98 %,
η = 90 %, f < fmax = 100 kHz for the input parameters Vin = 230 V, fin = 50 Hz and Pin = VLED ILED / (η PF). The
values chosen for the 10 W/230 V LED driver demo board are close to the following calculation results.
4.1
Transformer Parameters
For QR operation, a sufficiently large reflected voltage from the secondary to the primary side present during the
flyback time tf has to be provided. After discharge of the transformer by means of the secondary winding current
the energy coupled to the reflected voltage is needed to reduce the input voltage at switch-on instantly in the VDS
valley to Vdsmin = Vin - Vr. With the selection of Vdsmin = 0 a value Vr = Vinp, where
Vinp  2Vin
(3)
is received according to
Vr  Vinp  Vds min
(4)
The ratio q = Vr / Vinp with q ≤ 1 determines the maximum input voltage Vinp to be processed with zero-voltage
switching. The winding ratio rn = np / ns is then calculated as rn = 12.
rn 
Vr
VLED  Vdo
(5)
The driver operation frequency f varies over the line voltage phase and reaches its maximum values in the line
voltage phase close to the voltage peak Vinp. Consideration of a QR mode condition defines the primary
inductance L = 6.1 mH (6.3 mH chosen) based on
L
Vin 2
2 Pin f
 Vinp 
1  Vr 
2
(6)
With the selection f = fmax, the equation for the minimum on-time
ton 
2 L Pin 1
f Vin
(7)
yields ton = 5.0 µs and so the minimum duty cycle becomes Dmin = 0.5 as
D min 
Application Note
2 L Pin f
Vin
9
(8)
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Parameters
Now the maximum primary peak current arising in the line phase with Vin = Vinp can be fixed as Ipmaxp = 0.27 A
according to
Pin
Lf
I p max p  2
(9)
2
Assuming a saturation inductance of Bm = 0.4 T and Ae = 20.1 mm a minimum number of primary turns of
np ≈ 200 (np = 190 chosen) would be obtained.
npmin 
L I p max p
Ae Bm
(10)
The maximum magnetic permeance ALmax = 170 nH is obtained using
AL max 
L
np 2
(11)
which determines the air gap using the k-factors in
1
 AL max109  k 2
s

k1


(12)
For the often relevant E16-core, the k-factors are k1 = 42.2 and k2 = –0.701 resulting in s ≈ 0.15 mm. The
secondary number of turns is then given by
ns 
np
rn
(13)
By choosing ns = 14 and the centered IC supply voltage Vccc = ( Vccmax + Vccmin ) / 2 = 19 the auxiliary winding
number will be na = 10 according to
na  ns
4.2
Vcc max
VLED  Vdo
(14)
Power Switch
The maximum voltage VDSmax along the drain-source path of the MOSFET arises during the line phase with Vin
= Vinp at the instant just subsequent to switching off the primary current. In this phase the high-frequency
oscillation amplitude
Vosc  I p max p
Ls
Cds
(15)
adds to the superposition given by the input voltage and reflected voltage to
VDS max  Vinp (1  q )  Vosc
(16)
The VDSmax value can be limited by either reducing the QR effect (with q) at the cost of driver efficiency or by
damping the oscillation by means of a snubber circuit. These limiting approaches can be compared in terms of
Application Note
10
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Parameters
cost and efficiency reduction with the option of choosing MOSFETs with higher breakdown voltages. For high
efficiency it is recommended to make the adjustment VDSmax = 2 Vinp + Vosc < 800 V or 900 V respectively. If
necessary, a transformer design with q < 1 (meaning reduced QR operation effect) should be favored over high
snubber losses.
4.3
Primary Peak Current Control
The peak current charging the primary inductance of the transformer reaches its maximum value at the input
voltage Vinp. By choosing the maximum threshold Vcsmax = 0.75 V for the shunt voltage generated at the instant
of Vinp, the shunt resistance is Rs = 2.7 Ω based on
Rs 
Vcs max
I p max p
(17)
The upper resistor, Ro, in the input voltage divider is dimensioned in consideration of losses and PFC. For
efficiency optimization, high ohmic values are preferred. A low ohmic input voltage divider enables a maximum
power factor adjustment.
The resistance of the lower input voltage divider resistor is expressed as
Ru 
Ro Rs GPWM I p max p
Vinp  GPWM I p max p Rs
(18)
For the selection Vcsmax ≈ 0.75V and Rs = 2.7 Ω the Ru equation above helps to define (with the specification of
Ro = 560 kΩ) a resistance of Ru = 4.3 kΩ (choose Ru = 3.9 kΩ).
Application Note
11
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Parameters
4.4
Foldback Correction
When the line input voltage increases, the comparator threshold for switching off the power switch is reached in a
shorter time and the resulting increase in operation frequency would lead to a strong increase in the LED driver
power. To limit the input voltage dependence above a certain limit Vinth, the foldback correction reduces the
threshold Vcsmax(Icz) in accordance with the current detected at the ZCD pin as expressed here:
Icz (Vin ) 
Vin na
Rcz1 np
(19)
The characteristic curve Vcsmax(Izc) implemented in the IC shows the following steps:
Figure 4
Vcsmax(lzc) characteristic curve
For the adjusted setting Vinth = Vinp and selection of the foldback operation point Vcs-max = 0.75 V / Icz = 1 mA,
using the characteristic curve the detection resistance is calculated to be Rzc1 ≈ 15 kΩ according to
Rcz1 
Vinth na
Icz np
(20)
From the characterisitic curve it can be seen that the Vin dependence of Vcsmax is stronger in the range of higher
threshold values. This means that if a strong impact of the foldback correction is desired, the Vcs should reach
rather high values at Vinp. For increased power constancy under input voltage voltage variations, higher values
of R4 and R19 should be chosen.
Application Note
12
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Parameters
4.5
Switch-on Determination for Quasi-resonant Operation
For ideal quasi-resonant operation the switch-on of the power switch should occur when the transformer current
in the output circuit reaches zero and half of the oscillation period (oscillation 2 in Figure 5) defined by the leakage
inductance and the drain-source capacitance of the power switch has elapsed.
Figure 5
Determination of the switch-on instant
The delay td elapses from the instant of discharging the transformer to when the first VDS valley is reached. For
initial dimensioning of the delay circuit the relationship
td   L Cds
(21)
can be used. For the 230 V / 10 W demo board an initial value of td ≈ 1 µs is appropriate. For output overvoltage
protection with final shutdown at Vo > 45 V it turns out to be useful to choose Rzc2 ≈ 2 kΩ. Consequently, the
capacitance of the delay circuit Czc = 580 pF (choose Czc = 470 pF) will be obtained according to
Czc  td
Application Note
Rzc1  Rzc 2
Rzc1Rzc 2
13
(22)
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Parameters
4.6
Power Factor Correction
The PFC function provides a sinusoidal input current waveform. For this, the setpoint at pin VR should be related
strongly to the input voltage waveform Vin(t). The internal reference voltage Vref is connected via the internal
resistor VFB to the pin VR. The voltage waveform VR(t) over the line period can be influenced in such a way that
a more trapezoidal curve is generated and the input current waveform behaves accordingly. An input voltage
divider with Ro ≈ 500 kΩ at Vac = 230 V delivers a high power factor of typically PF > 95 % and dissipates low
power for high driver efficiency. High PF values above 99 % and low total harmonic distortion of typically 10 % are
achievable at decreased power tolerances. If high power factor adjustment has priority, it is also important to select
an elevated threshold for foldback correction Vinth ≈ Vinp.
Figure 6
Power factor correction
4.7
VCC for Dimming
The dimming feature is closely coupled to the PF control as the variation of the setpoint VR(t) determines the
power flow to the QR-operated flyback. This is also true if the input voltage is shaped by means of a phase cut
dimmer, regardless of whether this is a trailing or leading edge wall dimmer. Therefore the variation of the effective
input voltage determines the power level supplied to the LED load at the driver output. The low frequency variation
of the input power is smoothened by means of an output capacitor Co storing enough energy to cause no visible
light flickering.
For a stable IC supply voltage at phase cut dimming more severe conditions have to be observed. Extended
temporal gaps with zero input voltage, especially at low dimming levels, are present. Also a certain decrease in
output voltage is present in the continuous dimming state of the LED. The Vcc capacitor Cvcc can be designed
using
Cvcc 
IVCC Tgap
VCC dim
(23)
For dimming applications the assumptions of voltage variation ∆VCCdim = 5 V, IVCC = 5 mA and Tgap
≈ 2 x 10 ms for line half-periods with phase cut dimmer malfunction, a Cvcc ≈ 20 µF would be required, while for
non-dimming application the capacitance requirement could be half of that value. For dimmable drivers with Cvcc
= 22 µF a short time of 300 ms to light can be realized due to the charging current provided by the HV start-up cell.
Application Note
14
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
5
Design Optimizations
5.1
Dimming
5.1.1
Dimming Principle
The LED driver input voltage at phase cut dimming with the switching instant ts will generate an effective voltage
of Vineff(ts). From the applied dimming principle it can be seen that the LED power will depend on Vineff(ts)
according to the proportionality law
PLED (Vrms (ts))  (Vrms (ts )) x
(24)
where 1.5 < x < 2.0. Consequently, the light level depends on the RMS input voltage Vrms(ts) provided by the
phase cut dimmer and the dimming range is therefore given by the phase-cut limits of the wall dimmer. For the
example of a trailing edge dimmer, the following diagrams show the time dependence of the modified input voltage
V(t,ts) with the fixed parameter switching instant ts = T/4 and the RMS input voltage Vrms(ts) during a theoretical
full variation of switching instants ts for visualization of the mathematical function Vrms(ts).
Figure 7
Timing dependency on input voltage
Figure 8
Timing dependency on RMS input voltage
Application Note
15
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
Phase cut dimming arrangement
Phase Cut Dimmer
Vac
ICL8001G
+
LED Driver
_
Vac
Figure 9
+
VLED
LED Array
_
Phase cut dimming arrangement
The experimental dependence PLED(Vrms) is illustrated using a 40 Weq LED bulb driver controlled with a trailing
edge phase cut dimmer.
Figure 10
Output power versus input trail edge Vrms
Effective for the application is the dependence of the LED driver power on the switching instant ts / switching angle.
The following experimental diagram obtained with an Ehmann trailing edge dimmer shows the dependence of the
switching instant ts on the driver input power (upper curve) and LED power (lower curve) respectively.
Figure 11
Relationship between switch-off instant ts and power
Application Note
16
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
The lower curve corresponds closely to the lumen flux of the LED array supplied by the driver and illustrates a
continuous increase for the switching angle variation at the wall dimmer.
5.1.2
Dimmer Compatibility
The driver operation with TRIAC dimmers makes it necessary to observe the requirements regarding their holdup currents. In addition, the steep rising voltage slopes can excitate disturbing interactions with the parasitics of
the EMI filter of the driver. To dampen these interactions it is useful to implement a low-pass filter close to the EMI
filter. As placement of the filter in front of the input rectifier would decrease the EMI filter performance it is
recommendable to choose a position parallel to the input voltage divider. In the circuit diagram further below, an
R-C circuit is implemented. Dimensioning of R8 = 220 Ω and C8 = 68 nF (not integrated in the standard demo
board layout) provides stabilization for leading-edge dimmer operation. The non-dimming driver efficiency is
reduced by only typically 1 %.
T1
R1
Vin
5
6
3
8
D5
D21
C25 VLED
np ns
R19
R3 C18
R8
C12
R6
D6
C11
L1
D7
R7
Vac
R2
R17
R5
BR1
C1
1
C15
C8
2
na
C17
C5
L2
VR
ZCV
VCC
Continuous Mode
DIM Control
PFC
NC
HV
Start-Up
Cell
ICL8001G
PWM-Control
Protection
Gate Driver
GD
GND
Q1
CS
R4
Figure 12
Extended application circuit for dimming
Especially for applications with low input voltage and resulting higher input currents, shaping of the input current
waveforms has been shown to be effective in achieving dimming stability. For this, an increase in the ratio
R17 / (R17+ R19) and increase in R4 for keeping the LED power constant is suggested. The principal modification
of the modification on the different waveforms is indicated in Figure 13 below.
Application Note
17
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
Figure 13
Input current waveforms
For an input supply voltage of 100 V the resistors have been changed accordingly: R19 from 3.9 kΩ to 10 kΩ, R19
from 150 kΩ to 300 kΩ and R4 from 2.2 Ω to 2.7 Ω. The increased dimmer compatibility decreases the power
factor from 99 % to 86 %. This modification can be combined with dimensioning for a stronger effect of the
foldback correction. Designs with an increased VR voltage level will lead to a reduction in driver efficiency.
5.1.3
Dimmer List
Table 2
Input voltage 230 V / 10 W
Manufacturer
Type/designation
Power limit
Dimming range
PDL
Leading Edge CAT634LM
450 W
20 – 100 %
Busch
Leading Edge 2200
400 W
16 – 100 %
Ehmann
Leading Edge T10
300 W
3 – 100 %
Ehmann
Trailing Edge T46
315 W
23 – 100 %
HPM
Trailing Edge Cat400T
400 W
6 – 100 %
PDL
Trailing Edge CAT624TM
450 W
10 – 100 %
Table 3
Input voltage 100 V / 10 W
Manufacturer
Type/designation
Power limit
Dimming range
LEVITON
Leading Edge decora RPI06
600 W
11 – 100 %
LEVITON
Leading Edge TRIMATRON 6684
600 W
5 – 100 %
LUTRON
Leading Edge SKYLARK S-600H-WH
600 W
3 – 100 %
LUTRON
Leading Edge SKYLARK S-600P
600 W
4 – 100 %
LUTRON
Leading Edge TOGGLER TG-600PH-IV
600 W
9 – 100 %
Application Note
18
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
5.1.4
Dimmer Selection
With ICL8001G / ICLS8082G control the LED driver can be operated without a wall dimmer as well as using
trailing- and leading-edge dimmers. With the phase-cut dimmers applied above, a satisfying light quality and
variation in the LED lumen flux over the dimming range has been achieved. For both technologies the phase-cut
range should be sufficiently large to provide the dimming range required. The dimming range available with a
particular dimmer for incandescent lamps is directly transferred to the LED driver with ICL8001G / ICLS8082G,
but the lumen output of the LEDs at the minimum dimming position will be higher than with incandescent lamps
as they need a high amount of thermal power to enable the visible light radiation process. Particularly for operation
with 230 V input voltage, leading-edge wall dimmers with a lower rated power range – and hence lower hold-up
current requirements – should be selected in order to prevent the occurrence of repetitive TRIAC ignition.
Operation with lower input voltages, such as 110 V, provides a higher input current, which in principle supports
more stable TRIAC operation.
Touch dimmers with internal IC control, which depends exclusively on a resistive load for sufficient digital IC
supply, should not be selected for LED driver control.
5.2
Power Stabilization for Line Voltage Variations
For stable lumen output and limitation of LED power during mains voltage variations it may be useful to consider
power stabilization for the LED driver design. This can be achieved by appropriate use of IC foldback correction
without additional external components.
5.2.1
Stabilization using IC Foldback Correction
Driver design can be dimensioned in such a way that PLED variation becomes smaller in the range Vin,min < Vin
< Vin,max than would be expected from the approximate quadratic power law. The effect of the foldback correction
on LED power and LED current variations can be enhanced by means of an increased R17 value with a constant
R19. This will modify the input current waveform from sinusoidal to a more rectangular curve. The LED power can
be readjusted by means of the shunt resistor R4. The effect of the foldback correction on the LED power and on
LED current variation can be further increased by additionally lowering the Vcs limit. For this second step the R6
value is decreased.
PLED is readjusted using the shunt resistor R4. The R6 value can be decreased until the latch-off threshold of the
output overvoltage detection is reached. The principal experimental LED current increase for a 10 % input voltage
increase is shown in the following diagram. For input voltage drops, similar behavior is obtained for the LED
current decrease.
Application Note
19
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
Figure 14
LED current behavior according to power factor using foldback correction
The table below shows the relevant design parameters. Consideration of EMI and driver efficiency designs with
PF > 90 % are recommended.
Table 4
Design parameters for stabilization using foldback correction
R17
R19
R6
R4
Pin, rel
[%]
ILEDrel, dec
[%]
ILEDrel, inc
[%]
10k
560k
10k
3R3
4.9
3.8
4
10k
560k
6k8
3R3
6
5
5
3k9
560k
10k
2R7
12
11.6
10
The LED driver should be designed for operation at the maximum voltage arising in the input voltage range.
5.2.2
Discrete Power Stabilization Circuit for ICL8001G
For especially high requirements regarding input power or LED current stability for line voltage variations, such as
∆ILED ≤ +/- 2 % at ∆Vac = +/- 10 %, and for low harmonic distortion, a solution using a discrete differential amplifier
can be proposed. Here the Vin signal is sensed by means of a voltage divider consisting of R21 and R22, and
smoothed by C21. The Zener diode D21 is used for the reference voltage. When the smoothed voltage is much
smaller than the reference voltage, the current though R23 will flow only through R24 and there is no influence on
the output power of the demo board. When the value of the smoothed voltage is near the reference voltage, part
of the current flowing through R23 will begin flowing through R25, R26 and R4 to GND to generate an additional
voltage drop. Subsequently, its waveform shows the same slope at pin CS as the rectified input voltage Vin(t).
Application Note
20
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
Figure 15
Discrete power stabilization circuit
As the on-time of the power MOSFET Q1 is reduced the LED driver power is also reduced. The parameters should
be adjusted so that the smoothed voltage at the base of Q21 is equal to the reference voltage at the base of Q22
at the nominal input voltage. The maximal additional voltage drop on the CS pin should be approximately 10 % to
20 % of the original CS limit at the maximum input voltage. As the waveform of the additional voltage drop at the
CS pin has the same slope as Vin(t), the input current will not change significantly, but still enable designs with a
high power factor and low harmonic distortion. Careful optimization of the discrete stabilization circuit containing
the components stated in the BOM below when connected to the 230 V / 10 W driver delivers variations of
∆ILED ≤ + 0.3 % and ∆ILED ≤ –1.3 % at ∆Vac = +10 % and ∆Vac = –10 % respectively. There is no influence on
output short circuit and output open loop detection.
BOM of Discrete Power Stabilization Circuit for 230V / 10W
ICL8001G LED Driver Demoboard
Figure 16
Component
Value
Package
Q21
Q22
C21
R21
R22
R23
R24
R25
R26
R27
D21
BC860B (PNP)
BC860B (PNP)
470nF/50V
2.2MΩ
75kΩ
1MΩ
10k
10k
1k5
68k
ZMM6V8
SOT23
SOT23
0805
0805
0805
0805
0805
0805
0805
0805
Minimelf
Bill of Materials (BOM) for power stabilization circuit
Application Note
21
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Design Optimizations
5.3
Optimized Power Factor Correction Performance
Driver optimization for high power factor correction can deliver very high PF > 99 % and THD ≈ 10 %.
Corresponding measurements for a 100 V / 10 W LED driver result in the timing diagrams for the setpoint voltage
VR(t) and driver input current Iin(t).
Figure 17
Optimized power factor correction
Figure 18
Bill of Materials (BOM) for 100 V / 10 W LED driver with high PF
Application Note
22
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Protection Functions
6
Protection Functions
The following ICL8001G / ICLS8082G protection functions are provided.
Table 5
ICL8001G / ICLS8082G protection functions
VCC Overvoltage
Auto Restart Mode
VCC Undervoltage
Auto Restart Mode
Output Overvoltage
Latched Off Mode
Output Short Circuit
Auto Restart Mode
Short Winding
Latched Off Mode
Over temperature
Auto Restart Mode
6.1
Output OVP
By means of VCC overvoltage detection it is possible to switch to Auto Restart Mode, which causes a blinking
effect of the LED. The output voltage threshold Voovp_vcc can be adjusted by means of the resistor Rvcc in the
VCC supply circuit.
A second output overvoltage threshold Voovpth can be adjusted to set Latched Off Mode using the IC threshold
Vzcovp = 3.7 V. For a decreased output OV threshold Voovpth = 35 V the Rzc2 resistance value is increased to
Rzc2 = 2.7 kΩ based on the equation
Rzc 2 
6.2
ns Rzc1 Vzcovp
na Voovpth  ns Vzcovp
(25)
Output Short Circuit Protection
In the case of an output short circuit, the IC will switch to Auto Restart Mode by means of VCC undervoltage
detection. No special dimensioning is required for this purpose.
6.3
Input Overvoltage Protection (OVP)
Upon input overvoltage the foldback point correction becomes active and limits the LED driver input power at the
thresholds stated above.
Application Note
23
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
Explicit Questions and Answers
7
Explicit Questions and Answers
7.1
Application
What are the main applications of ICL8001G / ICLS8082G Control?
ICL8001G / ICLS8082G is optimized for offline LED lighting like dimmable LED bulbs for incandescent lamp
replacement, LED downlights and LED tubes. The PFC function enables a power range of typically 5 W to 50 W
to be applied.
Should LED drivers for universal input voltage be designed?
Universal input designs lead to essential efficiency reduction in the LED driver and are therefore not supported by
special IC features.
Is it possible to supply different LED modules with strong variations in total forward voltages?
The LED driver requirements under strongly varying conditions of phase-cut dimming are tough. The VCC concept
compatible with these requirements and proposed here does not allow for strong variations in the DC output
voltage but only for limited output voltage variation. Only in this case is a constant transformer winding ratio or a
design with single transformer connecting points at the auxiliary winding sufficient.
How can driver efficiencies exceeding 90 % be realized?
Generally at higher power applications with pin ≈ 15 W, driver efficiencies > 90% can be realized using optimized
QR control with ICL8001G / ICLS8082G.
Can we disable the dimming function and use it as a constant current LED driver with a good power
factor?
TDA 4863-2 is recommended for this kind of application. For details, see the Application Note AN186.
What will happen under no-load conditions?
The application circuits based on ICL8001G / ICLS8082G are designed to work with LEDs only. Under no-load
conditions it will go to either the auto restart mode due to Vcc OVP or latched off mode due to output OVP.
7.2
Control Principle
How does ICL8001G / ICLS8082G realize dimming control and power factor correction?
Both dimming control and PFC are achieved with the input mains voltage sensing at the VR pin. This signal is used
to set the peak current of the primary winding and consequently allows both PFC and phase-cut dimming
functionality by regulating the cycle energy.
What are the effects of constant power control compared to constant current?
Under constant power control, variations in the LED forward voltage may cause certain variations in the lumen
output while, in contrast, the output remains approximately unchanged under constant LED current control.
However, the power control concept also enables the LEDs to be operated thermally under stable conditions.
Combination of output power control with power stabilization under input voltage variation as proposed above
makes it possible to optimize the heat sink design of an LED bulb or LED lighting fixture and hence minimize
system costs.
How does the ICL8001G / ICLS8082G achieve the regulation of the current through the LEDs?
ICL8001G / ICLS8082G controls the cycle energy stored in the primary inductance of the flyback transformer.
When also considering the frequency behavior, the resulting LED power depends on quite a few parameters. The
influence of input voltage variation can be reduced with the design means described above. The tolerances of the
Application Note
24
Version 2.0, 2011-04-14
ICL8001G / ICLS8082G
Design Guidelines
References
current sense resistor R4, the voltage dividers formed by R17 and R19 as well as R6 & R3 have also to be
considered. The ICL8001G / ICLS8082G PWM parameter GPWM and offset voltage needed for low driver
tolerances possess especially high precision, as stated in the ICL8001G / ICLS8082G data sheet.
8
References
Data Sheet ICL8001G / ICLS8082G
AN EVAL-LED-ICL8001G-Bulb02 for LED Drivers 230 V / 10 W
AN EVAL-LED-ICL8082G-Bulb01 for LED Drivers 230 V / 10 W
AN186 – 40 W LED Street and Indoor Lighting Reference Design
Application Note
25
Version 2.0, 2011-04-14
w w w . i n f i n e o n . c o m
Published by Infineon Technologies AG
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