AN1258

AN1258
Op Amp Precision Design: PCB Layout Techniques
Kumen Blake
Microchip Technology Inc.
INTRODUCTION
This application note covers Printed Circuit Board
(PCB) effects encountered in high (DC) precision op
amp circuits. It provides techniques for improving the
performance, giving more flexibility in solving a given
design problem. It demonstrates one important factor
necessary to convert a good schematic into a working
precision design.
This material is for engineers who design slow
precision circuits, including those with op amps. It is
aimed at those engineers with little experience in this
kind of design, but can also help experienced
engineers that are looking for alternate solutions to a
design problem.
The information in this application note can be applied
to all precision (DC) analog designs, with some thought
and diligence. The focus is on common op amp circuits
so that the reader can quickly convert this material into
improvements in their own op amp designs.
Additional material at the end of the application note
includes references to the literature and the schematic
of a PCB used in the design example.
THERMOCOUPLE JUNCTION
BEHAVIOR
While thermocouples are a common temperature
sensor [5], it is not commonly known that every PCB
design includes many unintended thermocouple
junctions that modify the signal voltages. This section
covers the physics behind this effect and gives
practical illustrations.
Seebeck Effect
When two dissimilar conductors (or semiconductors)
are joined together, and their junction is heated, a
voltage results between them (Seebeck or
thermoelectric voltage); this is known as the Seebeck
effect. This voltage is roughly proportional to absolute
temperature. There are many references that discuss
this effect in detail, including the “Temperature
Products” section of reference [8]; see especially
pages Z-13, Z-14 and Z-23 through Z-32.
Figure 1 shows the Seebeck voltage as a function of
temperature for the standard type K thermocouple.
Notice that the response is not strictly linear, but can be
linearized over small temperature ranges (e.g., ±10°C).
ocouple Voltage (mV)
Thermo
Author:
•
•
•
•
•
•
•
•
Op Amp
Temperature
Thermal Gradient
Thermocouple Junction
Thermoelectric Voltage
IC Sockets
Contact Potential
PCB Surface Contamination
Related Application Notes
The following application notes, together with this one,
form a series about precision op amp design topics.
They cover both theory and practical methods to
improve a design’s performance.
• AN1177 on DC Errors [2]
• AN1228 on Random Noise [3]
 2009-2012 Microchip Technology Inc.
ITS-90
Type K Thermocouple
-300
-200
-100
0
100
200
300
400
500
600
700
800
900
1000
1100
1200
1300
1400
Key Words and Phrases
60
55
50
45
40
35
30
25
20
15
10
5
0
-5
-10
Thermocouple Temperature (°C)
FIGURE 1:
Response.
Type K Thermocouple’s
Most thermocouple junctions behave in a similar
manner. The following are examples of thermocouple
junctions on a PCB:
•
•
•
•
•
Components soldered to a copper pad
Wires mechanically attached to the PCB
Jumpers
Solder joints
PCB vias
DS01258B-page 1
AN1258
The linearized relationship between temperature and
thermoelectric voltage, for small temperature ranges, is
given in Equation 1. The Seebeck coefficients for the
junctions found on PCBs are typically, but not always,
below ±100 µV/°C.
EQUATION 1:
TABLE 1:
Junction No.
VREF
(mV)
kJ
(µV/°C)
1
10
40
2
-4
-10
3
4
10
-10
-40
SEEBECK VOLTAGE
VTH  k J  T J – T REF 
VTH = V REF + V TH
4
Where:
Note 1:
VTH = Change in Seebeck voltage (V)
kJ = Seebeck coefficient (V/°C)
In this illustration, temperature is constant across the
PCB. This means that the junctions are at the same
temperature. Let’s also assume that this temperature is
+125°C and that the voltage on the left trace is 0V. The
results are shown in Figure 3. Notice that VTH is the
voltage change from one conductor to the next.
TREF = Reference Temperature (°C)
VTH = Seebeck voltage (V)
VREF = Seebeck voltage at TREF (V)
Illustrations Using a Resistor
Three different temperature profiles will be shown that
illustrate how thermocouple junctions behave on PCB
designs. Obviously, many other components will also
produce thermoelectric voltages (e.g., PCB edge
connectors).
Figure 2 shows a surface mount resistor with two metal
(copper) traces on a PCB. The resistor is built with end
caps for soldering to the PCB and a very thin
conducting film that produces the desired resistance.
Thus, there are three conductor types shown in this
figure, with four junctions.
Copper
Traces
2:
VREF and kJ have polarities that assume a leftto-right horizontal direction.
TREF = 25 °C.
CONSTANT TEMPERATURE
TJ = Junction Temperature (°C)
Resistor
Film
ASSUMED THERMOCOUPLE
JUNCTION PARAMETERS
14 mV
0 mV
+125.00°C
+125.00°C
FIGURE 2:
on PCB.
Resistor and Metal Traces
For illustrative purposes, we’ll use the arbitrary values
shown in Table 1. Notice that junctions 1 and 4 are the
same, but the values are shown with opposite
polarities; this is one way to account for the direction
current flows through these junctions (the same applies
to junctions 2 and 3).
DS01258B-page 2
+125.00°C
+125.00°C
Location
Resistor
End Caps
Junction #4
Junction #3
14 mV
0 mV
VREF VTH VTH
(mV) (mV) (mV)
Junction #1
10
4
14
Junction #2
-4
-1
-5
Junction #3
4
1
5
Junction #4
-10
-4
-14
FIGURE 3:
Results.
Junction #1
Junction #2
9 mV
Constant Temperature
TEMPERATURE CHANGE IN THE NORMAL
DIRECTION
In this illustration, temperature changes vertically in
Figure 2 (normal to the resistor’s axial direction), but
does not change in the axial direction (horizontally).
The metal areas maintain almost constant voltages in
the normal direction, so this case is basically the same
as the previous one.
Note:
When temperature is constant along the
direction of current flow, the net change in
thermoelectric voltage between two
conductors of the same material is zero.
 2009-2012 Microchip Technology Inc.
AN1258
TEMPERATURE CHANGE IN THE AXIAL
DIRECTION
In this illustration, temperature changes horizontally in
Figure 2 (along the resistor’s axial direction), but does
not change in the normal direction (vertically). Let’s
assume 0V on the left copper trace, +125°C at
Junction #1, a temperature gradient of 10°C/in
(0.394°C/mm) from left to right (0 in the vertical
direction) and a 1206 SMD resistor.
The resistor is 0.12 inches long (3.05 mm) and
0.06 inches wide (1.52 mm). Assume the end caps are
about 0.01 inches long (0.25 mm) and the metal film is
about 0.10 inches long (2.54 mm). The results are
shown in Figure 4.
8.999 mV
14.000 mV
14.010 mV
0 mV
-0.038 mV
PREVENTING LARGE
THERMOELECTRIC VOLTAGES
This section includes several general techniques that
prevent the appearance of large temperature gradients
at critical components.
Reduced Heat Generation
When a PCB’s thermal gradient is mainly caused by
components attached to it, then find components that
dissipate less power. This can be easy to do (e.g.,
change resistors) or hard (change a PICmicro®
microcontroller).
Increasing the load resistance, and other resistor
values, also reduces the dissipated power. Choose
lower power supply voltages, where possible, to further
reduce the dissipated power.
Redirect the Heat Flow
Changing the direction that heat flows on a PCB, or in
its immediate environment, can significantly reduce
temperature gradients. The goal is to create nearly
constant temperatures in critical areas.
+125.0°C
+125.1°C
Location
+126.1°C
+126.2°C
VREF
(mV)
VTH
(mV)
VTH
(mV)
Junction #1
10
4.000
14.000
Junction #2
-4 -1.001
-5.001
Junction #3
Junction #4
FIGURE 4:
1.011
5.011
-10 -4.048
4
-14.048
Axial Gradient Results.
Thus, the temperature gradient of 10°C/in (1.2°C
increase from left to right) caused a total of -38 µV to
appear across this resistor. Notice that adding the
same temperature change to all junction temperatures
will not change this result.
Note:
Shifting all of the junction temperatures by
the same amount does not change the
temperature gradient. This means that the
voltage drop between any two points in the
circuit using the same conductive material
is the same (assuming we’re within the
linear region of response).
 2009-2012 Microchip Technology Inc.
ALTERNATE HEAT PATHS
Adding heat sinks to parts that dissipate a lot of power
will redirect the heat to the surrounding air. One form of
heat sink that is often overlooked is either ground
planes or power planes in the PCB; they have the
advantage of making temperature gradients on a PCB
lower because of their large (horizontal) thermal
conductivity.
Adding a fan to a design will also redirect heat to the
surrounding air, which reduces the temperature drop
on the PCB. This approach, however, is usually
avoided to minimize other design issues (random
temperature fluctuations, acoustic noise, power, cost,
etc.). It is important to minimize air (convection)
currents near critical components. Enclose either the
parts with significant temperature rise, or the critical
parts. Conformal coating may also help.
ISOLATION FROM HEAT GENERATORS
It is possible to thermally isolate critical areas on the
PCB. Regions with little or no metal act like a good
thermal insulator. Signals that need to cross these
regions can be sent through series resistors, which will
also act as poorly conducting thermal elements.
Place heat sources as far away from critical points as
possible. Since many heat sources are in the external
environment, it can be important to place these critical
points far away from the edges of the PCB.
Components that dissipate a lot of power should be
kept far away from critical areas of the PCB.
DS01258B-page 3
AN1258
Low profile components will have reduced exposure to
the external environment. They may have the
additional advantage of reduced electrical crosstalk.
CURING THERMOELECTRIC
VOLTAGE EFFECTS
Thermal barriers, such and conformal coating and PCB
enclosures, can be helpful too. They usually do not
have to be added unless there are other compelling
reasons to do so.
This section focuses on methods that minimize the
effects of a given temperature gradient. They can be
powerful aids in improving a design because they tend
to be low cost.
Slow Temperature Changes
Metallurgy
In some applications, sudden changes
thermoelectric voltages can also be a concern.
in
Avoid power-up and power-down thermal transient
problems by minimizing the currents drawn during
these times. Also, reducing the times can help.
Quick changes in voltages at heavy loads can be
another source of concern. If the load cannot be made
lighter, then isolation is usually the best approach to
solving this problem.
Critical points, that need to have the same total
thermoelectric voltage, should use the same
conductive material. For example, the inputs to an op
amp should connect to the same materials. The PCB
traces will match well, but components with different
constructions may be a source of trouble.
It is possible to find combinations of metals and solders
that have low Seebeck coefficients. While this
obviously reduces voltage errors, this can be
complicated and expensive to implement in
manufacturing.
Following Contour Lines
Place critical components so that their current flow
follows constant temperature contour lines; this
minimizes their thermoelectric voltages. Figure 5
shows an inverting amplifier that will be used to
illustrate this concept; RN, RG and RF are the critical
components in this circuit.
RG
RF
VIN
VOUT
VDD
RN
U1
FIGURE 5:
C1
Inverting Amplifier.
Figure 6 shows one implementation of this concept.
Constant temperature contour lines become
reasonably straight when they are far from the heat
source. Placing the resistors in parallel with these lines
minimizes the temperature drop across them.
Constant
Temperature
Contour
Lines
Heat Source
VIN RF
RG
RN
U1 VDD
C1
VOUT
FIGURE 6:
Contour Lines.
DS01258B-page 4
Resistors Aligned with
 2009-2012 Microchip Technology Inc.
AN1258
The main drawback to this technique is that the contour
lines change when the external thermal environment
changes. For instance, picking up a PCB with your
hands adds heat to the PCB, usually at locations not
accounted for in the design.
Cancellation of Thermoelectric Voltages
It is possible to cancel thermoelectric voltages when
the temperature gradient is constant. Several
examples will be given to make this technique easy to
understand.
TRADITIONAL OP AMP LAYOUT APPROACH
Figure 7 shows a non-inverting amplifier that needs to
have the resistors’ thermoelectric voltage effect
minimized. The traditional approach is to lay out the
input resistors (RN and RG) close together, at equal
distances from the op amp input pins and in parallel.
VIN
U1
C1
VOUT
RG
FIGURE 7:
RF
Non-inverting Amplifier.
Figure 8 shows one layout that follows the traditional
approach, together with a circuit diagram that includes
the resulting thermoelectric voltages (VTHx and VTHy).
VTHx is positive on the right side of a horizontally
oriented component (e.g., RN). VTHy is positive on the
top side of a vertically oriented component (e.g., RF).
RF
RG
RN
VIN
G N = 1 + RF  R G
VOUT =  V IN + V THx G N – V THx  G N – 1  + V THy
= V IN G N + V THx + VTHy
When the gain (GN) is high, the thermoelectric voltage
contributes little to the output error. This layout may be
good enough in that case. Notice that the cancellation
between RN and RG is critical to good performance.
When the gain is low, or the very best performance is
desired, this layout needs improvement. The following
sections give guidance that helps achieve this goal.
SINGLE RESISTOR SUBSTITUTIONS
VDD
Figure 9 shows the original resistor and its model on
the top, and a two series resistor substitution and its
model on the bottom.
The original resistor has a thermally induced voltage
VTHx that is based on the temperature gradient in the
x-direction (horizontal).
The two resistors on the bottom have thermally induced
voltages VTHy that are based on the temperature
gradient in the y-direction (vertical); they are equal
because the temperature gradient is constant and the
resistor lengths are equal. Due to their parallel alignment, these voltages cancel; the net thermally induced
voltage for this combination (as laid out) is zero.
C1
VTHx
C1
U1
R1A R1B
RN VTHx
RF V
THy
FIGURE 8:
One Possible Layout (not
recommended) and its Thermoelectric Voltage
Model.
VTHy
R1B
Where:
R1A = R1B = R1/2
FIGURE 9:
Note:
 2009-2012 Microchip Technology Inc.
VTHy
R1A
VOUT
RG V
THx
R1
R1
VOUT
VDD
VIN
U1
EQUATION 2:
A single resistor on a PCB will produce a thermoelectric
voltage, as discussed before. Replacing that resistor
with two resistors that are properly aligned will cancel
the two resulting thermoelectric voltages.
VDD
RN
The output has a simple relationship to the inputs (VIN
and the three VTHx and VTHy sources):
Series Resistor Substitution.
The orientation of these two resistors (R1A
and R1B) is critical to canceling the
thermoelectric voltages.
DS01258B-page 5
AN1258
Figure 10 shows the original resistor and its model on
the top, and a two parallel resistor substitution and its
model on the bottom.
RG
RN
VIN
The original resistor has a thermally induced voltage
VTHx that is based on the temperature gradient in the
x-direction (horizontal).
The two resistors on the bottom have thermally induced
voltages VTHy that are based on the temperature gradient in the y-direction (vertical); they are equal because
the temperature gradient is constant and the resistor
lengths are equal. Due to their orientation, and
because R1A = R1B, these voltages produce currents
that cancel. The net thermally induced voltage for this
combination (as laid out) is zero.
R1
R1
R1A
VTHx
C1
VDD
VIN
U1
RN VTHx
VOUT
RG V
THx
RF V
THx
FIGURE 12:
First Layout (not recommended) and its Thermoelectric Voltage Model.
The output has a simple relationship to the inputs (VIN
and the three VTHx sources):
R1B
R1A
VTHy
G N = 1 + RF  R G
VOUT =  V IN + V THx G N – V THx  G N – 1  + V THx
= V IN G N + 2V THx
Where:
R1A = R1B = 2R1
FIGURE 10:
Substitution.
Parallel Resistor
NON-INVERTING AMPLIFIER
Figure 11 shows a non-inverting amplifier. We will start
with the layout in Figure 12 (previously shown in
Figure 6). The resistor RF is horizontal so that all of the
thermoelectric voltages may be (hopefully) cancelled.
The model shows how the thermoelectric voltages
modify the circuit.
VIN
C1
VOUT
EQUATION 3:
VTHy
R1B
U1 VDD
RF
U1
When the gain (GN) is high, the thermoelectric
voltage’s contribution to the output error is relatively
small. This layout may be good enough in that case.
Notice that the cancellation between RN and RG is
critical.
We have a better layout shown in Figure 13.
Recognizing that subtracting the last term in the VOUT
equation (middle equation in Equation 3) completely
cancels the thermoelectric voltages, the resistor RF
was oriented in the reverse direction.
RF
RG
RN
VIN
VDD
C1
RN
VOUT
RG
FIGURE 11:
RF
Non-inverting Amplifier.
U1
C1
VOUT
VDD
VIN
VDD
C1
U1
RN VTHx
VOUT
RG V
THx
RF V
THx
FIGURE 13:
Second Layout and its
Thermoelectric Voltage Model.
DS01258B-page 6
 2009-2012 Microchip Technology Inc.
AN1258
With the reversed direction for RF, the output voltage is
now:
VDD
RF
U1
–
RG
VIN
RG
C1
+
RF
VREF VOUT
EQUATION 4:
GN = 1 + RF  RG
V OUT =  VIN + VTHx G N – VTHx  G N – 1  – VTHx
The cancellation between RN and RG is critical to this
layout; the change to RF’s position is not as important.
INVERTING AMPLIFIER
VTHx
VREF
C1
VDD
U1
VOUT
RG V
THx
RF V
THx
FIGURE 16:
Difference Amplifier Layout
and its Thermoelectric Voltage Model.
VDD
U1
The output has a simple relationship to the inputs (VIN,
VREF and the four VTHx sources):
C1
VOUT
RG
+
VIN
RF
–
Inverting amplifiers use the same components as noninverting amplifiers, so the resistor layout is the same;
see Figure 14.
RF
VIN
RG
RN
VTHx
RG
= VIN G N
EQUATION 5:
RF
VIN
VOUT
VDD
RN
U1
C1
G = RF  R G
VOUT =  V IN + V THx – V THx G
+  V REF + V THx – VTHx 
= V IN G + V REF
FIGURE 14:
Inverting Amplifier.
DIFFERENCE AMPLIFIER
Figure 15 shows a difference amplifier. This topology
has an inherent symmetry between the non-inverting
and inverting signal paths, which lends itself to
cancelling the thermoelectric voltages. Figure 16
shows the layout and its model.
RG
RF
VREF
U1
+
VIN
–
VDD
INSTRUMENTATION AMPLIFIER INPUT
STAGE
Figure 17 shows an instrumentation amplifier input
stage, which is sometimes used to drive the input of a
differential ADC. While this is a symmetrical circuit,
achieving good thermoelectric voltage cancellation on
the PCB presents difficulties. It is best to use a dual op
amp, so the RF resistors have to be on both sides of the
op amp, while RG connects both sides; the distances
between resistors are too large to be practical (thermal
gradient is not constant).
C1
U1A
VOUT
RG
FIGURE 15:
RN
RF
Difference Amplifier.
+
VIN
RG
–
VDD
C1
RF
RF
+
VOUT
–
RN
U1B
FIGURE 17:
Instrumentation Amplifier
Input Stage (not recommended).
 2009-2012 Microchip Technology Inc.
DS01258B-page 7
AN1258
The solution to this problem is very simple; split RG into
two equal series resistors so that we can use the
non-inverting layout (see Figure 13) on both sides of
the dual op amp. Each side of this amplifier will cancel
its thermoelectric voltages independently; this is shown
in Figure 18 and Figure 19.
The VTHx sources cancel, for the reasons already
given, so the differential output voltage is simply:
EQUATION 6:
G = 1 + 2R F  R G
VOUT = V IN G
U1A
VDD
+
VIN
–
MODIFICATIONS FOR NON-CONSTANT
TEMPERATURE GRADIENTS
C1
RN
RG/2
RF
RG/2
RF
Temperature gradients are never exactly constant. One
cause is the wide range of thermal conductivities (e.g.,
traces vs. FR4) on a PCB, which causes complex temperature profiles. Another cause is that many heat
sources act like point sources, and the heat is mainly
conducted by a two dimensional object (the PCB); the
temperature changes rapidly near the source and
slower far away.
+
VOUT
–
RN
U1B
FIGURE 18:
Input Stage.
Instrumentation Amplifier
C1
U1
VDD
RF
RG/2
RN
One method is to minimize the size of critical components (e.g., resistors). If we assume that temperature
has a quadratic shape, then using components that are
half as long should reduce the non-linearity error to
about one quarter the size.
VIN+
VOUT+
VDD
VOUT–
VIN–
RF
RG/2
RN
Another method is to keep all heat sources and sinks
far away from the critical components. This makes the
contour lines straighter.
C1
U1A
RN VTHx
+
VIN
–
RG/2 V
THx
RF V
THx
RG/2 VTHx
RF VTHx
+
VOUT
–
RN VTHx
U1B
FIGURE 19:
Instrumentation Amplifier
Input Stage Layout and its Thermoelectric
Voltage Model.
DS01258B-page 8
Non-constant temperature gradients will cause the
temperature profile to have significant curvature, which
causes all of the previous techniques to have less than
perfect success. Usually, the curvature is small enough
so that those techniques are still worth using. Sometimes, additional measures are needed to overcome
the problems caused by the curvature.
The contour lines can be deliberately changed in
shape. Using a ground plane (also power planes) to
conduct heat away from the sources helps equalize the
temperatures, which reduces the non-linear errors.
Adding guard traces or thermal heat sinks that
surround the critical components also help equalize the
temperatures.
We can modify the sizes of the critical components so
that the cancellation becomes closer to exact. In order
to match resistors, for instance, we need to make sure
that the temperature change across each of the
matched resistors is equal; see Figure 20 for an illustration.
 2009-2012 Microchip Technology Inc.
AN1258
(T = +1.0°C)
+25.0°C
+26.0°C
MEASUREMENT OF TEMPERATURE
RELATED QUANTITIES
While the techniques previously shown are a great help
in producing an initial PCB layout, it is important to
verify that your design functions as specified. This
section includes methods for measuring the response
of individual components and of a PCB. With this
information, it is possible to make intelligent design
tweaks.
TEMPERATURE
There are many ways to measure temperature [4, 5, 6].
We could use thermocouples, RTDs, thermistors,
diodes, ICs or thermal imagers (infrared cameras) to
measure the temperature.
+25.1°C
+26.1°C +26.3°C
(T = +1.0°C)
FIGURE 20:
Example of Mismatching the
Component Sizes.
Figure 21 shows a circuit based on the MCP9700 IC
temperature sensor. Because all of the components
draw very little current, their effect on PCB temperature
will be minimal. There is enough filtering and gain to
make VOUT easy to interpret. This circuit can be built on
a very small board of its own, which can be easily
placed on top of the PCB of interest.
VDD = 5.0V
U1
MCP9700
C1
1.0 µF
100 nF
Temp.
Sensor
R1
100 kΩ
VDD
TPCB
C4
100 nF
VDD
U2
R2
MCP6041
113 kΩ
R3
12.4 kΩ
R4
100 kΩ
C2
1.0 µF
FIGURE 21:
Circuit.
R5
1.00 MΩ
R6
1.00 kΩ
VOUT
C3
22 nF
IC Temperature Sensor
The MCP9700 outputs a voltage of about 500 mV plus
10.0 mV/°C times the board temperature (TPCB, in °C).
The amplifier provides a gain of 10 V/V centered on
500 mV (when VDD = 5.0V), giving:
EQUATION 7:
VOUT =  500 mV  + T PCB  100 mV/C 
 2009-2012 Microchip Technology Inc.
DS01258B-page 9
AN1258
Since the MCP9700 outputs a voltage proportional to
temperature, VOUT needs to be sampled by an ADC
that uses an absolute voltage reference.
The absolute accuracy of this circuit does not support
our application, so it is important to calibrate the errors.
Leave the PCB in a powered-off state (except for the
temperature sensor) for several minutes. Measure
VOUT at each point, with adequate averaging. The
changes in VOUT from the calibration value represents
the change in TPCB from the no power condition.
VDD
RF
VDD/2
VREF
RG
PACKAGE THERMAL RESISTANCE
The way to estimate the temperature (T in °C) of a component is to multiply its dissipated power (P in W) by the
package thermal resistance (JA in °C/W). This helps
establish temperature maximum points.
To measure JA, when it is not given in a data sheet,
place the temperature sensor at the IC (usually, a
thermocouple between the package and the PCB).
Insert a small resistor in the supply to measure the
supply current when on (IDD in A). Measure the change
in temperature (T in °C) between the off and on
conditions, supply voltage (VDD in V) and IDD. Then,
RF
RG
VTHx
VDD/2
VREF
C1
U1
VOUT
RG V
THx
RF V
THx
FIGURE 22:
Difference Amplifier with
Deliberately Unbalanced Thermoelectric
Voltages and Heat Generating Resistor.
We can also place a short across one component, of a
matched pair, with a copper trace on the PCB.
Figure 23 shows a non-inverting amplifier layout that
shorts RN (with a copper trace) to unbalance the
thermoelectric voltages. It also connects the two inputs
together, and uses larger resistors, to simplify
measurements (VOUT = VDD/2, ideally). The short is
easily removed from the PCB.
VDD/2
VDD
10 Ohm
U1
RG
(RN)
+
VX
–
C1
VOUT
THERMOELECTRIC VOLTAGES
The large resistor on the right of the layout can be used
to generate heat, causing a horizontal temperature
gradient at the resistors RG and RF. The gain (G) is set
high to make the measurements more accurate. The
thermoelectric voltage (VTHx) across one resistor is:
VTHx
RF
VDD
EQUATION 8:
The easiest way to measure thermoelectric voltages is
to thermally imbalance a difference amplifier circuit.
The thermoelectric voltages have a polarity that adds
(instead of cancelling) in Figure 22 (compare to
Figure 16). The differential input voltage is zero, and
the resistors are larger to emphasize the thermoelectric
voltages.
1W
VOUT
RF
T
 JA = -------------------V DD I DD
+
VX
–
C1
RG
THERMAL GRADIENTS
To measure thermal gradients, simply measure the
temperature at several points on the PCB. The gradient
is then the change in temperature divided by the
distance between points. More points give better
resolution on the gradient, but reduce the accuracy of
the numerical derivative.
10 Ohm
U1
RN
VTHx
VDD
1W
C1
U1
VDD/2
VOUT
RG V
THx
RF V
THx
FIGURE 23:
Shorted Resistor (RN) that
Unbalances Thermoelectric Voltages.
With the unbalance, we now have the thermoelectric
voltage:
EQUATION 10:
G N = 1 + RF  R G
EQUATION 9:
G = RF  RG
VOUT – VREF
V THx = --------------------------------2G + 2
DS01258B-page 10
VOUT – VDD  2
V THx = ------------------------------------–GN
 2009-2012 Microchip Technology Inc.
AN1258
TROUBLESHOOTING TIPS AND TRICKS
LEAKAGE CURRENTS
Using a strip chart to track the change in critical DC
voltages over time helps locate the physical source of
the errors. Not only can it show how large the change
is between two different thermal conditions (e.g., on
and off), but it shows the time constants of these shifts.
They can be roughly divided into the following three
categories:
Leakage currents cause voltage drops when they flow
through either resistors or parasitic resistances. This
section focuses on parasitic resistances presented by
the PCB: surface resistance and bulk (through the
dielectric) resistance.
• Time constant << 1 s, within component (e.g.,
thermal crosstalk within an op amp)
• Time constant  1 s, single component (e.g., in
an eight lead SOIC package)
• Time constant >> 1 s, PCB and its environment
To quickly and easily change the temperature at one
location on a PCB, do the following. Use a clean
drinking straw to blow air at the location (component) of
interest. Use a piece of paper to re-direct the airflow
away from other nearby components. When
troubleshooting, the paper can be used to divide a PCB
area in half to help locate the problem component. This
approach does not give exact numbers, but can be
used to quickly find problem components on a PCB.
You can use a heat sink (with electrically insulating heat
sink compound) to reduce the temperature difference
between two critical points on your PCB. The greater
the area covered at both ends of the heat sink, the
quicker and better this thermal “short” will work.
Leakage currents cause voltage ramps when they flow
into a capacitor. Common examples are the gain
capacitor of a transimpedance amplifier (see
Figure 32) and the non-inverting input of an op amp
with no DC path to ground (not recommended).
Op amp leakage (bias) currents are discussed in [2].
High Impedance Sources
High impedance signal sources are susceptible to
errors caused by leakage currents. These sources are
usually modeled as a current source with a high parallel
resistance (a Norton model):
VIN
IS
FIGURE 24:
RS
Norton Source Model.
One sensor that is modeled as a Norton current source
is the photodiode [1]. A common op amp circuit used
for photocurrent measurements is the transimpedance
amplifier (see Figure 32).
Sometimes, high impedance sources are modeled as a
voltage source with a high series resistance (a
Thevenin model):
VIN
VS
FIGURE 25:
RS
Thevenin Source Model.
The pH electrode [1] is one example with a Thevenin
source. A common op amp implementation is a noninverting amplifier.
PCB Surface Leakage
PARASITIC SURFACE RESISTANCES
Surface contamination on a PCB creates resistive
paths for leakage currents. These leakage currents can
cause appreciable voltage shifts, even in well-designed
circuits. The contamination can be humidity (moisture),
dust, chemical residue, etc.
 2009-2012 Microchip Technology Inc.
DS01258B-page 11
AN1258
On a PCB, leakage currents flow on the surface, to a
sensitive node (high resistance), from nearby bare
metal objects (including traces) at a different voltage.
To model sensitivity to surface contamination in your
circuit, add resistors between the high impedance node
and other nearby nodes. For example, Figure 26
shows an amplifier circuit with a Thevenin source (VS
and RS). Since RS is high and the amplifier’s input is
high impedance, VIN is a high impedance node. Parasitic resistances (RP1 to RP4) are connected to all other
(nearby) voltage nodes (traces on a PCB), including
ground. RP1 to RP4 are open-circuited for most design
work. For leakage current design calculations, they
take on high resistance values (usually one at a time).
RP1
RS
VS
VIN
RP4
CLEANING
A standard PCB clean step helps minimize surface
contamination, but may not eliminate the problem. An
additional cleaning step, using isopropyl alcohol, is
needed to clean the residue left by some PCB cleaning
solvents. This can then be blown dry using compressed
air (with an in-line moisture trap).
COATING
In order to maintain the PCB cleanliness after the initial
clean, you may coat the PCB surface. The coating
needs to be a barrier to moisture and other
contaminants; solder mask, epoxy and silicone rubber
are examples.
The coating will have internal (bulk) leakage currents;
this effect needs to be evaluated for your design.
VDD
RP2
GUARD RINGS
VOUT
RP3
FIGURE 26:
Thevenin Source and Parasitic Leakage Resistances.
Figure 27 shows an amplifier circuit with a Norton
source.
Guard rings surrounding critical signal traces, when
properly applied, can significantly reduce PCB surface
leakage currents into critical (high resistance) nodes.
These guard rings have no solder mask so that the
leakage currents flow into them, instead of into the
sensitive trace. The guard ring is biased at the same
voltage as the sensitive node; it needs to be driven by
a low impedance source.
Guard rings increase the capacitance at critical nodes.
Since they are driven by low impedance sources, these
capacitances have little effect on performance.
Unity Gain Buffer
RP1
VIN
IS
RS
RP2
VDD
VOUT
RP3
Figure 28 shows a unity gain buffer with a guard ring.
This guard ring is biased by VOUT and protects
(surrounds) the op amp’s non-inverting input (and all
top metal connected to it) on the PCB surface. The
diagram is for surface mount components only. RN is
an 0805 SMD to give sufficient clearance for the guard
ring trace between its pads.
The parasitic resistor values (RP) depend on your PCB
layout. For a typical layout, with today’s geometries
(traces are close and short) and materials, we have:
VIN
C1
RN
VOUT
• RP ~ 1000 GΩ, low humidity and contamination
• RP ~ 1 GΩ, high humidity and contamination
These RP values need to be modified for atypical
geometries; see Appendix B: “PCB Parasitic Resistance”. These RP values also need to be modified for
the worst case conditions for your application.
Measurements in your conditions, and with your PCB
layout, will give better estimates of RP.
DS01258B-page 12
VDD
U1
FIGURE 27:
Norton Source and Parasitic
Leakage Resistances.
VDD
VIN
RN
C1
U1
VOUT
FIGURE 28:
Ring.
Unity Gain Buffer with Guard
 2009-2012 Microchip Technology Inc.
AN1258
The parasitic resistances are connected as shown in
Figure 29. RP2 injects current into U1’s non-inverting
input (the high impedance node). The other parasitic
resistors inject current into the guard ring, which is
driven by VOUT; they do not affect the performance.
The voltage across RP2 is U1’s offset voltage (VOS), so
the leakage current is greatly reduced. For instance, if
VOS  ±2 mV and the voltage without the guard ring is
2V, the leakage current would be reduced by a factor of
about 1000.
The parasitic resistances are connected as shown in
Figure 31. Similar to the Unity Gain Buffer in the last
section, U1’s offset voltage (VOS) is across RP2, which
greatly reduces its leakage current. The other parasitic
resistances are driven by VOUT, RF and RG; they do not
affect the performance. The leakage current is typically
reduced by a factor of about 1000.
RP1
RP3
RP2
RP1
RP3
VIN
C1
U1
RP4
C1
U1
RN
RP2
RN
VIN
VDD
VDD
RG
VOUT
VOUT
RF
RP4
RP5
FIGURE 29:
Equivalent Circuit for Unity
Gain Buffer with Guard Ring.
FIGURE 31:
Equivalent Circuit for Noninverting Amplifier with Guard Ring.
One example of an application that sometimes uses
unity gain op amps are pH meters. In that case, however, both VIN and RN are located off the PCB; the
guard ring only needs to surround U1’s non-inverting
input.
Transimpedance Amplifier
Non-inverting Gain Amplifier
Figure 30 shows a non-inverting gain amplifier with a
guard ring. This guard ring is biased by VOUT, RF and
RG; it protects (surrounds) the op amp’s non-inverting
input (and all top metal connected to it) on the PCB
surface. RF and RG are low impedance, to drive the
guard ring properly. RN is a low valued resistor that
cancels thermojunction voltage effects, but has little
effect on bias current errors (e.g., IBRN << ±1 mV).
Figure 32 shows a photo-diode at the input of a
transimpedance amplifier, with a guard ring. This guard
ring is biased at ground; it protects (surrounds) the op
amp’s inverting input (and all top metal connected to it)
on the PCB surface. RF is high valued for the DC gain
(VOUT/ID1).
VDD
RF
U1
CF
C1
D1
VOUT
RF
CF
VDD
U1
RG
RF
VIN
C1
RN
VDD
VIN
C1
U1
FIGURE 30:
with Guard Ring.
RF
VOUT
Non-inverting Gain Amplifier
 2009-2012 Microchip Technology Inc.
VDD
C1
FIGURE 32:
Photo-diode and
Transimpedance Amplifier, with Guard Ring.
RN
RG
U1
D1
VOUT
VOUT
ID1
The parasitic resistances are connected as shown in
Figure 33. Similar to the Non-inverting Gain Amplifier in
the last section, U1’s offset voltage (VOS) is across RP1,
which greatly reduces its leakage current. The other
parasitic resistances are connected to ground; they do
not affect the performance. The leakage current is
typically reduced by a factor of about 1000.
DS01258B-page 13
AN1258
CF
RP1
Top Trace
RP2
RP1
RF
VOUT
ID1
U1
D1
C1
RP3
FIGURE 33:
Equivalent Circuit for Photodiode and Transimpedance Amplifier, with Guard
Ring.
Guard Rings on Both PCB Surfaces
For op amps in through-hole packages (e.g., PDIP),
guard rings are needed on both top and bottom
surfaces. The same design principles apply to both
surfaces.
Any jumper traces (via to other surface, trace and via
back to the original surface), connected to traces with
guard rings, also need guard rings around the jumper
traces. It is better, when possible, to avoid jumper
traces for critical nodes.
PCB Bulk Leakage
The dielectric material used in a PCB (e.g., FR4) is an
insulator. Its resistance to leakage currents through the
bulk (the dielectric) is described by its volume resistivity
(ρV). ρV values vary considerably, depending on the
dielectric and on ambient conditions.
Usually, bulk leakage currents are much smaller than
surface leakage currents; they can be neglected in
many designs. Designs that minimize surface leakage
currents, however, may be affected by bulk currents.
Example 1 shows one example of how bulk leakage
currents occur. Two traces run in parallel and are
separated by the dielectric. The leakage current
between the traces, flowing through the dielectric, is
modelled by a parasitic resistor (see RP1 in Figure 34).
EXAMPLE 1:
Bottom Trace
VDD
PARALLEL TRACES,
OPPOSITE SURFACES
FIGURE 34:
Equivalent Circuit for
Parallel Traces, Opposite Surfaces.
Any two metal areas on the PCB (on a surface or buried
in an inner layer), at different potentials, will have a
leakage current between them. The value of the
parasitic (bulk) resistance depends on:
• The geometry of the areas
- The distance between
- The cross sectional area seen by the current
• Nearby metal objects (e.g., a guard ring) that
modify the current flow path
• The volume resistivity (ρV)
- Dielectric material
- Exposure to chemicals (e.g., water)
The following discussion shows simple techniques to
minimize these leakage currents. See Appendix B:
“PCB Parasitic Resistance” for ways to estimate bulk
leakage currents.
SEPARATION
Moving traces to the surfaces, from inner layers,
increases the distance between them (e.g.,
Example 1).
Example 2 shows two parallel traces, with a distance
separating them. This extra distance increases the
parasitic resistance.
EXAMPLE 2:
TWO PARALLEL TRACES,
WITH OFFSET
Top View
End View
Side View
Top View
End View
DS01258B-page 14
Side View
 2009-2012 Microchip Technology Inc.
AN1258
CROSSED TRACES
GUARD PLANE
When two traces must cross, place them on opposite
surfaces and in normal directions to minimize the
parasitic resistance; see Example 3.
Example 5 shows a trace on the left (node 1), a guard
plane (node 2) and a sensitive trace (node 3). The
guard plane behaves similar to a guard ring, except
that it forms a distributed attenuator to the input voltage
(see Figure 36); this attenuation can be much greater.
EXAMPLE 3:
TWO NORMAL TRACES
Top View
EXAMPLE 5:
TWO PARALLEL TRACES,
WITH GUARD PLANE
Top View
End View
Side View
End View
Side View
GUARD RINGS
Example 4 shows a trace on the left (node 1), a guard
ring (node 2) and a sensitive trace (node 3). The guard
ring provides a low resistance path that redirects some
of the current between node 1 and node 3 to itself.
RPB2 in Figure 35 acts as an attenuator to the input
voltage (V1). When V2 ≈ V3, the parasitic current (into
V3) is significantly reduced.
EXAMPLE 4:
TWO PARALLEL TRACES,
WITH GUARD RING
Top View
RPB1
RPB3
RPB5
RPB7
V1
V3
RPB2
RPB4
RPB6
RPB8
V2 (guard plane)
FIGURE 36:
(Lumped) Equivalent Circuit
for Two Parallel Traces, With Guard Plane.
DIELECTRIC MATERIAL
Changing the dielectric material changes its bulk
resistivity (ρV) and susceptibility to humidity. For
designs that need exceptional performance, this is an
option worth exploring.
MOISTURE CONTROL
End View
Side View
When a dielectric is exposed to moisture for an
extended period of time, it can become wet. This
reduces its bulk resistivity (ρV).
Measures to control exposure to moisture reduce this
effect. One possibility is the use of coatings.
RPB1
ISOLATING SENSITIVE NODES
RPB3
V1
V3
RPB2
V2 (guard ring)
FIGURE 35:
Equivalent Circuit for Two
Parallel Traces, With Guard Ring.
 2009-2012 Microchip Technology Inc.
Another way to minimize PCB leakage currents is to
isolate sensitive nodes (wires, package pins, etc.) from
the board (i.e., not touching).
One approach is to use teflon stand-offs. This has
technical advantages, but can be costly to implement.
Another approach is to keep sensitive nodes in the air.
Bending package leads and routing holes in the PCB
are possible techniques to accomplish this.
DS01258B-page 15
AN1258
OTHER TIPS
DESIGN EXAMPLE
This section discusses other effects and design tips.
This section goes over the thermal design of a thermocouple PCB available from Microchip. This PCB has
the following descriptors:
Packages
IC packages contribute to leakage currents. Pins that
are close together (fine pitch) will see greater leakage
currents, due to dust and shorter leakage paths. The
package itself will have bulk leakage, which depends
on its chemistry.
• MCP6V01 Thermocouple Auto-Zeroed Reference
Design
• 104-00169-R2
• MCP6V01RD-TCPL
The application of this PCB is discussed in detail in
reference [7].
Piezoelectric Effect
Some capacitors accumulate extra charge from
mechanical stress (a variable capacitor), creating a DC
shift. Some ceramic capacitors (not all) suffer from this
effect. Minimize stresses with acoustic noise reduction
techniques, and by making the PCB assembly more
rigid.
Circuit Description
Figure 37 shows the general functionality of this design
(the schematic is shown in Figure A-1).
PC
(Thermal Management Software)
Triboelectric Effect
Mechanical friction can cause charges to accumulate
(a variable capacitor), causing a DC shift. Air flow over
a PCB is one source of mechanical friction. Flexing
wires and coax cables excessively can also cause this
to happen. Shield against air flow, and make bends in
wiring with a large radius. For remote sensors, use low
noise coax or triax cable.
Contact Potential
Sometimes, for convenience on the bench, a PCB has
sockets for critical components (e.g., an op amp).
While these sockets make it easy to change
components, they cause significant DC errors in high
precision designs.
The problem is that the socket and the IC pins are
made of different metals, and are mechanically forced
into contact. In this situation, there is a (contact) voltage
potential developed between the metals (the Volta
effect). Physicists explain this phenomenon through
the difference between their work functions. In our
bench tests of our auto-zeroed op amps, we saw
voltage potentials of ±1 µV to ±2 µV due to the IC
socket.
The solution is very simple; do not use sockets for
critical components. Instead, solder all critical
components to the PCB.
MCP1541
4.1V Reference
USB
PIC18F2250
(USB) Microcontroller
I2C™ Port
SDA,
SCLK,
ALERT
3
CVREF
(cold junction
compensation)
VOUT2
×1
nd
2 Order,
Low-pass
Filter
VSHIFT
MCP9800
Temp. Sensor
VREF
10-bit ADC
Module
VOUT1
TCJ
Type K
Thermocouple
(welded bead)
MCP6V01
Difference Amp.
VP
VM
TTC
Connector
(cold junction)
FIGURE 37:
Block Diagram.
Thermocouple Circuit’s
The (type K) thermocouple senses temperature at its
hot junction (TTC) and produces a voltage at the cold
junction (at temperature TCJ). The conversion constant
for type K thermocouples is roughly 40 µV/°C. This
voltage (VP – VM) is input to the Difference Amplifier
(MCP6V01).
The MCP9800 senses temperature at the Type K
Thermocouple’s cold junction (TCJ). The result is sent
to the PICmicro microcontroller via an I2C™ bus. The
firmware corrects the measured temperature for TCJ.
The difference amplifier uses the MCP6V01
auto-zeroed op amp to amplify the thermocouple’s
output voltage. The VREF input shifts the output voltage
down so that the temperature range includes -100°C.
The VSHIFT input shifts the output voltage, using a
DS01258B-page 16
 2009-2012 Microchip Technology Inc.
AN1258
digital POT internal to the microcontroller (CVREF), so
that the temperature range is segmented into 16
smaller ranges; this gives a greater range (-100°C to
+1000°C) with reasonable accuracy.
The MCP1541 provides an absolute reference voltage
because the thermocouple’s voltage depends only on
temperature (not on VDD). It sets the nominal VOUT and
serves as the reference for the ADC internal to the
microcontroller.
1.
2.
3.
nd
The 2 Order, Low-pass Filter reduces noise and
aliasing at the ADC input. A double R-C filter was
chosen to minimize DC errors and complexity.
CVREF is a digital POT with low accuracy and highly
variable output resistance. The buffer (×1 amplifier)
eliminates the output impedance problem, producing
the voltage VSHIFT. Since CVREF has 16 levels, we can
shift VOUT1 by 16 different amounts, creating 16
smaller ranges; this adds 4 bits resolution to the
measured results (the most significant bits). The 10 bits
produced by the ADC are the least significant bits; they
describe the measured values within one of the 16
different smaller ranges.
VSHIFT is brought back into the PICmicro microcontroller so that it can be sampled by the ADC. This gives
VSHIFT values the same accuracy as the ADC (“10
bits”), which is significantly better than CVREF’s accuracy. The measured value of VOUT2 is adjusted by this
measured VSHIFT value.
The Difference Amplifier is as close to the
sensor as possible, and is on the opposite PCB
surface from the PICmicro microcontroller. This
minimizes electrical and thermal crosstalk
between the two active devices.
Small resistors (0805 SMD) reduce the
thermoelectric voltages, for a given temperature
gradient.
The resistors that are a part of the Difference
Amplifier play a critical role in this design’s
accuracy.
a) R6 and R7 are at the input from the thermocouple, and give a gain of 1000 V/V to
VOUT1. They are arranged so that their
thermoelectric voltages cancel.
b) R9 and R10 are at the input from the range
selection circuitry (VSHIFT), and give a gain
of 17.9 V/V to VOUT1. Changing their
location and orientation on the PCB might
improve the performance.
c) R8 and R11 convert the inputs to the output
voltage (VOUT1). Changing their location
and orientation may not improve the
performance enough to be worth the
trouble.
Figure 39 shows the top metal layer of the PCB. The
sensitive analog and sensor circuitry is connected to
this layer.
The overall accuracy of this mixed signal solution is set
by the 10-bit ADC. The resolution is 14 bits, but the
accuracy cannot be better than the ADC, since it
calibrates the measurements.
4
6
PCB Layout
5
In the figures in this section (Figure 38 through
Figure 43), the red numbers (inside the circles) point to
key design choices, which are described by a list after
each figure.
Figure 38 shows the top silk screen layer of the PCB
designed
for
the
MCP6V01
Thermocouple
Auto-Zeroed Reference Design.
7
6
4
FIGURE 39:
4.
5.
3
2
6.
1
7.
FIGURE 38:
1st Layer – Top Silk.
 2009-2012 Microchip Technology Inc.
1st Layer – Top Metal.
Metal fill, connected to the ground plane,
minimizes thermal gradients at the cold junction
connector.
The MCP9800 Temperature Sensor (cold
junction compensation) is centered at the cold
junction connector to give the most accurate
reading possible.
Sensor traces are separated from power (top
layer) and digital (bottom layer) traces to reduce
crosstalk.
The MCP9800’s power traces are kept short,
straight and above ground plane for minimal
crosstalk.
DS01258B-page 17
AN1258
Figure 40 shows the power plane. It minimizes noise
conducted through the power supplies and isolates the
analog and digital signals.
8
10
9
14. This ground plane extension provides better
isolation between digital signals and the
MCP9800’s power supply. It also helps protect
the thermocouple signal lines. However, it
increases the thermal conduction between the
left and right sides of the PCB.
Figure 42 shows the bottom silk layer.
15
16
17
FIGURE 40:
2nd Layer – Power Plane.
8.
The power plane on the left helps keep the
temperature relatively constant near the
auto-zeroed op amp. It also provides isolation
from the microcontroller’s electrical and thermal
outputs.
9. The power plane on the right helps keep the
temperature relatively constant near the
thermocouple’s cold junction and MCP9800
cold junction temperature sensor.
10. The FR4 gap provides attenuation to heat flow
(a relatively high temperature drop) between the
active components on the left (MCP6V01 and
PIC18F2550) and the sensors on the right (thermocouple and MCP9800).
Figure 41 shows the ground plane. It also minimizes
noise conducted through the power supplies and
isolates the analog and digital signals.
FIGURE 42:
4th Layer – Bottom Silk.
15. The USB connector and its components are
isolated from the rest of the circuit.
16. The crystal (XTAL) oscillator is as far from everything else as possible, except from the clock
input pins of the microcontroller.
17. The microcontroller produces both thermal and
electrical crosstalk, so it is isolated from the analog components.
Figure 43 shows the bottom metal layer of the PCB.
The digital circuitry is connected to this layer.
18
11
13
12
19
14
FIGURE 43:
FIGURE 41:
3rd Layer – Ground Plane.
11. Same function as #8.
12. Same function as #9.
13. Same function as #10.
DS01258B-page 18
4th Layer – Bottom Metal.
18. Metal fill, connected to the ground plane,
minimizes thermal gradients at the cold junction.
19. The digital traces that run under the ground
plane extension have series resistors inserted
inside the FR4 gap. This reduces the thermal
conduction between the sides that solid traces
would produce; otherwise this would become
the worst case thermal conductor between the
microcontroller and the temperature sensors.
 2009-2012 Microchip Technology Inc.
AN1258
SUMMARY
This application note covers thermal effects on Printed
Circuit Boards (PCB) encountered in high (DC)
precision op amp circuits. Causes, effects and fixes
have been covered.
Thermocouple junctions are everywhere on a PCB.
The Seebeck effect tells us that these junctions create
a thermoelectric voltage. This was shown to produce a
voltage across resistors (and other components) in the
presence of a temperature gradient.
Preventing large thermoelectric voltages from
occurring is usually the most efficient way to deal with
thermocouple junctions. The amount of heat generated
on the PCB can be reduced, and the heat flow
redirected away from critical circuit areas. It also pays
to keep any temperature changes from occurring too
quickly.
Any remaining thermoelectric voltage effects need to
be reduced. Choosing the metals, in critical areas, to
have approximately the same work function will
minimize the thermoelectric coefficients of the metal
junctions. Critical components can be oriented so that
they follow constant temperature contour lines. It is
possible to cancel most of the thermoelectric voltage
effects at the input of op amps by correctly orienting
them. Smaller components, spaced closer together, will
also help.
Once a design has been implemented on a PCB, it
pays to measure its thermal response. Information on
where to focus design effort can greatly speed up the
design process. Information has been given on
measuring temperature, thermal gradients, package
JA‘s and troubleshooting tips and tricks.
REFERENCES
Related Application Notes
[1]
AN990, “Analog Sensor Conditioning Circuits –
An Overview,” Kumen Blake; Microchip
Technology Inc., DS00990, 2005.
[2]
AN1177, “Op Amp Precision Design: DC
Errors,” Kumen Blake; Microchip Technology
Inc., DS01177, 2008.
[3]
AN1228, “Op Amp Precision Design: Random
Noise,” Kumen Blake; Microchip Technology
Inc., DS01228, 2008.
Other Application Notes
[4]
AN990, “Analog Sensor Conditioning Circuits –
An Overview,” Kumen Blake; Microchip
Technology Inc., 2005.
[5]
AN684, “Single Supply Temperature Sensing
with Thermocouples,” Bonnie C. Baker; Microchip Technology Inc., DS00684, 1998.
[6]
AN679, “Temperature Sensing Technologies,”
Bonnie Baker,” Microchip Technology Inc.,
DS00679, 1998.
Other References
[7]
User’s Guide, “MCP6V01 Thermocouple
Auto-Zeroed Reference Design,” Microchip
Technology Inc., DS51738, 2008.
[8]
“The OMEGA® Made in the USA Handbook™,”
Vol. 1, OMEGA Engineering, Inc., 2002.
Leakage current effects also need to be minimized.
Methods to accomplish this for surface and bulk
leakage currents have been shown.
Other effects were also discussed.
A design example using the MCP6V01 Thermocouple
Auto-zeroed Reference Design PCB illustrates the
theory and recommendations given in this application
note. The circuit operation is described, then the PCB
layout choices are covered in detail.
At the end of this application note, references to the
literature and an appendix with the design example’s
schematic are provided.
 2009-2012 Microchip Technology Inc.
DS01258B-page 19
FIGURE A-1:
DS01258B-page 20
ALERT
1.0K
R16
1 VDD SDA 5
2
VSS
3ALERT SCLK4
U3
R14
1.0K
THERMO_CONN
J1
CVREF
R15
MCP9800_SOT23-5 1.0K
C11
0.1UF
VDD
U4
VIN–
VDD
4
MCP6001_SOT23-5
2 VSS
3 VIN+
1 VOUT
SCL
SDA
VSHIFT
VDD
R7 100
R6 100
VSHIFT
R9
5.6K
C14
0.1UF
R10
5.6K
VIN
VOUT
VSS
2
1
MCP1541
3
U2
NC
NC 8
7
VDD
6
VOUT
5
U5
R11
100K
VREF
C9
22pF
VREF
CVREF
VSHIFT
VOUT1
VDD
C8
0.47uF
C10
22pF
Y1
20 MHz
C5
1uF
MCP6V01_SOIC
VSS
1 NC
2 VIN–
3
VIN+
4
R8
100K
C12
0.1uF
VDD
VREF
VOUT
RA0/AN0
MCLR/VPP/RE3
RA5/AN4/SS/HLVDIN/C2
RA4/TOCKI/C1OUT/RCV
RA3/AN3/VREF+
R3
1 OHM
C3
10uF
TP1
RB7/KBI3/PGD
RB4/AN11/KBI0
RB5/KBI1/PGM
RB6/KBI2/PGC
SDA
TP3
TP4
RC4/D–/VM
RC5/D+/VP
RC6/TX/CK
RC7/RX/DT/SDO
TP5
ALERT
R4
100K
21
22
23
24
25
26
27
28
C1
0.1UF
USB CONN
7
11
8
6
SDA
10
J3
1 2 3 4 5
VDD
ALERT
R2
10K SCL
VDD
R5
10K
VDD
R1
10K
VDD
N/C
9
15
16
17
18
VDD 20
VSS 19
RB0/AN12/SDI/SDA
RB1/AN10/SCK/SCL
RB2/AN8/INT2/VMO
RB3/AN9/CCP2/VPO
C4
10uF
SCL
VOUT
VOUT2
L1
10uH
TP2
C7
0.1uF
R13
499
R12
499
C6
0.1uF
VDD
U1
PIC18F2550–SOIC28
2nd Order RC
Low-Pass Filter
C2
1uF
VUSB
RC2/CCP1
RC1/T1OSI/CCP2/UOE
RC0/T1OSO/T13CKI
OSC2/CLKO/RA6
OSCI/CLKI
VDD
C13
0.1UF
14
13
12
11
10
9
8 VSS
7
6
5
RA1/AN1
4 RA2/AN2/VREF–/CVREF
3
2
1
VDD
J2
APPENDIX A:
N/C
HDR1X6
AN1258
PCB SCHEMATIC (MCP6V01 THERMOCOUPLE AUTO-ZEROED
REFERENCE DESIGN)
THERMOCOUPLE CIRCUIT’S SCHEMATIC.
 2009-2012 Microchip Technology Inc.
AN1258
APPENDIX B:
PCB PARASITIC
RESISTANCE
The body of this application note emphasized better
PCB designs. For this reason, detailed analyses of
leakage currents were not included.
This appendix shows you how to estimate parasitic
resistances on a PCB. This will help adjust designs
sensitive to leakage currents.
Example B-2 shows two parallel planes on opposite
PCB surfaces. This geometry is useful for measuring
ρV on a given PCB process. Make the planes as large
as possible, to maximize the bulk leakage current.
For instance, a 6 in × 6 in PCB (60 mil thick) could
have two planes 5 in long and 5 in wide. With
ρS = 1 × 105 MΩ and ρV = 1 × 106 MΩ cm, we can
estimate RPS ≈ ∞ and RPB ≈ 0.9 GΩ.
EXAMPLE B-2:
B.1
PCB Resistivities
PARALLEL PLANES
Top View
Surface resistivity (ρS, in units of MΩ, or equivalent)
describes the local, physical behavior of a PCB surface
under a voltage gradient field. The aggregate behavior
over all such points creates an equivalent, parasitic
surface resistance (RPS) between any two points.
Bulk resistivity (ρV, in units of MΩ cm, or equivalent)
describes the local, physical behavior inside a PCB’s
dielectric under a voltage gradient field. The aggregate
behavior over all such points creates an equivalent,
parasitic bulk resistance (RPB) between any two points.
Typical resistivities for FR4, based on several
manufacturer’s data, are:
• ρS ≈ 1 × 105 MΩ
• ρV ≈ 1 × 106 MΩ cm
Different operating conditions and manufacturing flows
will produce different values; sometimes by two or
three orders of magnitude in either direction.
B.2
Measuring Resistivities
Example B-1 shows two parallel, serpentine traces on
the same PCB surface. This geometry is useful for
measuring ρS on a given PCB process. Make the
traces as long as possible, and as close together as
possible, to maximize the surface leakage current.
For example, a 6 in × 6 in PCB (60 mil thick) could
have two serpentine traces 10 mil wide and 10 mil
apart. If the overlap area is 5 in × 5 in, the equivalent
length would be 1250 in. With ρS = 1 × 105 MΩ and
ρV ≈ 1 × 106 MΩ cm, we estimate RPS ≈ 0.8 MΩ and
RPB ≈ 2 GΩ.
EXAMPLE B-1:
PARALLEL TRACES
Top View
End View
Side View
Be sure to measure ρS and ρV under various conditions
seen by your circuit. These values will also change
between different manufacturers and processes.
B.3
B.3.1
Numerical Solutions
COMMERCIAL SOFTWARE
When optimizing a complicated PCB geometry, it may
pay to use a PDE (partial differential equation) solver.
Searching the internet for “partial differential equation
software,” or the equivalent, should bring up several
commercially available software packages.
B.3.2
USING SPICE
It is possible to use SPICE to implement a network of
resistors representing heat flow between adjacent
points.
Select an array of equally spaced points. Resistors
connect adjacent points and represent the local
resistance to current flow, in that direction.
To simulate a particular parasitic resistance, force a
voltage at its input and measure the current at its
output. The voltage-to-current ratio is that resistance.
Figure B-1 shows a typical array point, in a two
dimensional (2D) array. The central point is at voltage
V0. The four adjacent points are connected by the four
resistors R1, R2, R3 and R4.
 2009-2012 Microchip Technology Inc.
DS01258B-page 21
AN1258
V2
R2
V0
V3
R3
R1
V1
R4
V4
FIGURE B-1:
TYPICAL 2D ARRAY.
The resistors represent the local resistivity between
adjacent points. Use very low valued resistors for
points connected by metal. For instance, R1 is:
EQUATION B-1:
R1 = ρS ∆x / ∆y,
for surface calculations
= ρV ∆x / (∆y ∆z),
for bulk calculations
Where:
ρS = Local Surface Resistivity (MΩ)
ρV = Local Bulk Resistivity (MΩ cm)
∆x = grid spacing in the x-direction
∆y = grid spacing in the y-direction
∆z = z-dimension (common to all objects)
This approach is easily extended to three dimensions
(3D).
B.3.3
USING A SPREADSHEET
It is possible to simulate in a spreadsheet by using an
iterative approach. Set up the resistive array like the
last section. The iteration equation at V0 is:
EQUATION B-2:
V0 = (V1/R1 + V2/R2 + V3/R3 + V4/R4)
/ (1/R1 + 1/R2 + 1/R3 + 1/R4)
= (V1 + V2 + V3 + V4) / 4,
all Rs equal
The convergence can be slow, so this approach should
be used for simple problems.
Once the voltages are determined, it is a simple matter
to calculate the sum of currents into (or out of) a node
of interest.
This approach is easily extended to three dimensions
(3D).
DS01258B-page 22
 2009-2012 Microchip Technology Inc.
AN1258
APPENDIX C:
REVISION HISTORY
Revision B (July 2012)
The following is the list of modifications:
1.
2.
3.
4.
5.
6.
Added power supply components to circuit
diagrams.
Re-wrote Section “Leakage Currents”,
starting on page 11.
a) Added information on high impedance
sources and parasitic leakage resistances.
b) Corrected guard ring connections so they
are driven by a low impedance source.
c) Added current source examples.
d) Added discussion of bulk leakage currents.
e) Expanded discussion on isolating sensitive
connections from a PCB.
Added Section “Other Tips”, on page 16.
Added AN990 to Section “References”, on
page 19.
Added Appendix B: “PCB Parasitic
Resistance”, starting on page 21.
Added Appendix C: “Revision History”, on
page 23.
Revision A (March 2009)
• Original Release of this Document.
 2009-2012 Microchip Technology Inc.
DS01258B-page 23
AN1258
NOTES:
DS01258B-page 24
 2009-2012 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
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Trademarks
The Microchip name and logo, the Microchip logo, dsPIC,
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Printed on recycled paper.
ISBN: 978-1-62076-404-6
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CERTIFIED BY DNV
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 2009-2012 Microchip Technology Inc.
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DS01258B-page 25
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