Application Note: STR-W6200D Series PWM Off-Line Switching Regulators

Application Information
STR-W6200D Series PWM Off-Line Switching Regulator ICs
General Description
The STR-W6200D series are power ICs for switching
power supplies, incorporating a power MOSFET and a
current mode PWM controller IC in one package. Including a startup circuit and a standby function in the controller,
the product achieves low power consumption, low standby
power, and high cost-effectiveness in power supply systems,
while reducing external components.
Features and Benefits
▪ TO-220 fully-molded package with 6 pins
▪ Current mode PWM control
▪ PWM and frequency modulation functions: reduces EMI
noise, simplifies EMI filters, and cuts cost by external part
reduction
▪ Built-in Slope Compensation circuit: avoids subharmonic
oscillation
▪ Automatic Standby Mode function (Input Power < 40
mW at no load)
▫ Normal operation: PWM mode
▫ Light load operation: Standby mode (burst oscillation)
▪ Built-in Audible Noise Suppression function during
Standby mode
▪ Built-in startup circuit: reduces power consumption in
standby operation, and eliminates external components
▪ Bias-Assist function: improves startup operation,
suppresses VCC pin voltage drop in operation, and allows
use of smaller VCC capacitor
▪ Built-in Leading Edge Blanking function
▪ Protection Functions:
▫ Overcurrent Protection function (OCP); pulse-by-pulse,
built-in compensation circuit to minimize OCP point
variation on AC input voltage
▫ Overload Protection function (OLP); auto restart, built-in
timer, reduces heat during overload condition, and few
external components required
▫ External Latch Protection function (ELP): latched
shutdown by external signal
▫ Overvoltage Protection function (OVP): latched
shutdown
▫ Thermal Shutdown function (TSD); latched shutdown
STRW6200D-AN Rev.2.0
Figure 1. STR-W6200 series packages are fully molded TO-220
package types. Pin 2 is deleted for greater isolation. A flange is
provided for heatsink mounting.
Applications
Switching power supplies for electronic devices such as:
• White goods
• Consumer electronics
• Office automation
• Industrial equipment
• Communications equipment
The product lineup for the STR-W6200D series provides
the following options:
Part Number
fOSC
(kHz)
MOSFET
VDSS(min)
(V)
STR-W6251D
STR-W6252D
STR-W6253D
67
650
POUT*
(W)
RDS(on)
(max)
(Ω)
230 VAC
85 to
265 VAC
3.95
45
30
2.8
60
40
1.9
90
60
*The listed output power is based on the thermal ratings, and the
peak output power can be 120% to 140% of the value stated here.
At low output voltage and short duty cycle, the output power may
be less than the value stated here.
SANKEN ELECTRIC CO., LTD.
http://www.sanken-ele.co.jp/en/
STARTUP
VCC
D/ST
7.1 V
UVLO
REG
VREG
28.5 V
OVP
7.1 V
R
SQ
ELP
TSD
RQ
S
Dmax 75%
PWM OSC
DRV
SQ
R
S2
Frequency
Modulation
FM /ELP
Istartup
=1.6 mA
RESET
15.5 V / 8.9 V
OLP
Q
Drain Peak Current
Compensation
CK
S1
tDLY =
tFM ×16 R
OCP
7.8 V
160 μA
Feedback
Control
FB
LEB
Slope
Compensation
S/OCP
GND
Pin List Table
Number
Name
Function
1
D/ST
2
–
3
S/OCP
4
VCC
Power supply voltage input for Control Part and input of Overvoltage
Protection (OVP) signal
5
GND
Ground
6
FB
7
FM/ELP
MOSFET drain and input of the startup current
(Pin removed)
MOSFET source and input of Overcurrent Protection (OCP) signal
Input for constant voltage control signal
Capacitor connection pin for frequency modulation and input of
External Latch Protection
Table of Contents
General Description
1
Product Lineup
1
Functional Block Diagram
2
Package Diagram
3
Electrical Characteristics
4
Typical Application Circuit
6
Functional Description
7
Startup Operation
Frequency Modulation Function
Automatic Standby Mode Function
STRW6200D-AN Rev.2.0
7
9
10
Constant Output Voltage Control
Fault Latch
External Latch Protection Function (ELP)
Overload Protection Function (OLP)
Overvoltage Protection Function (OVP)
Overcurrent Protection Function (OCP)
Thermal Shutdown Function (TSD)
Design Notes
10
11
11
12
12
12
13
14
Peripheral Components
14
Phase Compensation
14
Capacitance of External Capacitor at FM/ELP Pin14
Secondary Diode EMI Measure
15
PCB Trace Layout and Component Placement 15
SANKEN ELECTRIC CO., LTD.
2
Package Diagram
TO-220F-6L package
Leadform: 2003
10.0 ±0.2
4.2 ±0.2
Gate Burr
Ø3.2 ±0.2
16.9 ±0.3
7.9 ±0.2
4.0 ±2
0.5
2.8 ±0.2
+0.2
End of bend
(2
R1
)
(5.4)
6X0.65 –0.1
10.4 ±0.5
6X0.74 ±0.15
5.0 ±0.5
2.8
2.6 ±0.1
(At base of pin)
View B
View A
1.27 ±0.15
6×1.27 ±0.15 = 7.62 ±0.15
+0.2
0.45 –0.1
(At base of pin)
3
5
5.08 ±0.6
7
(At tip of pin)
1
2
4
6
0.5
0.5
View A
Bottom View
0.5
0.5
View B
Unit: mm
Dashed line at Gate Burr indicates
protrusion of 0.3 mm (max)
STR
W62xx
YMDDR
Part Number
Lot Number
Y is the last digit of the year (0 to 9)
M is the month (1 to 9, O, N, or D)
DD is a period of day (01 to 31)
R is the Sanken Registration Number
Pin treatment Pb-free. Device composition
compliant with the RoHS directive.
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
3
Electrical Characteristics
• This section provides separate sets of electrical characteristic
data for each product.
• The polarity value for current specifies a sink as "+ ," and a
source as “−,” referencing the IC.
Absolute Maximum Ratings Unless specifically noted, valid at TA = 25°C
Characteristic
Symbol
Note
Pin
STR-W6251D
Drain Peak Current
Maximum Switching Current
Avalanche Energy
IDPEAK
IDMAX
EAS
Rating
Unit
2.6
A
3.2
A
STR-W6253D
10
A
STR-W6251D
2.6
A
3.2
A
STR-W6252D Single Pulse
STR-W6252D TA = –20°C to 125°C
1-3
1-3
STR-W6253D
10
A
STR-W6251D ILPEAK = 2 A
47
mJ
62
mJ
86
mJ
Single Pulse,
STR-W6252D ILPEAK = 2.3 A VDD = 99 V,
STR-W6253D ILPEAK = 2.7 A L = 20 mH
1-3
S/OCP Pin Voltage
VOCP
3-5
–6 to 6
V
FM/ELP Pin Voltage
VFM
7-5
–0.3 to 12
V
FM/ELP Pin Sink Current
IFM
FB Pin Voltage
VFB
Controller Part Input Voltage
VCC
FB pin is open
7-5
3
mA
6-5
–0.3 to 9
V
4-5
0 to 32
V
25
W
STR-W6251D
MOSFET Power Dissipation
PD1
STR-W6252D With infinite heatsink
STR-W6253D
26
W
27.5
W
1.3
W
4-5
0.8
W
–
–20 to 115
°C
1-3
Without heatsink
Controller Part Power Dissipation
Internal Frame Temperature in Operation
PD2
TF
Recommended operating temperature is
TF = 105°C (max)
Operating Ambient Temperature
Top
–
–20 to 115
°C
Storage Temperature
Tstg
–
–40 to 125
°C
Channel Temperature
Tch
–
150
°C
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
4
Electrical Characteristics of Control Part Unless specifically noted, TA is 25°C, VCC = 18 V
Characteristic
Symbol
Pin
Min.
Typ.
Max
Unit
Operation Start Voltage
VCC(ON)
4-5
13.9
15.5
17.1
V
Operation Stop Voltage
VCC(OFF)
4-5
8.0
8.9
9.8
V
ICC(ON)
4-5
–
1.4
2.8
mA
Circuit Current in Non-Oscillation
ICC(STOP)
4-5
–
0.8
1.3
mA
Circuit Current in Non-Operation
ICC(OFF)
4-5
–
5
20
μA
Startup Current
ISTARTUP
4-5
–0.9
–1.6
–2.3
mA
Bias Assist Voltage
VCC(BIAS)
4-5
13.6
15.2
16.8
V
FM/ELP Pin High Threshold Voltage
VFM(H)
7-5
4.0
4.5
5.0
V
FM/ELP Pin Low Threshold Voltage
VFM(L)
7-5
2.4
2.8
3.2
V
Power Supply Startup Operation
Circuit Current in Operation
Normal Operation
FM/ELP Pin Voltage Difference
∆VFM
7-5
1.4
1.7
1.8
V
IFM(SRC)
7-5
–17.4
–13
–8.6
μA
FM/ELP Pin Sink Current
IFM(SNK)
7-5
8.6
13
17.4
μA
Average Switching Frequency
fOSC(AVG)
1-5
60
67
74
kHz
FM/ELP Pin Source Current
Frequency Modulation Deviation
∆f
1-5
4.8
6.9
9
kHz
Maximum Duty Cycle (On-duty)
DMAX
1-5
71
75
79
%
FB Pin Maximum Feedback Current
IFB(MAX)
6-5
–220
–160
–100
μA
Standby Operation Startup Voltage
VSTBY
6-5
0.99
1.10
1.21
V
Slope Compensation Startup Duty Cycle
DSLP
6-5
–
27
–
%
Slope Compensation Rate
SLP
6-5
–22
–17
–12
mV/μs
VOCP1
3-5
0.71
0.78
0.86
V
DPC
–
1.5
1.9
2.3
mV/D%
VOCP2
3-5
0.82
0.93
1.04
V
tBW
1-5
280
400
520
ns
Protection Operation
OCP Threshold Voltage at Zero Duty Cycle
(0% On-duty)
Drain Peak Current Compensation Coefficient
OCP Threshold Voltage After Compensation
LEB Time
tDLY
1-5
–
200
–
ms
Circuit Current in OLP-Operation
ICC(OLP)
4-5
–
410
700
μA
OVP Threshold Voltage
VCC(OVP)
4-5
27
28.5
30
V
Latch Circuit Holding Current
ICC(La.H)
4-5
–
140
220
μA
Latch Circuit Release Voltage
VCC(La.OFF)
4-5
6.4
7.1
7.8
V
ELP Threshold Voltage
VELP
7-5
6.4
7.1
7.8
V
Sink Current in ELP Operation
IELP
7-5
–
55
100
μA
TJ(TSD)
–
135
–
–
°C
OLP Delay Time*
Thermal Shutdown Activating Temperature
*Reference value of 47 nF capacitor between FM/ELP and GND Pins.
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
5
Electrical Characteristics of MOSFET Unless specifically noted, TA is 25°C
Characteristic
Symbol
Drain-to-Source Breakdown Voltage
VDSS
Drain Leakage Current
IDSS
Note
Pin
Min.
Typ.
Max.
Unit
1-3
650
–
–
V
1-3
–
–
300
μA
–
–
3.95
Ω
1-3
–
–
2.8
Ω
–
–
1.9
Ω
1-3
–
–
400
ns
–
–
2.23
°C/W
–
–
2.04
°C/W
–
–
1.75
°C/W
STR-W6251D
On-Resistance
RDS(ON)
Switching Time
tr
STR-W6252D
STR-W6253D
STR-W6251D
Thermal Resistance
Between
channel
and internal
frame
STR-W6252D
Rθch-F
STR-W6253D
–
Typical Application Circuit
Clamp Snubber Circuit
T1
VAC
C10
R9
V OUT
PC1
R3
P
R6
R4
D4
C1
L2
D3
S
U1
R7
C6
C5
R5
U2
C7
R8
GND
D1 R2
D/ST
S/OCP
VCC
GND
FB
FM/ELP
STR-W6200D
1
3 4 5 6 7
CV
ROCP
C2
D
External Latch Circuit
(Optional)
C4
PC1
C3
C8
Damper Snubber
The following design feature should be observed: In applications having a power supply specified such that VDS
has large transient surge voltages, a clamp snubber circuit of a capacitor-resistor-diode (CRD) combination should
be added on the primary winding, P, or a damper snubber circuit of a capacitor (C) or a resistor-capacitor (CR)
combination should be added between the D/ST pins and the S/OCP pin.
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
6
Functional Description
With regard to current direction, "+" indicates sink current
(toward the IC) and "–" indicates source current (from the IC).
Startup Operation
Figure 2 shows the VCC pin peripheral circuit. The built-in
startup circuit is connected to the D/ST pin, and it generates a
constant current, ISTARTUP (–1.6 mA typical) to charge capacitor C2 connected to the VCC pin. During this process, when the
VCC pin voltage reaches VCC(ON) (15.5 V typical), the control
circuit starts operation. After that, the startup circuit stops automatically, in order to eliminate its own power consumption.
The startup time is determined by the C2 capacitance, and a
value of 10 to 47 μF is generally recommended. The approximate startup time, tSTART , can be calculated as follows:
tSTARTUP (s) = C2 (µF) ×
VCC(ON) (V) – VCC(INT) (V)
|ISTARTUP |(mA)
where VCC(INT) is the initial voltage on the VCC pin.
C1
P
D1
VCC 4
R2
C2
STR-W6200D
The rectified voltage from the auxiliary winding, D (figure 2),
becomes a power source to the control circuit after the operation
start.
The VCC pin voltage should become as follows within the
specification of input voltage range and the output load range
of power supply, taking account of the winding turns of the D
winding; the target voltage of the VCC pin voltage is about
15 to 20 V:
VCC(OFF) = 9.8 V (max) < VCC
< VCC(OVP) = 27.0 V (min)
(2)
Figure 4 shows the VCC pin voltage behavior during the startup
period. When the VCC pin voltage reaches VCC(ON) , the control
circuit starts operation, the circuit current, ICC , increases, and
thus the VCC pin voltage begins dropping. At the same time,
the auxiliary winding voltage, VD , increases in proportion to the
output voltage rise. Thus, the VCC pin voltage is set by the balance between dropping by the increase of ICC and rising by the
increase of the auxiliary winding voltage, VD .
Just at the turning-off of the power MOSFET, a surge voltage
occurs at the output winding. If the feedback control is activated
by the surge voltage on light load condition at startup, and the
VCC pin voltage decreases to VCC(OFF) , a startup failure can
occur, because the output power is restricted and the output voltage decreases.
1
D/ST
(1)
Figure 3 shows the relationship of VCC and ICC. When the VCC
pin voltage increases to VCC(ON), the control circuit starts operation and the circuit current, ICC , increases. In operation, when
the VCC pin voltage decreases to VCC(OFF) (8.9 V typical), the
control circuit stops operation, by the UVLO (Undervoltage
Lockout) circuit, and reverts to the state before startup.
D
GND 5
In order to prevent this, when the VCC pin voltage falls to
the Bias Assist Voltage, VCC(BIAS) (15.2 V typical), the Bias
Figure 2. VCC pin peripheral circuit
VCC (V)
ICC
Bias Assist Period
Target Operation Voltage
1.4 mA
(Typ.)
15.5 (typ)
15.2 (typ)
Stop
Bias Assist Voltage
8.9 (typ)
Star t
Operation Stop Voltage
Startup Failure Event
5 μA
(Typ.)
8. 9 V
(Typ.)
Figure 3. VCC versus ICC
STRW6200D-AN Rev.2.0
13.8 V
(Typ.)
15. 5 V
(Typ.)
VCC
VIN(AC)
turn on
Start of normal
operation
time
Figure 4. VCC during startup period
SANKEN ELECTRIC CO., LTD.
7
Assist function is activated. While this function is operating,
the decrease of the VCC pin voltage is suppressed by providing the startup current, ISTARTUP , from the startup circuit. By
this function, the use of a small value C2 capacitor is allowed,
resulting in shortened startup time. Also, because the increase
of VCC pin voltage becomes faster when the output runs with
excess voltage, the response time of the OVP function can also
be shortened.
After the IC starts switching operation, the Bias Assist function
is available until the FM/ELP pin voltage reaches the FM/ELP
pin High Threshold Voltage, VFM(H) (4.5 V typical), and at this
voltage, this function stops.
In actual power supply circuits, there are cases in which the
VCC pin voltage fluctuates in proportion to the output of the
SMPS (see figure 5), and the Overvoltage Protection (OVP)
on the VCC pin may be activated. This happens because C2 is
charged to a peak voltage on the auxiliary winding D, which is
caused by the transient surge voltage coupled from the primary
winding when the power MOSFET turns off.
For alleviating C2 peak charging, it is effective to add some
value R2, of several tenths of ohms to several ohms, in series
with D1 (see figure 6). The optimal value of R2 should be determined using a transformer matching what will be used in the
actual application, because the variation of the auxiliary winding
voltage is affected by the transformer structural design.
Bobbin
Barrier
VCC
P1
S1
2
out R
With
P2
S2
D
Barrier
With R2
P1 ,P2
S1
IOUT
S2
D
Figure 5. VCC versus IOUT with and without resistor R2
Primary winding
Secondary output winding
controlled to constant voltage
Secondary output winding
Auxiliary winding for VCC
Figure 7. Winding structural example (a)
Bobbin
D1
4
VCC
STR-W6200D
Added
Barrier
R2
P1
D
C2
S1
D
S2
S1
P2
Barrier
GND
5
P1 ,P2
S1
S2
D
Figure 6. VCC pin peripheral circuit with R2
STRW6200D-AN Rev.2.0
Primary winding
Secondary output winding
controlled to constant voltage
Secondary output winding
Auxiliary winding for VCC
Figure 8. Winding structural example (b)
SANKEN ELECTRIC CO., LTD.
8
The variation of VCC pin voltage becomes worse if:
• The coupling between the primary and secondary windings of
the transformer gets worse and the surge voltage increases (low
output voltage, large current load specification, for example).
• The coupling of the auxiliary winding, D, and the secondary
side stabilization output winding (winding of the output line
which is controlling constant voltage) gets worse and it is subject
to surge voltage.
In order to reduce the influence of surge voltages on the VCC
pin, alternative structures of the auxiliary winding, D, can be
used; as examples of transformer structural designs see figures 7
and 8.
• Winding structural example (a): Separating the auxiliary winding D from the primary side windings P1 and P2.
The primary side winding is divided into two windings, P1 and
P2.
• Winding structural example (b): Placing the auxiliary winding D within the secondary winding S1 in order to improve the
coupling of those windings.
The output winding S1 is a stabilized output winding, controlled to constant voltage.
Frequency Modulation Function
The frequency modulation is superposed on the PWM frequency,
helping to reduce the conductive EMI noise, and simplify noise
filtering on input lines.
Figure 9 shows the VDS and ID waveforms at an average frequency of 67 kHz and an internally-fixed fluctuation width, Δf,
of 6.9 kHz typical. Figure 10 shows the relationship between the
FM/ELP pin voltage and the frequency modulation period.
The C3 connected to the FM/ELP pin is charged by the constant source current, IFM(SRC) (–13 μA typical), until its voltage increases to about 4.5 V. After that, it is discharged by the
Frequency modulation signal
FM/ELP
Voltage
2.8 V
(typ)
∆f = 6.9 kHz
VDS
4.5 V
(typ)
Drain
Current, ID
ID
f(min) ≈ 63.55 kHz
f(max) ≈ 70.45 kHz
tFM = (2 × C3 × 1.7) /13 μA (s)
Figure 9. VDS and ID waveforms in frequency modulation
Figure 10. FM/ELP pin voltage and frequency modulation period
Burst Oscillation Mode
Output Current, IOUT
Less than a few kilohertz
Drain Current, ID
Normal Load
Standby Load
Normal Load
Figure 11. Automatic-Standby mode operation
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
9
constant sink current, IFM(SNK) (13 μA typical), until its voltage
decreases to about 2.8 V, and then the mode is reverted to the
constant charge by IFM(SRC). The repetition of this makes the frequency modulation signal a triangle waveform on the FM/ELP
pin. By inputting this signal to the PWM oscillation circuit, the
frequency modulation occurs.
The frequency modulation period, tFM , can be adjusted by the
value of C3. The tFM can be calculated as follows.
tFM (s) = 2 × C3 × ∆VFM
13 (μA)
where ∆VFM is 1.7 V typical.
(3)
In general, a C3 value in the range of 0.01 to 0.047 μF is recommended, and should be determined based on actual operation in
the application.
VROCP
GND FB
5
6
RROCP
PC1
The operation mode becomes burst oscillation, as shown in figure 11. Burst oscillation reduces switching losses and improves
power supply efficiency because of periodic non-switching
intervals. Generally, to improve efficiency under light load conditions, the frequency of the burst oscillation becomes just a few
kilohertz. When the burst oscillation frequency is in the human
audible range (20 Hz to 20 kHz), audible noise may occur from
the transformer.
This IC keeps the peak drain current low during burst oscillation
mode, and suppresses the audible noise of the transformer.
Constant Output Voltage Control
The constant output voltage control function uses current-mode
control (peak current mode), which enhances response speed
and provides stable operation. This IC compares the voltage,
VROCP , of the current detection resistor with the target voltage,
VSC , by the internal FB comparator, and controls the peak value
of VROCP so that it gets close to VSC. VSC is internally generated
by inputting the FB pin voltage to the feedback control (see the
Functional Block diagram) and adding the slope compensation
value (refer to figures 12 and 13).
STR-W6200D
S/OCP
3
Automatic Standby Mode Function
The Automatic Standby mode is activated automatically when
the drain current, ID , reduces under light load conditions, at
which ID is less than 15% of the maximum drain current (it is in
the Overcurrent Protection state).
IFB
C4
• Light load conditions When load conditions become lighter,
the output voltage, VOUT, rises, and the feedback current from
the error amplifier on the secondary side also increases. The
feedback current is sunk at the FB pin, transferred through a
Figure 12. FB pin peripheral circuit
–
+
VSC
Target voltage without Slope Compensation
Target voltage including
Slope Compensation
VROCP
S/OCP signal
voltage across ROCP
FB Comparator
ton1
Drain
Current,
ID
T
Figure 13. Drain current, ID, and FB comparator operation in
steady operation
STRW6200D-AN Rev.2.0
ton2
T
T
Figure 14. Drain current, ID, waveform in subharmonic oscillation
SANKEN ELECTRIC CO., LTD.
10
photo-coupler, PC1, and the FB pin voltage decreases. Thus, VSC
decreases, the peak value of VROCP is controlled to be low, and
the peak drain current of ID decreases. This control prevents the
output voltage from increasing.
• Heavy load conditions When load conditions become greater,
the control circuit performs the inverse operation to that
described above. Thus, VSC increases and the peak drain current
of ID increases. This control prevents the output voltage from
decreasing.
In the current-mode control method, when the drain current
waveform becomes trapezoidal in continuous operating mode,
even if the peak current level set by the target voltage is constant, the on-time fluctuates based on the initial value of the
drain current. This results in the on-time fluctuating in multiples
of the fundamental operating frequency as shown in figure 14.
This is called the subharmonics phenomenon.
In order to avoid this, the IC incorporates the Slope Compensation function. Because the target voltage is added, a down-slope
compensation signal that reduces the peak drain current as the
on-duty gets wider relative to the FB pin signal to compensate
VSC , the subharmonics phenomenon is suppressed.
Even if subharmonic oscillations occur when the IC has some
excess supply being out of feedback control, such as during
startup and load shorted, this does not affect performance during
normal operation.
In the current-mode control method, the FB comparator and/or
the OCP comparator may respond to the surge voltage resulting
from the drain surge current in turning-on the power MOSFET,
and may turn off the power MOSFET irregularly. Leading Edge
Blanking, tBW (400 ns typical), is built-in to prevent malfunc-
Vcc
tions caused by surge voltage in turning-on the power MOSFET.
Fault Latch
When the OVP, ELP, and TSD functions are activated, the latch
circuit is also activated, and then the IC stops switching operation, in latch mode.
After that, the VCC pin voltage decreases to VCC(OFF) (8.9 V
typical), and then the startup circuit is activated. When the VCC
pin voltage increases to VCC(ON) (15.5 V typical), the circuit
current increases, and the VCC pin voltage decreases again. As a
result, the VCC pin voltage fluctuates between 8.9 V typical and
15.5 V typical as shown in figure 15, and it prevents the VCC
pin voltage from increasing.
Releasing the latched state is done by turning off the input voltage and allowing the VCC pin voltage to drop below VCC(La.OFF)
(7.1 V typical).
External Latch Protection Function (ELP)
This function forces a latched shutdown if more than the ELP
Threshold Voltage, VELP (7.1 V typical), is applied between
the FM/ELP and the GND pins. The applied voltage should
be within –0.3 to 12 V (the Absolute Maximum Rating of the
FM/ELP pin voltage).
The Typical Application Circuit shows an example, whereby a
current limitation resistor and a latch trigger switch (that is, a
photocoupler), are inserted between the VCC and FM/ELP pins.
Because the FM/ELP pin has a built-in Zener diode, the current
limitation resistor value should be such that the current going
into the FM/ELP pin would be below the Absolute Maximum
Rating of 3 mA for IFM .
8.9 V (typ)
Circuit current is small
Circuit current is large
7
C3
time
Figure 15. VCC pin voltage at fault latch
STRW6200D-AN Rev.2.0
5
FB
FM/ELP
15.5 V (typ)
GND
STR-W6200D
6
PC1
C4 IFB
Figure 16. FB pin and FM/ELP pin peripheral circuit
SANKEN ELECTRIC CO., LTD.
11
Overload Protection Function (OLP)
When the peak drain current of ID is limited by OCP operation,
the output voltage, VOUT , decreases and the feedback current
from the secondary photo-coupler, IFB (see figure 16), becomes
zero. When this state remains for the OLP Delay Time, tDLY , the
OLP function is activated, and the IC stops switching operation.
When the VCC pin voltage decreases to VCC(OFF) , the control
circuit stops its operation by the UVLO circuit, and reverts to the
state before startup. After that, the VCC pin voltage increases to
VCC(ON) because the startup circuit is activated. As a result, the
operation becomes the intermittent oscillation mode by UVLO
as shown in figure 17, during OLP operation.
This operation reduces stresses on the power MOSFET and on
the secondary rectifier diode, furthermore, it reduces power
consumption because it reduces the frequency of the intermittent
oscillation and the ratio of the switching period, by lowering the
circuit current to ICC(OLP) (410 μA typical) during OLP operation. The approximate value of tDLY is equal to 16 times the
charge-discharge cycle of C3, connected to the FM/ELP pin (see
figure 16), and its equation is:
tDLY (s) = tFM × 16
=
2 × C3 (µF) × 1.7 (V)
× 16
13 (µA) (typ)
(4)
where tFM is determined by equation 3.
The Delay Time, tDLY , should be longer than the period from a
point where the VCC pin voltage increases to VCC(ON) , to other
where the output voltage reaches its target value, in order to
prevent startup failure caused by OLP operation.
In general, a C3 value is recommended to be about
0.01 to 0.047 μF, and the optimal value should be determined
based on actual operation in the application.
Switching stopped
interval
VCC pin voltage
Swtiching
turns off
Overvoltage Protection Function (OVP)
When the voltage between the VCC pin and the GND pin
increases to VCC(OVP) (28.5 V typical) or more, the OVP function
is activated and stops switching operation. When the auxiliary
winding supplies the VCC pin voltage, the OVP function is able
to detect an excessive output voltage, such as when the detection
circuit for output control is open on the secondary side, because
the VCC pin voltage is proportional to the output voltage.
The secondary side output voltage, which initiates OVP operation, is calculated approximately as follows:
V
(normal operation)
× 28.5 (V) (typ.)
VOUT(OVP) = OUT
(5)
VCC(normal operation)
Overcurrent Protection Function (OCP)
The OCP function detects each peak drain current level of a
power MOSFET on a pulse-by-pulse basis, by monitoring the
voltage across the current detection resistor ROCP between the S/
OCP pin and the GND pin, and limits the output power accordingly. When the voltage drop on both sides of ROCP increases to
the OCP threshold voltage, the power MOSFET is turned off.
ICs with PWM control usually have some detection delay time
on OCP detection. The steeper the slope of the actual drain current at a high AC input voltage is, the later the actual detection
point is, compared to the internal OCP threshold voltage, VOCP.
Thus, the actual OCP point limiting the output current usually
has some variation depending on the AC input voltage, as shown
in figure 18.
The IC incorporates a built-in AC Input Compensation function
that superposes a signal with a defined slope into the detection
signal from the S/OCP pin as shown in figures 19 and 20. When
the AC input voltage is lower and the duty cycle is higher, the
OCP compensation level of this function is increased. Therefore,
the OCP point in low AC input voltage is increased to minimize
the difference of OCP points between low AC input voltage and
high AC input voltage, without additional external components.
VOUT
VCC(OFF)= 8.9 V typical
Variance resulting from
propagation delay
t
npu
Ci
A
t
Low
npu
Ci
A
h
Hig
FM/ELP pin voltage
Delay Time, tDLY
Drain Current, ID
Figure 17. Waveforms during OLP operation
STRW6200D-AN Rev.2.0
IOUT
Figure 18. Output current at OCP without input compensation
SANKEN ELECTRIC CO., LTD.
12
Because the compensation signal level is designed to depend
upon the duty cycle, the OCP threshold voltage after compensation, VOCP(D%) , is calculated as follows:
VOCP(D%) (V) = VOCP1 (V) + DPC (mV/duty%) × D (%)
where
VOCP1 is the OCP threshold voltage at zero duty cycle (V),
0.78Vtypical,
(6)
DPC is the OCP compensation coefficient (mV/D%),
1.9 mV/D% typical, and
D is the duty cycle.
Assuming an AC input voltage of 85 V, if the transformer is
designed so the duty cycle, D, at a maximum load is 50%, then,
according to equation 6, VOCP(50%) = 0.875 V typical.
Thermal Shutdown Function (TSD)
When the controller chip temperature increases to 135°C (min)
or more, the IC stops switching operation, in latch mode.
1.05
0. 95
Max.
Typ.
Min.
0.95
≈ 0.875
0. 85
VOCP2(D%) Typical (V)
VOCP(D%) (V)
1.00
≈ 0.923
0. 90
0.90
0.85
0.80
≈ 0.809
0. 80
0. 75
0. 70
0. 60
0. 50
0.75
0.70
0
10
20
30
40
50
60
70
80
0
0
Duty Cycle, D (%)
15. 5
50
75
Duty Cycle, D (%)
Figure 19. Duty Cycle versus OCP Threshold Voltage After
Compensation
STRW6200D-AN Rev.2.0
85 VAC
(as an example)
265 VAC
(as an example)
Figure 20. Duty Cycle versus typical value of OCP Threshold Voltage
After Compensation relative to input voltage, for examaple
SANKEN ELECTRIC CO., LTD.
13
Design Notes
Peripheral Components
Take care to use the proper rating and proper type of components.
• Input and output electrolytic capacitors
▫ Apply proper design margin to accommodate ripple current,
voltage, and temperature rise.
▫ Use of high ripple current and low impedance types, designed
for switch-mode power supplies, is recommended, depending
on their purposes.
• Transformer
▫ Apply proper design margin to core temperature rise by core
loss and copper loss.
▫ Because the switching circuits contain high frequency
currents, the skin effect may become a consideration.
▫ In consideration of the skin effect, choose a suitable wire
gauge in consideration of the rms current and a current
density of about 3 to 4 A/mm2.
▫ If measures to further reduce temperature are still necessary,
use paralleled wires or litz wires to increase the total surface
area of the wiring.
L2
D3
Frequency of Frequency Modulation versus C3 Value
10.00
C6
C5
Capacitance of External Capacitor at FM/ELP Pin
The capacitor C3 at the FM/ELP pin determines the frequency
of frequency modulation, fFM , and the OLP delay time tDLY . Figure 23 shows the relationship between them and the C3 value.
A C3 value of 0.01 to 0.047 μF is recommended, and should be
selected based on actual operation in the application.
R6
R4
S
Place C4 between the FB pin and the GND pin, as shown in
figure 22, to perform high frequency noise reduction and phase
compensation. The value for C4 is recommended to be about
2200 pF to 0.01 μF, and should be selected based on actual
operation in the application.
OUTPUT
PC1
R3
Phase Compensation
A typical phase compensation circuit with a secondary shunt
regulator (U2) is shown in figure 21. The value for C6 is recommended to be about 0.047 to 0.47 μF, and should be selected
based on actual operation in the application.
R7
C7
R5
U2
fFM (kHz)
T1
• Current detection resistor, ROCP
▫ A high frequency switching current flows to ROCP , and may
cause poor operation if a high inductance resistor is used.
▫ Choose a low inductance and high surge tolerant type.
R8
1.00
0..10
GND
0.01
0.001
Figure 21. Peripheral circuit around secondary shunt regulator (U2)
D1
C2
S/OCP
VCC
GND
FB
FM/ELP
D/ST
1
3 4 5 6 7
R1
PC1
STRW6200D-AN Rev.2.0
1000
D
0.1
OLP Delay Time versus C3 Value
100
10
1
0.001
C4
Figure 22. FB pin peripheral circuit
T1
tDLY (ms)
STR-W6200D
R2
0.01
C3 (μF)
C3
0.01
C3 (μF)
0.1
Figure 23. The relationship between tFM, tDLY , and C3 (calculated values)
SANKEN ELECTRIC CO., LTD.
14
Secondary Diode EMI Measure
A ceramic capacitor, CDI , parallel to the secondary rectifier as
shown in figure 24, may become necessary in some cases to
reduce EMI noise reduction. If ringing occurs on the drain current, it is recommended to connect a damper resistor, RDI , in
series with CDI , as shown in figure 25, in order to reduce ringing
waveforms, and to stabilize switching operation. Note: The values chosen for RDI and CDI should take those temperature rises
into consideration, based on actual operation in the application.
PCB Trace Layout and Component Placement
PCB circuit trace design and component layout significantly
affect operation, EMI noise, and power dissipation. Therefore,
pay extra attention to these designs. In general, where high frequency current traces form a loop, as shown in figure 26, wide,
short traces, and small circuit loops are important to reduce line
impedance. In addition, earth ground traces affect radiated EMI
noise, and the same measures should be taken into account.
Switch-mode power supplies consist of current traces with high
frequency and high voltage, and thus trace design and compo-
nent layouts should be done to comply with all safety guidelines.
Furthermore, because the incorporated power MOSFET has a
positive thermal coefficient of RDS(ON) , consider it when preparing a thermal design.
Figure 27 shows a circuit layout design example for the IC
peripheral circuit and secondary smoothing circuit.
• S/OCP Trace Layout: S/OCP pin to ROCP to C1 to T1 (winding P) to D/ST pin. This is the main trace containing switching
currents, and thus it should be as wide and short as possible.
If C1 and the IC are distant from each other, an electrolytic
capacitor or film capacitor (about 0.1 μF and with proper voltage
rating) near the IC or the transformer is recommended to reduce
impedance of the high frequency current loop.
• GND Trace Layout: GND pin to C2 (negative pin) to T1
(winding D) to R2 to D1 to C2 (positive pin) to VCC pin. This
trace also must be as wide and short as possible. If C2 and the IC
are distant from each other, placing a capacitor (approximately
0.1 to 1.0 μF film capacitor) close to the VCC pin and the GND
pin is recommended.
CDI
T1
D3
ID
P
C2
S
C5
D1 R2
D
Figure 24. Rectifier measure example
RDI CDI
T1
D3
ID
P
C2
S
C5
Figure 26. High-frequency current loops (hatched areas)
D1 R2
D
Figure 25. Damper resistor example
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
15
• ROCP Trace Layout: ROCP should be placed as close as possible
to the S/OCP pin. The connection between the power ground
of the main trace and the control circuit ground should be at a
single point ground (A in figure 27) to remove common impedance, and to avoid interference from switching currents to the
control circuit.
• Secondary Smoothing Circuit Trace Layout: T1 (winding S)
to D3 to C5. This trace should be as wide and short as possible.
If the loop distance is lengthy, leakage inductance resulting from
the long loop may increase surge voltage at turning off the incorporated power MOSFET. Proper secondary trace layout helps to
increase margin against the power MOSFET breakdown voltage,
and reduces stress on the clamp snubber circuit and losses in it.
T1
C10
D3
R9
P
D4
C1
S
C5
D1 R2
D/ST
S/OCP
VCC
GND
FB
FM/ELP
U1
STR-W6200D
1
3 4 5 6 7
CV
C2
D
Main power circuit trace
GND trace for the IC
C4
ROCP
PC1
A
C3
C8
Figure 27. Peripheral circuit example around the IC
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
16
• The contents in this document are subject to changes, for improvement and other purposes, without notice. Make sure that this is the
latest revision of the document before use.
• Application and operation examples described in this document are quoted for the sole purpose of reference for the use of the products herein and Sanken can assume no responsibility for any infringement of industrial property rights, intellectual property rights or
any other rights of Sanken or any third party which may result from its use.
• Although Sanken undertakes to enhance the quality and reliability of its products, the occurrence of failure and defect of semiconductor products at a certain rate is inevitable. Users of Sanken products are requested to take, at their own risk, preventative measures
including safety design of the equipment or systems against any possible injury, death, fires or damages to the society due to device
failure or malfunction.
• Sanken products listed in this document are designed and intended for the use as components in general purpose electronic equipment or apparatus (home appliances, office equipment, telecommunication equipment, measuring equipment, etc.).
When considering the use of Sanken products in the applications where higher reliability is required (transportation equipment and
its control systems, traffic signal control systems or equipment, fire/crime alarm systems, various safety devices, etc.), and whenever
long life expectancy is required even in general purpose electronic equipment or apparatus, please contact your nearest Sanken sales
representative to discuss, prior to the use of the products herein.
The use of Sanken products without the written consent of Sanken in the applications where extremely high reliability is required
(aerospace equipment, nuclear power control systems, life support systems, etc.) is strictly prohibited.
• In the case that you use Sanken products or design your products by using Sanken products, the reliability largely depends on the
degree of derating to be made to the rated values. Derating may be interpreted as a case that an operation range is set by derating the
load from each rated value or surge voltage or noise is considered for derating in order to assure or improve the reliability. In general,
derating factors include electric stresses such as electric voltage, electric current, electric power etc., environmental stresses such
as ambient temperature, humidity etc. and thermal stress caused due to self-heating of semiconductor products. For these stresses,
instantaneous values, maximum values and minimum values must be taken into consideration.
In addition, it should be noted that since power devices or IC's including power devices have large self-heating value, the degree of
derating of junction temperature affects the reliability significantly.
• When using the products specified herein by either (i) combining other products or materials therewith or (ii) physically, chemically
or otherwise processing or treating the products, please duly consider all possible risks that may result from all such uses in advance
and proceed therewith at your own responsibility.
• Anti radioactive ray design is not considered for the products listed herein.
• Sanken assumes no responsibility for any troubles, such as dropping products caused during transportation out of Sanken's distribution network.
• The contents in this document must not be transcribed or copied without Sanken's written consent.
STRW6200D-AN Rev.2.0
SANKEN ELECTRIC CO., LTD.
17
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