Design guide for ICE1PCS01/02(G)

Application Note, V1.3, April 2007
ICE1PCS01/02
Boost Type CCM PFC Design with
ICE1PCS01/02
Power Management & Supply
N e v e r
s t o p
t h i n k i n g .
Edition 2007-04-11
Published by Infineon Technologies Asia Pacific,
168 Kallang Way,
349253 Singapore, Singapore
© Infineon Technologies AP 2005.
All Rights Reserved.
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ICE1PCS02
Revision History:
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2007-04
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Subjects (major changes since last revision)
Boost Type CCM PFC Design with ICE1PCS01/02
License to Infineon Technologies Asia Pacific Pte Ltd
Luo Junyang
Liu Jianwei
Jeoh Meng Kiat
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V1.3
ICE1PCS01
Table of Contents
Page
1
Introduction ...................................................................................................................................5
2
2.1
2.2
2.3
2.4
2.5
2.6
2.7
2.8
2.9
2.10
2.11
2.12
2.13
Boost PFC design with ICE1PCS01/02 .......................................................................................7
Target specification .........................................................................................................................7
Bridge rectifier .................................................................................................................................7
Power MOSFET and Gate Drive Circuit .........................................................................................7
Boost Diode.....................................................................................................................................8
Boost inductor .................................................................................................................................9
AC line current filter.......................................................................................................................10
Boost Output Bulk Capacitance ....................................................................................................11
Current Sense Resistor.................................................................................................................11
Output voltage sensing divider......................................................................................................12
Frequency setting (only for ICE1PCS01)......................................................................................12
AC Brown-out Shutdown (only for ICE1PCS02) ...........................................................................13
IC supply .......................................................................................................................................15
Voltage loop and current loop compensation................................................................................15
Application Note
4
2007-04-11
Abstract
Continuous conduction mode (CCM) PFC controllers, named ICE1PCS01/02, are developed based on a
new control scheme. Compared to the conventional PFC solution, the new ICs does not need the direct sinewave sensing reference signal from the AC mains. Average current control is implemented to achieve the
unity power factor. In this application note, the design process for the boost PFC with ICE1PCS01/02 is
presented and the design details for a 300W output power PFC with the universal input voltage range of
85~265VAC are included.
1
Introduction
The Pin layout of ICE1PCS01 and ICE1PCS02 is shown in Figure 1.
GND
1
8
GATE
ICOMP
2
7
VCC
ISENSE
3
6
VSENSE
FREQ
4
5
VCOMP
ICE1PCS01
Figure 1
GND
1
8
GATE
ICOMP
2
7
VCC
ISENSE
3
6
VSENSE
VINS
4
5
VCOMP
ICE1PCS02
Pin Layout of ICE1PCS01 and ICE1PCS02
From the layout, it can be seen that most of Pins in ICE1PCS02 are the same as ICE1PCS01 except Pin 4.
In ICE1PCS01, Pin 4 is to set the switching frequency. However, for ICE1PCS02, Pin 4 is for AC brown out
detection and the switching frequency is fixed by internal oscillator at 65kHz. The typical application circuits
of ICE1PCS01 and ICE1PCS02 are shown in Figure 2 and Figure 3 respectively.
Application Note
5
2007-04-11
Rectifier
EMI Filter
D1
L1
COUT
R1
VOUT =400VDC
T1
R2
RSENSE
VIN=85V ...265V AC
R3
ISENSE
Auxiliary Supply
GATE
ICE1PCS01
VCC
ICOMP
VCOMP
FREQ
R4
C2
RFREQ
C1
Figure 2
GND VSENSE
C3
Typical application circuit of ICE1PCS01
Rectifier
EMI Filter
D1
L1
COUT
R1
VOUT =400VDC
T1
D2
R3
ISENSE
GATE
R5
R6
Figure 3
GND VSENSE
ICE1PCS02
VINS
Application Note
R2
RSENSE
VIN=85V ...265V AC
ICOMP
C1
C4
Auxiliary Supply
VCC
VCOMP
R4
C2
C3
Typical application circuit of ICE1PCS02
6
2007-04-11
2
Boost PFC design with ICE1PCS01/02
2.1
Target specification
The fundamental electrical data of the circuit are the input voltage range Vin, the output power Pout, the
output voltage Vout, the operating switching frequency fSW and the value of the high frequency ripple of the
AC line current Iripple. Table 1 shows the relevant values for the system calculated in this Application Note.
The efficiency at rated output power Pout is estimated to 90 % over the complete input voltage range.
Input voltage
85VAC~265VAC
Input frequency
50Hz
Output voltage and current
390VDC, 0.76A
Output power
300W
Efficiency
>90% at full load
Switching Frequency
65kHz
Maximum Ambient temperature around PFC
70ºC
Table 1 Design parameter for the proposed design
2.2
Bridge rectifier
In order to obtain 300W output power at 85 V minimum AC input voltage, the maximum input RMS current is
I in _ RMS =
Pout
Vin _ min ⋅ η
=
300
= 3.92 A
85 ⋅ 90%
and the sinusoidal peak value of AC current is
I in _ pk = 2 ⋅ I in _ RMS = 2 ⋅ 3.92 = 5.54 A
For these values a bridge rectifier with an average current capability of 6A or higher is a good choice. Please
note here, that due to a power dissipation of approximately
PBR = 2 ⋅ VF ⋅ I in _ RMS = 2 ⋅ 1V ⋅ 3.92 A = 7.84W
the rectifier bridge should be connected to an appropriate heatsink. Assuming a maximum junction
temperature TJmax of 125°C, a maximum ambient temperature TAmax of 70°C, the thermal junction-to-case
RthJC of approximate 2.5 K/W and the thermal case to heatsink RthCHS of approximate 1K/W, the heatsink
must have a maximum thermal resistance of
RthHS _ BR =
TJ max − T A max
125 − 70
− RthJC − RthCHS =
− 2.5 − 1 = 3.52 K / W
PBR
7.84
2.3
Power MOSFET and Gate Drive Circuit
Due to the switch mode operation, the losses are only active during the on-time of the MOSFET. The duty
cycle of the transistor in boost converters operating in CCM at minimum AC input RMS voltage is
Don = 1 −
Vin _ min
Vout
Application Note
= 1−
85
= 0.782
390
7
2007-04-11
Since rms-values have the same effect on a system as DC-values, it is possible to calculate a characteristic
duty cycle for the rms-value. Therefore, the on-state losses of the MOSFET in CCM-mode at a junctiontemperature of 125°C are
2
Pcond = I in _ RMS ⋅ Don ⋅ Rdson (125C )
the MOSFET switching loss can be estimated as
PSW = ( E on + E off ) ⋅ f SW
where, Eon and Eoff are the switch-on and switch-off energy loss which data can be found in MOSFET
datasheet, fSW is the switching frequency.
For 300W design, if SPP20N60C3 is used, the conduction loss is
Pcond = 3.92 2 ⋅ 0.782 ⋅ 0.42 = 5.05W
assuming the switching current is about 6A and gate drive resistance Rg=3.6Ω, then the switching loss is
PSW = (0.007mWs + 0.015mWs) * 65kHz = 1.43W
the total loss is
PMOS _ total = Pcond + PSW = 6.48W
the required heatsink for the MOSFET is
RthHS _ MOS =
TJ max − T A max
125 − 70
− RthJC _ MOS − RthCHS =
− 0.6 − 1 = 6.89 K / W
PMOS _ total
6.48
the gate drive resistance is used to drive MOSFET as fast as possible but also keep dv/dt within EMI
specification. In this 300W example, 3.6Ω gate resistor is chosen for SPP20N60C3 MOSFET.
Beside gate drive resistance, one 10kΩ resistor is also commonly connected between MOSFET gate and
source to discharge gate capacitor.
2.4
Boost Diode
The boost diode D1 has a big influence on the system’s performance due to the reverse recovery behaviour.
So the Ultra-fast diode with very low trr and Qrr is necessary to reduce the switching loss. The new diode
technology of silicon carbide (SiC) Schottky shows its outstanding performance with almost no reverse
recovery behaviour. The switching loss due to the boost diode can be ignored with choosing SiC Schottky
diode. Only conduction loss is calculated as below.
Pdiode = VF ⋅ I in _ RMS ⋅ (1 − Don ) = 2V ⋅ 3.92 A ⋅ (1 − 0.782) = 1.71W
For a rule of thumb, the SiC diodes provide a output power Pout of a CCM-PFC-system of 100 W to120 W
per rated ampere of the diode. This means for example, that the SDT04S60 from Infineon Technologies
which is rated for a forward current IF = 4 A is capable for a system of Pout = 4*100 W = 400 W system in
minimum. Therefore, this diode would be suitable for the proposed design.
The required heatsink for boost diode is
RthHS _ diode =
TJ max − T A max
125 − 70
− RthJC _ diode − RthCHS =
− 4.1 − 1 = 27.06 K / W
Pdiode
1.71
Application Note
8
2007-04-11
The SiC boost diodes often have a poor surge current capability, so that they may break down. Therefore a
so called bypass diode is necessary such as the diode D3 as Figure 4. For the proposed system, 1N5408 is
suitable.
D3
Rectifier
D1
L1
R1
COUT
T1
R2
RSENSE
Figure 4
2.5
inrush current bypass diode
Boost inductor
The peak current that the inductor must carry is the peak line current at the lowest input voltage plus the high
frequency ripple current. The high frequency ripple current peak to peak, IHF, can be related to maximum
input power and minmum input voltage as equation below.
I HF = k ⋅ 2 ⋅
Pin _ max
Vin _ min
Where, k must be kept reasonably small, and is usually optimized in the range of 15% to 25% for cost
effective design based on the current magnetic component status. If k is too high, the larger AC input filter is
required to filter out this ripple noise. If k is too low, the value of the inductance is too large and leads to big
size of the magnetic core.
For example, we choose k 22%. Then,
I HF = 22% ⋅ ⋅ 2 ⋅
Pin _ max
Vin _ min
= 1 .2 A
The peak current passing through inductor is
I L _ pk = I in _ peak +
I HF
1 .2
= 5.54 +
= 6.14 A
2
2
The boost inductance must be
Lboost ≥
D ⋅ (1 − D ) ⋅ Vout
I HF ⋅ f SW
D=0.5 will generate the maximum value for the above equation.
Lboost ≥
0.5 ⋅ (1 − 0.5) ⋅ 390V
= 1.25mH
1.2 A ⋅ 65kHz
The magnetic core of the boost choke can be either magnetic powder or ferrite material.
(1) sendust powder toroid core
The required effective magnetic volume of the core, Ve, is
Application Note
9
2007-04-11
Ve ≥ µ r µ 0 Lboost (
I L _ pk
Bmax
) 2 = 125 ⋅ 1.257 e − 6 ⋅ 1.25mH (
6.14 A 2
) = 11.6e − 6m 3 = 11.6cm 3
0.8T
where, µr is the relative permeability which is fixed by core manufacturer; µ0 is magnetic field constant
which is equal to 1.257e-6; Bmax is the maximum magnetic flux density for the selected magnetic material
(for sendust, Bmax is up to 0.8T.) Select a core with similar Ve value from the magnetic core datasheet.
For example, the core type CS468125 from Chang Sung Corporation is suitable for this case. The
parameters of CS468125 are Ve=15.584cm3, Ae=1.34cm2, C=11.63cm, µr=125. The turn number of the
boost choke winding is
Lboost ⋅ C
µ r µ 0 Ae
N toroid _ boost =
where, C is the magnetic path length and Ae is the effective magnetic cross section area.
The copper loss of the winding wire can be calculated on Iin_RMS.
2
PL _ boost = I in _ RMS ⋅ RL _ boost
Selecting the proper wire type to fullfil the loss and thermal requirement for the choke.
(2) ferrite core
To make sure the ferrite core will not go into saturation, the turn number of the boost choke winding with
ferrite core is
N ferrite _ boost ≥
I L _ pk ⋅ Lboost
Bmax ⋅ Amin
where, Bmax is up to 0.3T according to ferrite material specification; Amin is the minimum magnetic cross
section area.
The winding wire copper loss calculation is the same as in the above section of sendust powder toroid
core.
2.6
AC line current filter
As decribed in section 2.5, there is high frequency ripple current peak to peak IHF passing through boost
choke. This ripple will also go into AC line power network. The current filter is necessary to reduce the
amplitude of high frequency current component. The filtering circuit consists of a capacitor and an inductor
as shown in Figure 5.
Rectifier
IHF_spec Current Filter
IHF
Lfilter Cfilter
VIN=85V ...265VAC
Figure 5
AC line current filter
The required Lfilter is
Application Note
10
2007-04-11
I HF
L filter ≥
I HF _ spec
+1
(2πf SW ) 2 C filter
normally there is one EMI X2 capacitor which can act as Cfilter. In this example, if we define IHF_spec as 0.2A
peak to peak and asumming X2 capacitance 0.47µF, then
L filter
1.2 A
+1
0
.
2
A
≥
= 89 µH
(2π ⋅ 65kHz ) 2 ⋅ 0.47 µF
The leakage inductance of EMI common mode choke can be used for current filter. If the leakage inductance
is large enough, no need to add the additional differential mode inductor for filtering. Otherwise, a current
filter choke is necessary. The calculation method for the current filter choke is the same as for boost choke.
2.7
Boost Output Bulk Capacitance
The bulk capacitance has to fullfil two requirements, output double line frequency ripple and holdup time.
(1) output double line frequency ripple limit.
The inherent PFC always presents 2*fL ripple. The amplitude of ripple voltage is dependant on output
current and bulk capacitance as below.
C out ≥
I out
π ⋅ 2 * f L ⋅ Vout _ ripple _ pp
where, Iout is the PFC output current, Vout_ripple_pp is the output voltage ripple (peak to peak), and fL is the
AC line frequency.
Please note that ICE1PCS01/02 has enhance dynamic block which is active when Vout exceed ±5% of
regulated level. The enchanc dynamic block should be designed to work only during load or line change.
During steady state with constant load, the enhance dynamic block should not be triggered, otherwise
THD will be deteriorated. That means the target Vout_ripple_pp must be lower than 10% of Vout. For this
example, Vout=390VDC, then Vout_ripple_pp must be lower than 39V. if we define Vout_ripple_pp=8V, then
C out ≥
I out
= 306µF
π ⋅ 2 * f L ⋅ Vout _ ripple _ pp
(2) holdup time requirement
After the PFC stage, there is commonly a PWM stage to provide isolated DC output for end user. Some
applications, especially computing, have the holdup time requirement. It means that PWM stage should
be able to provide the isolated output even if AC input voltage become zero for a short holdup time. The
common specification for this holdup time is 20ms. If minimum input voltage for PWM stage is defined as
250VDC, then the bulk capacitance will be
C out ≥
2 ⋅ Pout ⋅ t holdup
2
Vout − Vout _ min
2
=
2 ⋅ 300W ⋅ 20ms
= 134 µF
390 2 − 250 2
the final Cout capacitance should be higher value calculated from the above two requirement.
2.8
Current Sense Resistor
Application Note
11
2007-04-11
The current sense resistance is calculated based on the IC soft over current control threshold and peak
current carried by boost choke.
When the Isense signal reaches the soft over control threshold, IC will reduce the internal control voltage and
accordingly the duty cycle is reduced in the following cycles. Finally the boost choke current is limited.
According to IC datasheet, soft over current control threshold is -0.66V maximum. So the current sense
resistor should be
Rsense ≤
0.66V 0.66V
=
= 0.11Ω
6.14 A
I L _ pk
According to Figure 2 and Figure 3, the transistor current as well as the diode current is sensed with Rsense.
That means that also the inrush current is sensed there leading to a large negative voltage drop at Rsense,
because the inrush current is in the range of about 150 A to 200 A. It is therefore necessary to limit the
current into Pin 2 (ISENSE) to 1 mA, which is realized with resistor R3. A value of R3 = 220Ω is sufficient for
this resistor.
2.9
Output voltage sensing divider
The output voltage is set with the voltage divider represented by R1 and R2 in Figure 2 and Figure 3. First,
choose the value of the lower resistor R2. Then the value of the upper resistor R1 is
R1 =
Vout − Vref
Vref
⋅ R2
where, Vref is IC internal reference voltage for voltage sensing, 5V typical.
If R2=10kΩ,
R1 =
390 − 5
⋅ 10kΩ = 770kΩ
5
It is recommended to take resistor values with a tolerance of 1% for R1 and R2. Due to the voltage stress of
R1, it is recommended to split this value into few resistors in series.
2.10
Frequency setting (only for ICE1PCS01)
The frequency of the ICE1PCS01 is adjustable in the range of 50 kHz up to 200 kHz. The external resistor
RFREQ according to Figure 2 programs a current which controls the oscillator. The given points of the resistorfrequency-characteristic are (250 kHz / 18 k.), (125 kHz / 33 k.) and (50 kHz / 82k.) and Figure 6 shows the
curve through those points.
Application Note
12
2007-04-11
450000
400000
350000
Freq /Hz
300000
250000
200000
150000
100000
50000
0
0
50000
100000
150000
200000
Resistance /ohm
Figure 6
2.11
Resistor-frequency characteristic
AC Brown-out Shutdown (only for ICE1PCS02)
Brown-out occurs when the input voltage VAC falls below the minimum input voltage of the design (i.e. 85V
for universal input voltage range) and the VCC has not entered into the VCCUVLO level yet. For a system
without input brown out protection (IBOP), the boost converter will increasingly draw a higher current from
the mains at a given output power which may exceed the maximum design values of the input current and
lead to over heat of MOSFET and boost diode. ICE1PCS02 provides a new IBOP feature whereby it senses
directly the input voltage for Input Brown-Out condition via an external resistor/capacitor/diode network as
shown in Figure 7. This network provides a filtered value of VIN which turns the IC on when the voltage at
Pin 4 (VINS) is more than 1.5V. The IC enters into the standby mode and gate is off when VINS goes below
0.8V. The hysteresis prevents the system to oscillate between normal and standby mode. Note also that
input voltage needs to at least 16% of the rated VOUT in order to overcome open loop protection and
powerup the system (referred to application note of ICE1PCS01).
Application Note
13
2007-04-11
VBR
D2
ICE1PCS02
R5
1.5V
C5
VINS
R
S
Protection
Logic
C4
0.8V
R6
Figure 7
C4
Block diagram of voltage loop
Because of the high input impedence of comparator of C4 and C5, R5 can be high ohmic resistance to
reduce the loss. From the datasheet, the bias current on VINS Pin is 1µA maximum. In order to have the
design consistence, the current passing through R5 and R6 has to be much higher than this bias current, for
example 7µA. Then R6 is:
R6 =
0.8V
= 114kΩ
7 µA
R6 is selected 120KΩ. R5 is selcted by
R5 =
2 ⋅ V AC _ on − 1.5V
1.5V
⋅ R6
where, VAC_on is the minimum AC input voltage (RMS) to start PFC, for example 70VAC.
R5 =
2 ⋅ 70V − 1.5V
⋅ 120kΩ = 7.8MΩ
1.5V
Due to the voltage stress of R5, it is recommended to split this value into few resistors in series.
C4 is used to modulate the ripple at the VINS pin. The timing diagram of VINS pin when IC enters brown-out
shutdown is shown in Figure 8.
Application Note
14
2007-04-11
VBR
2*VINS_AVE-0.8V
VINS_AVE
0.8V
Figure 8
tdischarge
Timing diagram of VINS Pin when IC enters brown-out shutdown
If the bottom level of the ripple voltage touches 0.8V, PFC is in standby mode and gate is off. The ripple
voltage defines PFC brown out off threshold of AC input voltage (RMS), VAC_off. C4 can be obtained from the
following equation. Assuming V INS _ AVE =
R6
⋅ V AC _ off , where, VAC_off is the maximum AC input voltage
R5 + R6
(RMS) to switch off PFC, for example 65VAC.
−
R6
(2 ⋅
⋅ V AC _ off − 0.8) ⋅ e
R5 + R6
t disch arg e
R6C4
= 0.8V
assuming tdischarge is equal to half cycle time of line frequency, ie. t disch arg e =
R6


2⋅
V AC _ off − 0.8V 

R5 + R6

C 4 =  2 f L R6 ln


0.8V




−1
120kΩ

2⋅
65V − 0.8V

7
.
8
M
120
k
Ω
+
Ω

C 4 = 2 ⋅ 50 Hz ⋅ 120kΩ ln
0.8V



2.12
1
, then
2 fL






−1
= 219nF
IC supply
Because of the low Vcc turn-on/off hysteresis, ICE1PCS01/02 is not supposed to be supplied by self supply
method with auxiliary winding on boost choke. The ICs can only be supplied by external DC applying to Vcc.
One Electrolyte capacitor 47uF and one ceramic capacitor 100nF is recommended to connect between Vcc
and GND to filter out the noise.
2.13
Voltage loop and current loop compensation
Application Note
15
2007-04-11
Please refer to Reference [5] for detail.
References
[1]
Infineon Technologies: ICE1PCS01 - Standalone Power Factor Correction Controller in Continuous
Conduction Mode; Preliminary datasheet; Infineon Technologies; Munich; Germany; Sept. 2002.
[2]
Luo Junyang, Jeoh Meng Kiat, Huang Heng Cheong, A New Continuous Conduction Mode PFC IC
With Average Current Mode Control, PEDS 2003; pp. 1110-1114, Nov 2003.
[3]
Luo Junyang, Jeoh Meng Kiat, Yew Ming Lik, 300W CCM PFC Evaluation Board with ICE1PCS01,
CoolMOS™ and SiC Diode thinQ!™, Application note, Infineon Technologies, Munich, Germany,
Oct. 2003.
[4]
Infineon Technologies: ICE1PCS02 - Standalone Power Factor Correction (PFC) Controller in
Continuous Conduction Mode (CCM) at Fixed Frequency, Preliminary datasheet; Infineon
Technologies; Munich; Germany; Sept. 2004.
[5]
Luo Junyang, Huang Heng Cheong, Jeoh Meng Kiat, ICE1PCS01 Based Boost Type CCM PFC
Design Guide - Control Loop Modeling, Application note, Infineon Technologies, Munich, Germany,
July, 2004.
Application Note
16
2007-04-11
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