Version 1.2 , Oct. 2003
Application Note
AN-PFC-TDA4862-1
TDA4862
TDA4862 - Technical Description
Authors:
Wolfgang Frank
Michael Herfurth
Published by Infineon Technologies AG
http://www.infineon.com/pfc
Power Management & Supply
N e v e r
s t o p
t h i n k i n g
TDA4862 - Technical Description
Contents:
Short Description ..................................................................................................................................... 3
Technical Description TDA4862 .............................................................................................................. 3
Control Method.............................................................................................................................. 3
Characteristics............................................................................................................................... 3
Power Supply and Self-Start............................................................................................... 3
Driver output........................................................................................................................ 4
Control amplifier .................................................................................................................. 4
Overvoltage control ............................................................................................................. 5
Multiplier.............................................................................................................................. 6
Current comparator ............................................................................................................. 6
Detector............................................................................................................................... 7
Applications of the TDA4862 ................................................................................................................... 7
Design steps.................................................................................................................................. 9
Input and output section...................................................................................................... 9
Multiplier section ............................................................................................................... 10
Boost inductor section....................................................................................................... 11
Operating frequency fp versus peak input voltage VinPk at constant output power Pout ............... 13
Output voltage controller: ............................................................................................................ 14
Zero Current Detector ................................................................................................................. 15
Auxilliary Power Supply............................................................................................................... 15
Summary of used Nomenclature ........................................................................................................... 26
References ............................................................................................................................................ 27
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TDA4862 - Technical Description
Short Description
The TDA 4862 integrated circuit controls a boost converter in a way that sinusoidal current is taken
from the single-phase line supply and stabilized DC voltage is available at the output. The circuit acts
as a harmonic filter which limits the harmonic currents resulting from the pulse charge currents of the
capacitor during rectification in a conventional capacitive input rectifier circuit. The power factor which
describes the ratio between active and apparent power is almost 1 and line voltage fluctuations are
compensated very efficiently, as well.
Technical Description TDA4862
Control Method
The control method of the harmonic filter is based on the physical relationship between current and
voltage at the boost converter choke. The transistor does not switch on until the current in the boost
converter diode turns zero. This creates triangular currents at a high frequency in the choke as it is
principally shown in figure 1, avoiding high-loss reverse recovery currents of the diode. If triangular
currents flow through the boost converter choke uninterruptedly, the mean input current calculated
over a high-frequency period is exactly half as high as the peak value of the high frequency choke
current. If the peak values of the choke current are on an envelope which is proportional to a
sinusoidal low-frequency input voltage, a sinusoidal
V, I
input current will be drawn from the mains after
VOUT
smoothing by means of an RFI suppression filter. The
RFI suppression filter is designed in a way that the valid
VIN
EMI limits at the inputs are not exceeded. Using this
control method, the operating frequency of the active
IL
IIN
harmonic filter changes with the input voltage and the
load.
t
Figure 1: Electrical input parameters and
Characteristics
choke current at discontinuous conduction mode operation
Power Supply and Self-Start
An undervoltage lockout with a turn-on threshold of 11
V typically and a turn-off threshold of 8.5 V typically assures that the IC is functional before the driver
output is enabled. In the stand-by state prior to enabling the driver the IC consumes a current of less
than 0.2 mA. A startup timer generates a set of pulses for the turn-off flip-flop, if the driver output stays
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in low state levels for longer than 150 µs. In order to guarantee safe supply from a current source the
supply voltage pin 8 is internally limited to 17 V to ground. Thus, the IC has all functions necessary for
low-loss self-start.
TDA 4862
2,5V Reference
VOLTAGE
AMPLIFIER
OUTPUT
Undervoltage Lockout
11V - 8.5V
Internal Supply
1.2V
C2
2
C3
Restart Timer
2.2V
1
0.9V
0
0
&
0
36k
0
0
1.3V
&
0
S
Q
R
Q
&
M2
2.5V...4.5V
M1
0V...4V
VCC
0
7
DRIVE
OUTPUT
6
GROUND
4
CURRENT
SENSE
0
0
0
3
8
16V
C4
MULTIPLIER
ZERO
CURRENT
DETECTOR
5V
2.5V
1.9V
VOP
VOLTAGE
SENSE
5
Multiplier
M3
C1
QM=M1*(M2-VFB)*K
Cm=0.65V-1
VFB=2.5V
30k
10p
1.3V
Figure 2: Scheme of TDA4862
Driver output
The driver output has been designed to drive power MOSFET with a current capability of ± 500 mA. In
order to avoid reverse currents the driver output is equipped with clamping diodes connected to
ground and supply voltage with a current rating of 100 mA. In standby state the driver output actively
asserts a LOW level with a residual voltage of 1.5 V and 5 mA dissipation current.
Control amplifier
The control amplifier compares the divided output voltage at its inverting input with a highly accurate
reference voltage of 2.5 V, with a maximum deviation of less than ±2% over the total temperature
range (–40°C < TJ < 150°C), at its non-inverting input. For the purpose of control loop compensation a
feedback network is inserted between the amplifier output (pin 2) and its inverting input (pin 1). A
feedback design using only one capacitor as an I-controller causes oscillating transient response,
because the boost converter, as a controlled current source, with the storage capacitor at its output
delays the phase by almost 90° in no-load and in low-load operation. The transient response is more
favorable if the control amplifier is designed as a PIT1-controller (see design steps).
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TDA4862 - Technical Description
The output voltage of the control amplifier ranges from 0.9 V to 4.3 V and can be loaded with a current
of 1 mA (source) and 2 mA (sink), respectively. The output voltage of the control amplifier is monitored
by a comparator. If the output voltage drops 0.3 V below the reference level of 2.5 V (i.e. reference
voltage) of the M2 multiplier input the driver output will be blocked directly via the turn-off flip-flop. This
measure guarantees the stability of the output voltage in complete no-load operation, without
interferences from offset voltages at the multiplier output or at the comparator input.
The output DC voltage of the boost converter is superimposed by double the mains frequency AC voltage ripple. The amplitude of the superimposed AC voltage depends on the capacity of the storage
capacitor and the load. The superimposed AC, which is also fed back via the control amplifier, causes
an undesirableed modulation of the line current drawn. Therefore the bandwidth of the controll
amplifier is chosen which is considerably lower than twice the mains-frequency. However, this causes
the controller to react slowly to sudden load changes which results in voltage overshoots and output
breakdowns.
Overvoltage control
If at the boost converter output a higher voltage than the rated output voltage is generated as a result
of voltage transients or load rejection, a current flows back from the output voltage divider to the
operational amplifier output via the feedback network. This is shown in figure 3. The current ∆I is
measured and in case of a threshold of 30 µA (typ.) is exceeded the multiplier output is controlled to
zero potential via a third input M3. This measure causes the input current to be continuously
+
C4
-
0.9V
V0
450V
400V
498V
400V
VREF
Vcc
RH
1600k
RH
1600k
M3
M2
OP
∆I
1.3V
V0
V0=400V
36k
+
-
I=250µA
Voltage
Sense
∆ I=0
2.5V
1
I=280µA
∆I
2.5V
=30µA
I=250µA
Threshold
∆I=30µA
RH
3200k
I=155µA
∆I
=30µA
I=250µA
2.5V
I=125µA
∆I
TDA 4862
RL
10k
RL
10k
RL
20k
2
Voltage Amplifier Output
Compensation
Network
Figure 3: Examples of the output voltage divider
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TDA4862 - Technical Description
compensated back, thus avoiding uncontrolled oscillations of the line current drawn, as they usually
appear with digital measures.
The switch-off level of the overvoltage control can be adjusted via the internal resistance of the output
voltage divider. In normal operation state the voltage at the tap of the divider is 2.5 V (i.e. reference
voltage). In case of higher than rated output voltage the excess divider current flows from the tap to
the operational amplifier output via the feedback network. The overvoltage control is also guaranteed
in the operational phases when the output voltage of the control amplifier reaches the upper limit
threshold, because the dissipation current is measured as well. As soon as the output voltage of the
control amplifier tends towards the minimum level, the comparator turns off at a level of 2.2 V to
guarantee safe no-load operation.
Multiplier
The multiplier generates the turn-off threshold of the current comparator giving consideration to the
waveshape of the feed voltage. In a typical application the rectified and divided supply voltage is
applied to the input M1 (pin 3). The output voltage of the control amplifier is applied at the input M2
which – under constant load and ideal conditions – appears as DC voltage without superimposed AC
shares. At the output of the multiplier a signal in the wave form of the rectified voltage corresponding
to input M1 is generated which can be modified in its amplitude via the DC voltage at input M2. Superimposed AC voltage shares at the input M2 cause an undesired modulation of the line current drawn,
unless they are part of the dynamic control processes. The level control range of the input M1 is 0 V to
4.0 V, the reference level being 0 V. The level control range of the input M2 is 2.5 V to 4.5 V, the
-1
reference level being 2.5 V. For multiplication a further, constant factor Cm = 0.65 V , which is an
-1
internal factor of the multiplier, is effective. Its dimension is V in order to comply with the following
equation. In this way the current comparator level can be calculated as VQm = Cm (Vpin2 - Vref) Vpin3.
The output voltage of the multiplier is limited to 1.3 V. This measure causes a defined turn-off
threshold for current limitation. In this way, dangerous excess currents are avoided which can arise in
particular in the case of an expanded input voltage range because the multiplier with its restricted
dynamics re-stabilized the current consumption.
Current comparator
The current comparator detects the voltage decline at the shunt which is in the source path of the
power MOSFET via its inverting input (pin 4) and which should have an intrinsic inductance as low as
possible. When switching on the transistor voltage, spikes are generated at the shunt as a result of the
intrinsic inductance of the shunt with turn on and the influence of the driver currents. An integrated
low-pass filter suppresses these voltage spikes. As soon as the voltage decline at the shunt reaches
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the turn-off threshold defined by the multiplier, the turn-off flip-flop is reset and the driver switches off.
The turn-off flip-flop prevents multiple pulses during the switching waveform of the power MOSFET.
The turn-off delay time between comparator input and driver output is below 250 ns.
Detector
The detector finds the point of time when the current in the boost converter choke turned zero and
then enables the control of a new pulse cycle. After the current comparator triggers the turn-off
process and the power MOSFET blocks, the boost converter diode takes over the current. In this case
the polarity of the voltage at the choke winding changes in a way that now a higher level voltage levell
(Vout) is available at the drainside terminal of the choke compared with the mains rectifier side terminall
(level Vin) of the choke. As soon as the choke current reaches zero and the boost converter diode
blocks, the voltage reverses at the drain side terminal of the choke. The voltage at the choke winding
turns zero or changes polarity. A second winding (detector winding) on the choke, which has
approximately 1/5 of the number of turns compared with the mains winding, permits the change of
polarity of the choke voltage to be registered without detrimental influences. Evaluation is effected by
the detector function (pin 5) of the IC, with the drain side polarity of the detector winding being
measured by means of a hysteresis-determined comparator.
The level for the acceptance of the „MOSFET blocks“ command from the turn-off flip-flop and for
setting the flip-flop is 2.5 V (i.e. reference voltage) with rising voltage. In case of a voltage decline,
which signals the zero crossing of the current, the switching level enabling the driver stage is 1.9 V.
The voltage of the detector winding is applied to pin 5 via a high-ohmic resistance (10k to 47k).
Clamping structures are available in the IC which limit the voltage at the input to +5 V and +0.6 V,
respectively, at 10 mA maximum.
There are cases in which there is no significant detector signal to set the turn-off flip-flop. This may be
the case when the supply voltage is switched on, in case of line overvoltage exceeding the output
voltage and in no-load and low-load operation, when the voltage controlller specifies intermittant
operation. In that case a startup generator is activated which supplies a set of pulses to the turn off
flip-flop if the driver output stays on LOW-level longer than 150 µs.
Applications of the TDA4862
The following applications demontrate the good performance of the TDA4862 controlling a power
factor preconverter. The design steps indicate the method of the calculation of the components values.
Lamp ballast designs as well as a design for switched mode power supplies (SMPS) are given here as
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examples. Circuit diagrams and measurement results at different operating conditions establish a
good basis for evaluation.
The tables of page 17 ff. also consists of a column called IZ which contains the values for the surplus
current of the auxiliary power supply for the IC bypassed with a 15 V zener diode. The zener current
indicates a suffucient IC supply. Therefore it should be low enough to avoid unnecessary losses.
There may be also states of operation when the zener current reaches zero. Then the actual supply
voltage VCC of the IC is figured.
Usually a single stage RFI-filter does not accomplish the RFI-standards. Therefore multiple stage RFIfilters are designed into these applications as an example how to suppress resonant oscillations of
these filters.
Discontinuous conduction mode always results in a high switching efficiency, because it avoids
reverse recovery losses of the boost converter diode. A high power factor, low harmonics, a wide input
voltage range and a feedback controlled output voltage are the most omportant features of a power
factor preconverter. The TDA4862 contains all control and monitoring functions to meet these
demands.
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Design steps
Input and output section
Application
2L-Ballast
1L-Ballast
3L-Ballast
SMPS
120V AC
230V AC
277 V AC
90 V – 270 V
Nominal input voltage
Vinnom
Minimum input voltage
Vinmin
= Vinnom – 20%
96V AC
184V AC
221 V AC
90 V AC
Maximum input voltage
Vinmax
= Vinnom + 20%
144V AC
276V AC
332 V AC
270 V AC
Minimum peak input voltage
VinPkmin
=√2 Vinnin
136V
260V
313 V
127 V
Maximum peak input voltage
VinPkmax
=√2 Vinmax
204V
390V
470 V
382 V
Estimated minimum efficiency
η
= 0.9
Output power
Pout
= η Pin
75W
53W
110 W
150 W
Maximum peak input current
IinPkmax
= 2 Pout / (Vinpmin η)
1.225A
0.453A
0.781A
2.625 A
Maximum high frequency peak current
IPkmaxHF
= 2 IinPkmax
2.45A
0.906A
1.562 A
5.25 A
Maximum current sense threshold
VISensemax
= 1.3 V
Shunt resistor
R11
= VISensem / IPkmaxHF
0.53Ω
1.44Ω
0.83 Ω
0.25 Ω
Nominal output voltage
Vout
Recommended minimum:VinPkmax+30V
230 V DC
410V
480V DC
410 V DC
Reference voltage
Vref
= 2.5 V
Controller current at pin VAOUT
IVAOUT
= 30 µA
Output voltage divider
R5
= Vref / IR5
10k
10k
10k
10k
(Select IR5 = 250 µA)
R4
= R5 (Vout—Vref) / Vref
910k
1640k
1910k
1640k
Overvoltage threshold
VOV
(Recommended: 1.1Vout)
257 V
462 V
537 V
462 V
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Multiplier section
Application
Multiplier inputs M1 and M2 dynamic
2-Lamp-Ballast
1L-Ballast
3L-Ballast
SMPS
136 V
260V
313V
127V
1M
2M
2M
940K
Vm1R = 3.8 V; Vm2R = 4.5V – Vref = 2V
voltage range
Multiplier output limitation
VQmmax
=VISensemax = 1.3V
Multiplier gain
Cm
= 0.65
Vm1(@ VQm = 1.3V; Vm2R = 2V)
=1.3V / (2V* Cm) =1V
From multiplier output characteristic
Vm1lim(@ VQm = 1.3V; Vm2R = 2V) = 1.2 V
Select Vm1 = Vm1lim = 1.2 V
Select upper resistor of input voltage
divider
@ VinPkmin =
R6
Lower resistor of input voltage divider
R7
=R6!Vm1lim / (VinPkmin – Vm1lim)
8.89k
9.27k
7.69k
8.95k
Low pass filter capacitor
C4
=1 / (2π!R2!f) {1 kHz<f<3kHz}
10 nF
10 nF
10 nF
10 nF
1.80V=OK
1.80V=OK
1.80V=OK
3.62V=OK
Test: Input range:
Vm1(@VinPkmax) < Vm1R = 3.8 V ?
otherwise select
Vm1(@VinPkmin) < Vm1lim = 1.2 V
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Boost inductor section
In this section two different approaches for the calculation of the transformer primary inductance LP are presented. The first one is recommended for a small input
voltage range application or for applications with nearly constant output power, e.g. lamp ballasts. Therefore only one example is executed here. The other one is
suitable for the demands of wide range applications like they occur in SMPS. All the values of the sections before are still valid.
2-Lamp-Ballast
SMPS-preconverter
On-time of power switch:
Ton = LP ! IPkmaxHF / Vin , IPkmaxHF = 2 IinPkmax
Off-time of power switch:
Toff = LP ! IPkmaxHF / (Vout - Vin)
Pulse frequency:
fp
=
Ton
1
+ Toff
=
Vin ⋅ (Vout − Vin )
Vout ⋅ LP ⋅ IPk max HF
Design criterion:
Design criterion:
Calculate LP according to desired range of pulse frequency (e.g. 80 kHz < fp Calculate LP in order to obtain pulse frequencies higher than 25 kHz at
<110 kHz) at nominal input voltage Vinnom and rated output power Pout
LP
=
=
Vinnom ⋅ (Vout − Vinnom )
Vout ⋅ f p ⋅ IPk max HF
=
Vinnom ⋅ (Vout − Vinnom ) ⋅ η ⋅ Vinnom
Vout ⋅ f p ⋅ 2Pout
120 V ⋅ (230 V − 120 V ) ⋅ 0,9 ⋅ 120 V
230 V ⋅ 90 kHz ⋅ 2 ⋅ 75 W
=
maximum peak input voltage and twice of nominal output power and on
minimum peak input voltage and twice of nominal output power
=
LP
459 µH
LP
Calculate LP by selecting the on-time Ton in the range of 3 µs < Ton < 6 µs
=
Ton ⋅ Vinnom
IPk max HF
=
2
Ton ⋅ V innom
⋅η
2 ⋅ Pout
Both inductances will be appropriate.
2
VinPk
(382V )2 ⋅ ( 410 V − 382 V ) ⋅ 0,9
max ⋅ (Vout − VinPk max ) ⋅ η
= 598 µH
=
Vout ⋅ fp ⋅ 2 ⋅ 2Pout
410 V ⋅ 25 kHz ⋅ 2 ⋅ 2 ⋅ 150 W
<
2
VinPk
(127V )2 ⋅ ( 410 V − 127 V ) ⋅ 0,9
min ⋅ (Vout − VinPk min ) ⋅ η
= 668 µH
=
Vout ⋅ f p ⋅ 2 ⋅ 2Pout
410 V ⋅ 25 kHz ⋅ 2 ⋅ 2 ⋅ 150 W
and
Also possible:
LP
<
=
5 µs ⋅ (120V 2 ) ⋅ 0,9
2 ⋅ 75 W
=
432 µH
We therefore select LP < 598 µH
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Application
Ballast, Vinnom = 120 V
Ballast, Vinnom = 230 V
Ballast, Vinnom = 277 V
SMPS, Vin = 90 V – 270 V
Example
POUT =75 W
L=
(120V )² ⋅ (230V − 120V ) ⋅ 0.9
230V ⋅ 90kHz ⋅ 2 ⋅ POUT
=
34,4mH ⋅ W
POUT
L=
(230V )² ⋅ ( 410V − 230V ) ⋅ 0,9
410V ⋅ 90kHz ⋅ 2 ⋅ POUT
=
116mH ⋅ W
POUT
POUT = 55 W
L=
(277V )² ⋅ ( 480V − 277V ) ⋅ 0,9
480V ⋅ 90kHz ⋅ 2 ⋅ POUT
=
162mH ⋅ W
POUT
POUT = 110W
L=
90mH ⋅ W
POUT
L = 459 µH
L = 2,1 mH
L = 1,47mH
POUT = 150W
L = 600µH
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TDA4862 - Technical Description
Operating frequency fp versus peak input voltage VinPk at constant output power Pout
f p (VinPk max ) =
V inPk max ⋅ (Vout − V inPk max ) V inPk max ⋅ (Vout − VinPk max ) ⋅ VinPk max V ² inPk max ⋅ (Vout − VinPk max ) ⋅ η
=
=
Vout ⋅ LP ⋅ 2 ⋅ I inPk max
Vout ⋅ LP ⋅ 2 ⋅ 2Pin
Vout ⋅ LP ⋅ 4 ⋅ Pout
Vin ⋅ 2 ⋅ sin ωt ⋅ (Vout − Vin ⋅ 2 sin ωt )
Operating frequency
fp(ωt) =
Example
f p (ωt ) =
Vout ⋅ LP ⋅ 2 ⋅ I in ⋅ 2 ⋅ sin ωt
=
Vin ⋅ (Vout − Vin ⋅ 2 sin ωt )
(Vin )² ⋅ η
=
(Vout − Vin ⋅ 2 ⋅ sin ωt )
Vout ⋅ LP ⋅ 2 ⋅ I in
Vout ⋅ LP ⋅ 2 ⋅ Pout
(120V )² ⋅ 0.9
⋅ (230V − 120V ⋅ 2 ⋅ sin ωt )
230V ⋅ 450 µH ⋅ 2 ⋅ 75W
Figure 4 shows the pulse frequency dependent on
the peak value of the input voltage for constant
1
0.9
output power or constant primary inductance
respectively. For example, the lower limit of the
0.8
input voltage in wide range applications is about
0.7
0.3 Vout. The corresponding pulse frequency is
f HFACP V INACP
then 40 % of the maximum pulse frequency. The
f MAX
upper limit in such applications is about 90 % of
the output voltage Vout, which leads to a pulse
frequncy of about 50 % of the maximal value.
0.6
0.5
0.4
0.3
0.2
0.1
0
0
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
V INACP
It is important, that those two frequencies
mentioned above are still above 25 kHz.
0.1
V OUT
Figure 4: Pulse frequency fp as a function of the input peak voltage
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Output voltage controller:
Usually a PIT1-design is used in PFC-circuits like it is shown in figure 8. The setting of the values of
C1, C2 and R1 should hit the following targets:
-
Good suppression of superimposed AC-share of the output voltage which has twice the
frequency of the input voltage,
-
good behaviour at load changes or changes of the input voltage,
-
good behaviour at low load conditions.
V OUT= 2 3 0 V
20dB
910k
2 .5V
10dB
G
C1
33Hz
3,3Hz
100Hz
f
1 0k
-10dB
Figure 5: Output voltage controller with integral component
VOUT=230V
2,5V
20dB
910k
10dB
33Hz
C1
C2
10k
R2
G
1,6Hz
f
3,3Hz
16Hz
100Hz
-10dB
Figure 6: Output voltage controller in PIT1-design
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NP
Zero Current Detector
The upper threshold of the ZCD is max. 2.75V. For
a)
NZCD
C10
a continuous operation the difference between out-
R12
put voltage Vout and maximum input voltage VinPkmax
and the transformation ratio of the inductor
D6
ZCD
Vcc
NP
windings have to accomplish the following relation
(Vout
b)
N
− VinPk max ) ⋅ ZCD > 2.75V
NP
D7
C10
D6
The recommended transformation ratio of NZCD/NP
C13
c)
frequency.
D7
C10
R9
NZCD
R12A
D6
Auxilliary Power Supply
R12
NP
the detector input voltage doesn’t achieve the upper
threshold, the IC is operating with the timer
NZCD
ZCD
Vcc
= 1/5 meets a minimal voltage difference of 14 V. If
R9
C13
Vcc
L5
R9
ZCD
An obvious way to supply the IC is to use the detector winding. We have to care, that the supply circuit
doesn’t influence the detector signal. First, in a
Figure 7: Auxiliary power supply realized with
rectifier (a) and charge pump (b and c)
simple voltage mode supply, we use a diode, a storage capacitor C10 and a current limiting resistor
R12. We achieve good results in ballast applications with the following design of the transformation
ratio:
NZCD
VZCD
=
NP
Vout − Vinnom
VZCD = 22V...24V;
R12 = 220Ω...270Ω
Second in a charge pump supply, we use two diodes, two capacitors C10, C13 and one decoupling
Resistor R12 or a decoupling inductor L5 (lower losses) and a current limiting resistor R12A, to avoid
burn down at resonance frequency. This method of supply is to prefer in SMPS applications with wide
input voltage range.
The supply current increases with the operating frequency at low load and is not dependend on the
input voltage. We achieve good results with the following design of the transformation ratio:
N ZCD VZCD
=
NP
Vout
VZCD ≈ 80V,
C13 = 3nF...4nF
R12 = 390Ω...270Ω
Or C13 = 1 nF...1.5nF, L5 = 50 µH...100µH, R12A designed with C13 and L5 as a low-pass filter of
Bessel characteristic.
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Applications
5 0 0 µH
L1
D1
R 15
C11
µ47
L1: 500µH
EF25, N27, gap=1mm
W1=75Wdg. 0,40d
W2=15Wdg. 0,22d
Q1: BUZ60 (400V; 1Ω)
100Ω
C6
µ47
R 12
D6
270Ω
R8
R6
1M
U
VIN
90-150V
AC
C5
3300p
M U R 115
R 10
3
R7
9k1
D3
Q1
5
R 14
C7
3300p
R9
33k
100k
C4
10n
8
D4
C9
µ1
C10
47µ/25V
TDA 4862
1A
1m H
L3
VOUT
D5
D2
S10K150
2x 10m H
L2
F use
230V DC
1N4937
6
7
1
R4
910k
12Ω
C2
1µ
C8
47µF
350V
R2
5k1
C1
1µ
2
4
R 11
0Ω5
R5
10k
GND
Figure 8: 75W Power Factor Preconverter with TDA 4862 and Vinnom = 120V
Page 16 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
Table 1: Measurement of input and output values
120V input for 2 x 35W lamp ballast
(Cout = 47µF, L1=500µH)
V RM S
I RM S
P IN
PF
THD
V OUT
I OUT
P OUT
V OUTAC
η
real power
I z (15V)
or V CC
93V
100V
120V
140V
150V
0.882A
0.812A
0.663A
0.563A
0.524A
82.20W
81.21W
79.55W
78.68W
78.44W
0.999
0.999
0.999
0.997
0.996
2.0%
2.5%
3.2%
4.3%
4.7%
229V
229V
229V
229V
229V
0.328A
0.328A
0.328A
0.328A
0.328A
75W
75W
75W
75W
75W
25V
25V
25V
25V
25V
0.912
0.924
0.934
0.953
0.956
9.0mA
8.0mA
3.0mA
1.0mA
0.4mA
90V
120V
140V
90V
120V
140V
120V
120V
0.392A
0.289A
0.249A
0.185A
0.141A
0.124A
0.081A
35.21W
34.45W
34.25W
16.54W
16.32W
16.24W
8.65W
0.998
0.993
0.984
0.991
0.965
0.933
0.890
3.0%
5.0%
6.5%
4.8%
6.8%
9.5%
9.4%
229V
229V
229V
229V
229V
229V
229V
0.124A
0.124A
0.124A
0.066A
0.066A
0.066A
0.033A
32.5W
32.5W
32.5W
15W
15W
15W
7.5W
13V
13V
13V
6V
6V
6.5V
3V
30V
0.923
0.943
0.949
0.907
0.919
0.924
0.867
5.6mA
0.3mA
11.9V
1.7mA
11.7V
9.6V
9.8V
Page 17 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
15 0 0 µH
L1
D1
VIN
180-270V
AC
C11
µ22
L1: 1500µH
EF20, N27, gap=1mm
W1=144 Wdg. 0,33d
W2=18 Wdg. 0,22d
Q1: BUZ77A (400V, 4Ω)
150Ω
C6
µ22
R 12
D6
R8
270Ω
R 8A
100k
100k
R6
1M
C7
3300p
Q1
R 10
3
R7
9k1
D3
R9
33k
5
R 14
C4
10n
8
D4
C9
µ1
C10
47µ/25V
TDA 4862
R 15
M U R 115
S10K250
0,5A
C5
3300p
U
3m H
L3
VOUT
D5
D2
R6A
1M
2x 18m H
L2
F use
410V DC
M U R 1 1 00
6
7
1
R4
820k
R4
820k
12Ω
C2
1µ
C8
10µF
450V
R2
5k1
C1
1µ
2
4
R 11
1Ω5
R5
10k
GND
Figure 9: 53W Power Factor Preconverter with TDA 4862 and Vinnom = 230V Input
Page 18 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
Table 2: Measurement of input and output values
230V input for 50W lamp ballast
(Cout = 10µF, L1=1.5mH)
V RM S
I RM S
P IN
PF
THD
V OUT
I OUT
P OUT
V OUTAC
η
I z (15V)
od.V CC
real power
180V
200V
230V
250V
270V
0.317A
0.282A
0.245A
0.225A
0.209A
57.16W
56.38W
56.02W
55.76W
55.61W
0.998
0.997
0.993
0.989
0.984
3.0%
2.5%
4.0%
5.3%
6.3%
409V
409V
409V
409V
409V
0.130A
0.130A
0.130A
0.130A
0.130A
53W
53W
53W
53W
53W
30V
30V
30V
30V
30V
0.927
0.940
0.946
0.950
0.953
9.9mA
6.9mA
4.0mA
2.8mA
1.9mA
180V
230V
270V
180V
230
270V
230V
0.146A
0.130A
0.113A
0.075A
0.063A
0.061A
29.36W
28.95W
28.8W
12.68W
12.63W
12.60W
0.991
0.970
0.941
0.944
0.865
0.764
4.3%
7.8%
10.7%
9.8%
12%
19%
409V
409V
409V
409V
409V
409V
409V
0.066A
0.066A
0.066A
0.027A
0.027A
0.027A
27W
27W
27W
11W
11W
11W
0
16V
16V
16V
8V
8V
8V
50V
0.920
0.933
0.937
0.868
0.871
0.871
0.871
7.6mA
1.8mA
0.2mA
4.0mA
12.8V
12.0V
10.8V
Page 19 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
15 0 0 µH
L1
D1
VIN
220-330V
AC
C11
µ22
150Ω
L1: 1500µH
EF25, N27, gap=2mm
W1=158 Wdg. 0,38d
W2=17 Wdg. 0,22d
Q1: BUZ80A (800V, 3Ω)
C6
µ22
R 12
D6
270Ω
R 8A
R6
1M
R 14
5
7
C7
3300p
Q1
R 10
3
R7
8k2
D3
R9
33k
120k
C4
10n
8
D4
C9
µ1
C10
47µ/25V
TDA 4862
R 15
M U R 115
S10K300
0,8A
C5
3300p
U
3m H
L3
VOUT
D5
D2
R6A
1M
2x 18m H
L2
F use
480V DC
M U R 1 1 00
6
1
R 4A
910k
R4
1M
12Ω
C2
1µ
C8
47µF
350V
R2
5k1
C1
1µ
2
4
R 11
0Ω8
R5
10k
GND
Figure 10: 110W Power Factor Preconverter with TDA 4862 and Vinnom = 277V Input
Page 20 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
Table 3: Measurement of input and output values
277V input for 3 x 35W lamp ballast
(Cout = 22µF, L1=1.5mH)
V RMS
I RMS
P IN
PF
THD
V OUT
I OUT
P OUT
V OUTAC
η
I z (15V)
or V CC
real power
220V
250V
277V
300V
0.527A
0.461A
0.415A
0.382A
115.8W
115.1W
114.6W
114.2W
0.999
0.998
0.996
0.994
2.7%
3.8%
4.5%
5.2%
479V
479V
479V
479V
0.229A
0.229A
0.229A
0.229A
110W
110W
110W
110W
35V
35V
35V
35V
0.950
0.956
0.960
0.963
7mA
3.7mA
2.1mA
1.1mA
220V
277V
300V
220V
277V
300V
0.396A
0.284A
0.263A
0.114A
0.095A
0.090A
79.3W
78.1W
77.9W
24.3W
24.2W
24.2W
0.998
0.991
0.987
0.964
0.916
0.889
3.2%
5.7%
6.8%
9.5%
11.0%
11.5%
479V
479V
479V
479V
479V
479V
0.156A
0.156A
0.156A
0.046A
0.046A
0.046A
75W
75W
75W
22W
22W
22W
25V
25V
25V
8V
8V
8V
0.946
0.960
0.963
0.905
0.910
0.910
7.5mA
0.7mA
0.2mA
0.3mA
10.7V
9.9V
479V
0
0
60V
220V300V
Page 21 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
5 0 0 µH
L1
R 12
R15
C 11
µ68(x)
L1: 500µH
RM14, N67, AL=160
W 1=56 Wdg. 60x0.1d
W 2=11 Wdg. 0,3d
BUZ91 : 600V, 0Ω8
120Ω
R 14
C6
µ68(x) S14
K250
R 6A
470k
B250C5000/3300
VIN
90-270V
AC
1.2m H
SIOV
M 2,5A
L4
1.2m H
L3
U
2x6.8m H
L2
Fuse
C5
3n3(Y )
C7
3n3(Y )
Z15
D8
C 13
D7
470Ω
D6
2xB YT 01
-400
L5
1n/400V 70µH
R8
R8A
120k
120k
5
R6
470k
R 10
3
R7
9k1
D3
VOUT
D5
D2
C4
10n
8
D4
C9
µ68
C10
47µ/25V
TDA 4862
D1
410V DC
M UR856
6
7
1
R9
33k
Q1
BUZ
91
R 4A
820k
R4
820k
12Ω
C2
1µ
C8
C 12
µ15
R2
5k1
150µF
450V
C1
1µ
2
4
R 13
C 11
270p
1k
R11
0Ω23
R5
10k
GND
Figure 11: 150W Universal Input Power Factor Preconverter with TDA 4862
Page 22 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
Table 4: Measurement of input and output values
90V - 270V / 150W Universal input for SMPS
(Cout = 150µF, L1=500µH)
V RM S
I RM S
P IN
PF
THD
V OUT
I OUT
P OUT
V OUTAC
η
I z (15V)
or V CC
real power
90V
120V
180V
230V
270V
1.844A
1.346A
0.876A
0.686A
0.590A
166.4W
161.0W
157.2W
155.9W
155.0W
0.998
0.999
0.996
0.987
0.973
2.8%
2.8%
4.9%
7.0%
9.5%
410V
409V
409V
409V
409V
366mA
366mA
366mA
366mA
366mA
150W
150W
150W
150W
150W
10Vpp
10Vpp
10Vpp
10Vpp
10Vpp
0.901
0.932
0.954
0.962
0.968
0.2mA
1.4mA
4.4mA
6.1mA
6.6mA
90V
120V
180V
230V
270V
0.379A
0.290A
0.209A
0.187A
0.178A
33.9W
34.0W
34.3W
34.3W
34.0W
0.994
0.981
0.911
0.798
0.708
6.6%
8.1%
9.8%
11.2%
14.8%
409V
409V
409V
409V
409V
73mA
73mA
73mA
73mA
73mA
30W
30W
30W
30W
30W
2Vpp
2Vpp
2Vpp
2Vpp
2Vpp
0.885
0.882
0.875
0.875
0.882
5.3mA
8.8mA
12.1mA
9.5mA
4.5mA
180V
90V270V
0.119A
14.1W
0.66
24.5%
409V
23mA
9.4W
0.667
409V
0
0
0.8Vpp
max.
6Vpp
9.6mA
selfsupply
Page 23 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
7 5 0 µH
L1
2x1N4148
R 15
C 11
µ47(x )
L1: 750µH
E36/11, N27, gap=2mm
W 1=85 Wdg., 40x0.1d
W 2=17 Wdg., 0,3d
BUZ90 : 600V, 1Ω6
120Ω
R 14
C6
µ47(x ) S14
K250
C7
3n3(Y )
D6
R9
33k
R 12
3n3/400V 470Ω
R8
R 8A
120k
120k
5
R6
470k
R 10
3
R7
9k1
D3
C 13
D7
R 6A
470k
B250C1500/1000
VIN
90-270V
AC
C5
3n3(Y )
SIOV
M 1.6A
1.2m H
L3
U
2x 6.8m H
L2
F use
VOUT
D5
D2
C4
10n
8
D4
C9
µ22
C 10
47µ/25V
TDA 4862
D1
410V DC
MUR856
6
7
1
R 4A
820k
Q1
BUZ
90
R4
820k
12Ω
C2
1µ
C8
100µF
450V
R2
5k1
C1
1µ
2
4
R 13
C 11
270p
1k
R 11
0Ω3
R5
10k
GND
Figure 12: 110W Universal Input Power Factor Preconverter with TDA 4862
Page 24 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
Table 5: Measurement of input and output values
90V 270V/110W Universal input for SMPS
(Cout = 100µF, L1=750µH)
V RM S
I RM S
P IN
PF
THD
V OUT
I OUT
P OUT
V OUTAC
η
real power
I z (15V)
or V CC
90V
120V
180V
230V
270V
1.355A
0.984A
0.643A
0.505A
0.434A
122.7W
118.0W
115.3W
115.0W
114.4W
0.999
0.999
0.995
0.986
0.972
2.9%
3.0%
5.6%
8.6%
11.5%
410V
410V
410V
410V
410V
268mA
268mA
268mA
268mA
268mA
110W
110W
110W
110W
110W
11Vpp
11Vpp
11Vpp
11Vpp
11Vpp
0.896
0.932
0.954
0.956
0.961
1.2mA
3.2mA
7.0mA
8.2mA
7.3mA
90V
120V
180V
230V
270V
90V270V
0.280A
0.213A
0.153A
0.132A
0.141A
25.0W
25.2W
25.4W
25.3W
24.5W
0.994
0.984
0.921
0.830
0.646
7.5%
7.8%
10.2%
9.5%
42%
410V
410V
410V
410V
410V
53.6mA
53.6mA
53.6mA
53.6mA
53.6mA
22W
22W
22W
22W
22W
2Vpp
2Vpp
2Vpp
2Vpp
10Vpp
0.880
0.873
0.866
0.870
0.898
5.8mA
8.3mA
9.6mA
7.5mA
0.1mA
410V
0
0
33Vpp
Page 25 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
Summary of used Nomenclature
Physics:
General identifiers:
Special identifiers:
A ......... cross area
b, B ..... magnetic inductance
C ......... capacitance
d, D ..... duty cycle
f........... frequency
i, I........ current
L.......... inductance
N ......... number of turns
p, P ..... power
t, T ...... time, time-intervals
v, V ..... voltage
W ........ energy
η.......... efficiency
AL ........ inductance factor
V(BR)CES collector-emitter breakdown voltage of
IGBT
VF ........ forward voltage of diodes
Vrrm ...... maximum reverse voltage of diodes
K1, K2 .. ferrite core constants
big letters: contant values and time intervals
small letters: time variant values
Components:
C ......... capacitor
D ......... diode
IC ........ integrated circuit
L.......... inductor
R ......... resistor
TR....... transformer
Indices:
AC....... alternating current value
DC ...... direct current value
BE....... basis-emitter value
CS....... current sense value
J.......... Junction value
OPTO . optocoupler value
P ......... primary side value
Pk ....... peak value
R ......... reflected from secondary to primary side
S ......... secondary side value
Sh ....... shunt value
UVLO.. undervoltage lockout value
Z ......... zener value
fmin..... value at minimum pulse frequency
i........... running variable
in......... input value
max..... maximum value
min...... minimum value
off ....... turn-off value
on ....... turn-on value
out ...... output value
p ......... pulsed
rip ....... ripple value
1,2,3 ... on-going designator
Page 26 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
References
[1]
Infineon Technologies AG: TDA4862 – Power factor and boost converter controller for high
power factor and low THD; data sheet; Infineon Technologies AG; Munich; Germany; 07/01.
Page 27 of 28
AN-PFC-TDA4862-1
TDA4862 - Technical Description
Revision History
Application Note AN-PFC-TDA4862-1
Actual Release: V1.2 Date: 18.09.2001
Page
of Page
actual
Previous Release: 1.1
of Subjects changed since last release
prev. Rel.
Rel.
29
29
Deleted
28
28
updated
For questions on technology, delivery and prices please contact the Infineon Technologies Offices in
Germany or the Infineon Technologies Companies and Representatives worldwide: see the address
list on the last page or our webpage at
http://www.infineon.com
CoolMOS and CoolSET are trademarks of Infineon Technologies AG.
Edition 2001-03-01
Published by Infineon Technologies AG,
St.-Martin-Strasse 53,
D-81541 München
© Infineon Technologies AG 2000.
All Rights Reserved.
Attention please!
The information herein is given to describe certain components and shall not be considered as warranted characteristics.
Terms of delivery and rights to technical change reserved.
We hereby disclaim any and all warranties, including but not limited to warranties of non-infringement, regarding circuits, descriptions and charts
stated herein.
Infineon Technologies is an approved CECC manufacturer.
Information
For further information on technology, delivery terms and conditions and prices please contact your nearest Infineon Technologies Office in
Germany or our Infineon Technologies Representatives worldwide (see address list).
Warnings
Due to technical requirements components may contain dangerous substances. For information on the types in question please contact your
nearest Infineon Technologies Office.
Infineon Technologies Components may only be used in life-support devices or systems with the express written approval of Infineon
Technologies, if a failure of such components can reasonably be expected to cause the failure of that life-support device or system, or to affect the
safety or effectiveness of that device or system. Life support devices or systems are intended to be implanted in the human body, or to support
and/or maintain and sustain and/or protect human life. If they fail, it is reasonable to assume that the health of the user or other persons may be
endangered.
Page 28 of 28
AN-PFC-TDA4862-1