DATASHEET

DATASHEET
40V 2.5A Buck Controller with Integrated High-side
MOSFET
ISL78206
Features
The ISL78206 is an AEC-Q100 qualified 40V, 2.5A synchronous
buck controller with a high-side MOSFET and low-side driver
integrated. The ISL78206 supports a wide input voltage range
from 3V to 40V, output current up to 2.5A, and can be
operated in synchronous and non-synchronous buck
topologies. The ISL78206 provides a low system cost, high
efficiency, single part buck solution for automotive
applications for a wide variety of input voltages. For vehicle
systems that must be kept powered during cold-cranking or
start-stop operation, the ISL78206 can be replaced by its
pin-to-pin alternative, the ISL78201 which offers boost-buck
operation. Together, these devices offer two functional
alternatives on the same PCB layout, providing convenience,
flexibility, and facilitating extensive design reuse.
• Ultra wide input voltage range 3V to 40V (refer to “Input
Voltage” on page 12 for more details)
The ISL78206 offers the most robust current protections. It
uses peak current mode control with cycle-by-cycle current
limiting. It is implemented with frequency foldback
undercurrent limit conditions. In addition, the hiccup
overcurrent mode is also implemented to guarantee reliable
operations under harsh short conditions.
The ISL78206 has comprehensive protections against various
faults, including overvoltage, undervoltage, programmable
overcurrent and over-temperature. Built-in soft-start allows the
IC to start-up smoothly and is reactivated during hiccup mode
fault recovery.
• Less than 5µA (max) shutdown input current (IC disabled)
• Temperature range -40°C to +105°C
• Integrated high-side MOSFET
• Operational topologies
- Synchronous buck
- Non-synchronous buck
• Programmable frequency from 200kHz to 2.2MHz and
frequency synchronization capability
• ±1% tight voltage regulation accuracy
• Reliable cycle-by-cycle overcurrent protection
- Temperature compensated current sense
- Frequency foldback
- Programmable OC limit
- Hiccup mode protection in worst case short condition
• 20 Ld HTSSOP package, pin-to-pin compatible with
ISL78201 boost buck
• AEC-Q100 qualified
• Pb-free (RoHS compliant)
Applications
• Automotive applications
• General purpose power regulator
• 24V bus power
• Battery power
• Embedded processor and I/O supplies
SYNC
BOOT
FS
VCC
ILIMIT
SS
VIN
ISL78206
PHASE
V OUT
LGATE
PGND
DGND
SGND
FB
COMP
EFFICIENCY (%)
VIN
PGOOD
EN
100
95 6V
90
85
80
75
70
65
60
55
50
45
40
35
30
0.0
12V
24V
40V
0.5
1.0
1.5
2.0
2.5
LOAD CURRENT (A)
FIGURE 1. TYPICAL APPLICATION SCHEMATIC I - SYNCHRONOUS
BUCK
March 25, 2015
FN8618.2
1
FIGURE 2. EFFICIENCY, SYNCHRONOUS BUCK, 500kHz, VOUT 5V,
TA = +25°C
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2013-2015. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL78206
Pin Configuration
ISL78206
(20 LD HTSSOP)
TOP VIEW
20 LGATE
PGND 1
19 SYNC
BOOT 2
VIN 3
18 NC
VIN 4
SGND 5
VCC 6
17 PHASE
21
PAD
16 PHASE
15 PGOOD
NC 7
14 DGND
EN 8
13 ILIMIT
FS 9
12 COMP
SS 10
11 FB
Functional Pin Description
PIN NAME
PIN #
DESCRIPTION
PGND
1
This pin is used as the ground connection of the power flow, including the driver.
BOOT
2
This pin provides bias voltage to the high-side MOSFET driver. A bootstrap circuit is used to create a voltage suitable to drive
the internal N-channel MOSFET. The boot charge circuitries are integrated inside of the IC. No external boot diode is needed.
A 1µF ceramic capacitor is recommended to be used between the BOOT and PHASE pin.
VIN
3, 4
Connect the input rail to these pins that are connected to the drain of the integrated high-side MOSFET, as well as the source
for the internal linear regulator that provides the bias of the IC.
With the part switching, the operating input voltage applied to the VIN pins must be under 40V. This recommendation allows
for short voltage ringing spikes (within a couple of ns time range) due to switching while not exceeding Absolute Maximum
Ratings.
SGND
5
This pin provides the return path for the control and monitor portions of the IC.
VCC
6
This pin is the output of the internal linear regulator that supplies the bias for the IC, including the driver. A minimum 4.7µF
decoupling ceramic capacitor is recommended between VCC to ground.
EN
8
The controller is enabled when this pin is pulled HIGH or left floating. The IC is disabled when this pin is pulled LOW. Range:
0V to 5.5V.
FS
9
Tying this pin to VCC, or GND, or leaving it open will force the IC to have 500kHz switching frequency. The oscillator switching
frequency can also be programmed by adjusting the resistor from this pin to GND.
SS
10
Connect a capacitor from this pin to ground. This capacitor, along with an internal 5µA current source, sets the soft-start
interval of the converter. Also, this pin can be used to track a ramp on this pin.
FB
11
This pin is the inverting input of the voltage feedback error amplifier. With a properly selected resistor divider connected from
VOUT to FB, the output voltage can be set to any voltage between the input rail (reduced by maximum duty cycle and voltage
drop) and the 0.8V reference. Loop compensation is achieved by connecting an RC network across COMP and FB. The FB pin
is also monitored for overvoltage events.
COMP
12
Output of the voltage feedback error amplifier.
ILIMIT
13
Programmable current limit pin. With this pin connected to VCC pin, or to GND, or left open, the current limit threshold is set
to a default of 3.6A; the current limit threshold can be programmed with a resistor from this pin to GND.
DGND
14
Digital ground pin. Connect to SGND at quiet ground copper plane.
PGOOD
15
PGOOD is an open drain output and pull up this pin with a resistor to VCC for proper function. PGOOD will be pulled low
under the events when the output is out of regulation (OV or UV) or EN pin is pulled low. PGOOD rising has a fixed 128
cycles delay.
PHASE
16, 17
These pins are the PHASE nodes that should be connected to the output inductor. These pins are connected to the source of
the high side N-channel MOSFET.
SYNC
19
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This pin can be used to synchronize two or more ISL78206 controllers. Multiple ISL78206s can be synchronized with their
SYNC pins connected together. 180 degree phase shift is automatically generated between the master and slave ICs.
The internal oscillator can also lock to an external frequency source applied to this pin with square pulse waveform (with
frequency 10% higher than the IC’s local frequency, and pulse width higher than 150ns).
This pin should be left floating if not used.
2
FN8618.2
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ISL78206
Functional Pin Description (Continued)
PIN NAME
PIN #
DESCRIPTION
LGATE
20
In synchronous buck mode, this pin is used to drive the lower side MOSFET to improve efficiency. A 5.1k or smaller value
resistor has to be added to connect LGATE to ground to avoid falsely turn-on of LGATE caused by coupling noise.
In non-synchronous buck when a diode is used as the bottom side power device, this pin should be connected to VCC through
a resistor (less than 5k) before VCC start-up to disable the low side driver (LGATE).
NC
7, 18
PAD
21
No connection pin. Connect these pins to SGND at quiet ground copper plane.
Bottom thermal pad. It is not connected to any electrical potential of the IC. In layout, it must be connected to PCB ground
copper plane with area as large as possible to effectively reduce the thermal impedance.
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
ISL78206AVEZ
PART
MARKING
78206 AVEZ
TEMP. RANGE
(°C)
-40 to +105
PACKAGE
(Pb-Free)
20 Ld HTSSOP
PKG.
DWG. #
M20.173A
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL78206. For more information on MSL please see techbrief TB363.
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March 25, 2015
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Block Diagram
PGOOD
VCC
VIN (x2)
VIN
CURRENT
MONITOR
BIAS LDO
4
ILIMIT
POWER-ON
RESET
SGND
VCC
BOOT
CONTROL LOGIC
OCP, OVP, OTP
EN
VOLTAGE
MONITOR
SYNC
FS
SLOPE
COMPENSATION
OSCILLATOR
+
SOFT-START
LOGIC
VCC
5 µA
BOOT REFRESH
0.8V
REFERENCE
COMPARATOR
EA
SS
+
LGATE
FB
COMP
FIGURE 3. BLOCK DIAGRAM
PGND
ISL78206
PHASE (x2)
GATE DRIVE
FN8618.2
March 25, 2015
ISL78206
Typical Application Schematic I - Synchronous Buck
VIN
PGOOD
EN
SYNC
BOOT
FS
VCC
VIN
ISL78206
ILIMIT
V OUT
PHASE
LGATE
SS
PGND
FB
DGND
SGND
COMP
FIGURE 4. TYPICAL APPLICATION SCHEMATIC I - SYNCHRONOUS BUCK
Typical Application Schematic II - Non-Synchronous Buck
VIN
PGOOD
EN
SYNC
BOOT
FS
VCC
ILIMIT
SS
DGND
SGND
VIN
ISL78206
PHASE
V OUT
LGATE
PGND
FB
COMP
FIGURE 5. TYPICAL APPLICATION SCHEMATIC II - NON-SYNCHRONOUS BUCK
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ISL78206
Absolute Maximum Ratings
Thermal Information
VIN, PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +44V
VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +6.0V
Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +50.0V
Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . +6.0V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to VCC + 0.3V
ESD Rating
Human Body Model (Tested per JESD22-A114F) . . . . . . . . . . . . . . 2000V
Machine Model (Tested per JESD22-A115C) . . . . . . . . . . . . . . . . . . 250V
Charged Device Model (Tested per AEC-Q100-11) . . . . . . . . . . . . . 1000V
Latch-up Rating (Tested per JESD78B; Class II, Level A) . . . . . . . . . 100mA
Thermal Resistance
JA (°C/W) JC (°C/W)
20 Ld HTSSOP Package (Notes 4, 5) . . . . .
32
3.5
Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range. . . . . . . . . . . . . . . . . -65°C to +150°C
Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see TB493
Recommended Operating Conditions
Supply Voltage on VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3V to 40V
Ambient Temperature Range (Automotive). . . . . . . . . . . . . . -40°C to +105°C
Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . . -40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Refer to “Block Diagram” on page 4 and Typical Application Schematics on page 5. Operating Conditions
Unless Otherwise Noted: VIN = 12V, or VCC = 4.5V, TA = -40°C to +105°C. Typicals are at TA = +25°C. Boldface limits apply across the operating
temperature range, -40°C to +105°C.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 7)
TYP
MAX
(Note 7) UNITS
VIN SUPPLY
VIN Voltage Range
Operating Supply Current
IQ
Shut Down Supply Current
IIN_SD
VIN
3.05
40
V
VIN connected to VCC
3.05
5.5
V
IC operating, not including driving current,
VIN = 12V
1.3
EN connected to GND, VIN = 12V
2.8
4.5
µA
4.5
4.8
V
0.3
0.52
V
0.25
0.42
V
mA
INTERNAL MAIN LINEAR REGULATOR
MAIN LDO VCC Voltage
VCC
MAIN LDO Dropout Voltage
VIN > 5V
4.2
VDROPOUT_MAIN VIN = 4.2V, IVCC = 35mA
VIN = 3V, IVCC = 25mA
VCC CURRENT LIMIT of MAIN LDO
60
mA
POWER-ON RESET
Rising VCC POR Threshold
VPORH_RISE
Falling VCC POR Threshold
VCC POR Hysteresis
2.82
2.9
3.05
V
VPORL_FALL
2.6
2.8
V
VPORL_HYS
0.3
V
ENABLE
Required Enable On Voltage
VENH
Required Enable Off Voltage
VENL
EN Pull-up Current
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IEN_PULLUP
6
1.7
V
1
V
VEN = 1.2V, VIN = 24V
1.5
µA
VEN = 1.2V, VIN = 12V
1.2
µA
VEN = 1.2V, VIN = 5V
0.9
µA
FN8618.2
March 25, 2015
ISL78206
Electrical Specifications
Refer to “Block Diagram” on page 4 and Typical Application Schematics on page 5. Operating Conditions
Unless Otherwise Noted: VIN = 12V, or VCC = 4.5V, TA = -40°C to +105°C. Typicals are at TA = +25°C. Boldface limits apply across the operating
temperature range, -40°C to +105°C. (Continued)
PARAMETER
MAX
(Note 7) UNITS
MIN
(Note 7)
TYP
RT = 665kΩ
160
200
240
kHz
RT = 51.1kΩ
1870
2200
2530
kHz
FS Pin connected to VCC or Floating or GND
450
500
550
kHz
SYMBOL
TEST CONDITIONS
OSCILLATOR
PWM Frequency
fOSC
MIN ON Time
tMIN_ON
130
225
ns
MIN OFF Time
tMIN_OFF
210
330
ns
Input High Threshold
VIH
2
V
Input Low Threshold
VIL
0.5
V
Input Minimum Pulse Width
25
ns
Input Impedance
100
kΩ
Input Minimum Frequency Divided by Free Running
Frequency
1.1
Input Maximum Frequency Divided by Free Running
Frequency
1.6
SYNCHRONIZATION
Output Pulse Width
CSYNC = 100pF
100
ns
RLOAD = 1kΩ
VCC0.25
V
Output Pulse High
VOH
Output Pulse Low
VOL
GND
V
VREF
0.8
V
REFERENCE VOLTAGE
Reference Voltage
System Accuracy
-1.0
FB Pin Source Current
1.0
5
%
nA
SOFT-START
Soft-start Current
ISS
3
5
7
µA
ERROR AMPLIFIER
Unity Gain-bandwidth
CLOAD = 50pF
DC Gain
CLOAD = 50pF
10
MHz
88
dB
Maximum Output Voltage
3.6
V
Minimum Output Voltage
0.5
V
5
V/µs
Slew Rate
SR
CLOAD = 50pF
INTERNAL HIGH-SIDE MOSFET
Upper MOSFET rDS(ON)
rDS(ON)_UP
(Note 6) Limits apply for +25°C
127
140
mΩ
LGate Source Resistance
100mA Source Current
3.5
Ω
LGATE Sink Resistance
100mA Sink Current
2.8
Ω
LOW-SIDE MOSFET GATE DRIVER
POWER GOOD MONITOR
Overvoltage Rising Trip Point
VFB/VREF
Percentage of Reference Point
Overvoltage Rising Hysteresis
VFB/VOVTRIP
Percentage Below OV Trip Point
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7
104
110
3
116
%
%
FN8618.2
March 25, 2015
ISL78206
Electrical Specifications
Refer to “Block Diagram” on page 4 and Typical Application Schematics on page 5. Operating Conditions
Unless Otherwise Noted: VIN = 12V, or VCC = 4.5V, TA = -40°C to +105°C. Typicals are at TA = +25°C. Boldface limits apply across the operating
temperature range, -40°C to +105°C. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
Undervoltage Falling Trip Point
VFB/VREF
Percentage of reference point
Undervoltage Falling Hysteresis
VFB/VUVTRIP
Percentage above UV trip point
MIN
(Note 7)
TYP
84
90
MAX
(Note 7) UNITS
96
%
3
%
128
cycles
PGOOD HIGH, VPGOOD = 4.5V
10
nA
VPGOOD
PGOOD LOW, IPGOOD = 0.2mA
0.10
V
Default Cycle-by-cycle Current Limit Threshold
IOC_1
ILIMIT = GND or VCC or Floating
Hiccup Current Limit Threshold
IOC_2
Hiccup, IOC_2/IOC_1
115
%
OV 120% Trip Point
Active in and after soft-start
Percentage of Reference Point
LG = UG = LOW
120
%
OV 120% Release Point
Active in and after soft-start
Percentage of Reference Point
102.5
%
OV 110% Trip Point
Active after soft-start done
Percentage of Reference Point
LG = UG = LOW
110
%
OV 110% Release Point
Active after soft-start done.
Percentage of Reference Point
102.5
%
Over-temperature Trip Point
160
°C
Over-temperature Recovery Threshold
140
°C
PGOOD Rising Delay
tPGOODR_DELAY fOSC = 500kHz
PGOOD Leakage Current
PGOOD Low Voltage
OVERCURRENT PROTECTION
3
3.6
4.2
A
OVERVOLTAGE PROTECTION
OVER-TEMPERATURE PROTECTION
NOTES:
6. Wire bonds included.
7. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
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ISL78206
100
95 6V
90
85
80
75
70
65
60
55
50
45
40
35
30
0.0
12V
24V
EFFICIENCY (%)
EFFICIENCY (%)
Typical Performance Curves
40V
0.5
1.0
1.5
2.0
2.5
100
6V
95
90
85
80
75
70
65
60
55
50
45
40
35
30
0.0
12V
24V
40V
0.5
LOAD CURRENT (A)
4.970
4.970
4.968
4.968
4.966
4.966
4.964
4.964
24V
4.960
4.958
4.956
4.954
12V
6V
2.5
4.960
4.958
IO = 2A
4.954
IO = 1A
4.952
4.950
0.0
0.5
1.0
1.5
2.0
4.950
2.5
0
FIGURE 8. LOAD REGULATION, VOUT 5V, TA = +25°C
80
75
75
IC DIE TEMPERATURE (°C)
85
80
VOUT = 20V
65
60
VOUT = 12V
55
50
45
40
VOUT = 5V
35
25
30
35
40
45
VIN (V)
FIGURE 10. IC DIE TEMPERATURE UNDER +25°C AMBIENT
TEMPERATURE, STILL AIR, 500kHz, IO = 2A
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30
35
40
45
50
50
VIN = 40V
55
50
45
40
35
25
1.0
20
25
60
25
15
20
65
30
10
15
70
30
5
10
FIGURE 9. LINE REGULATION, VOUT 5V, TA = +25°C
85
70
5
INPUT VOLTAGE (V)
LOAD CURRENT (A)
IC DIE TEMPERATURE (°C)
2.0
IO = 0A
4.962
4.956
40V
4.952
0
1.5
FIGURE 7. EFFICIENCY, SYNCHRONOUS BUCK, 500kHz,
VOUT 3.3V, TA = +25°C
VOUT (V)
VOUT (V)
FIGURE 6. EFFICIENCY, SYNCHRONOUS BUCK, 500kHz, VOUT 5V,
TA = +25°C
4.962
1.0
LOAD CURRENT (A)
VIN = 6.5V
VIN = 12V
1.5
VIN = 24V
2.0
2.5
IOUT (A)
FIGURE 11. IC DIE TEMPERATURE UNDER +25°C AMBIENT
TEMPERATURE, STILL AIR, 500kHz, VOUT = 5V
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ISL78206
Typical Performance Curves (Continued)
180
170
160
150
140
130
120
130
110
120
90
100
80
70
60
50
40
30
20
0
10
-10
-20
-50
100
-30
110
-40
UPPER MOSFET rDS(ON) (mΩ)
190
DIE TEMPERATURE (°C)
FIGURE 12. UPPER MOSFET rDS(ON) (mΩ) OVER-TEMPERATURE
VOUT 2V/DIV
VOUT 2V/DIV
PHASE 20V/DIV
PHASE 20V/DIV
2ms/DIV
FIGURE 13. VIN 36V, PRE-BIASED START-UP
2ms/DIV
FIGURE 14. SYNCHRONOUS BUCK MODE, VIN 36V, IO 2A,
ENABLE ON
VOUT 20mV/DIV (5V OFFSET)
VOUT 2V/DIV
PHASE 20V/DIV
PHASE 20V/DIV
5µs/DIV
FIGURE 15. SYNCHRONOUS BUCK, VIN 36V, IO 2A
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2ms/DIV
FIGURE 16. SYNCHRONOUS BUCK MODE, VIN 36V, IO 2A,
ENABLE OFF
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ISL78206
Typical Performance Curves (Continued)
VOUT 100mV/DIV (5V OFFSET)
VOUT 10mV/DIV (5V OFFSET)
IOUT 1A/DIV
PHASE 5V/DIV
PHASE 20V/DIV
20µs/DIV
1ms/DIV
FIGURE 17. VIN 24V, 0A TO 2A STEP LOAD
FIGURE 18. NON-SYNCHRONOUS BUCK, FORCE PWM MODE,
VIN 12V, NO LOAD
VOUT 10mV/DIV (5V OFFSET)
PHASE 10V/DIV
5µs/DIV
FIGURE 19. NON-SYNCHRONOUS BUCK, FORCE PWM MODE, VIN 12V, 2A
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ISL78206
Functional Description
Synchronous and Non-synchronous Buck
Initialization
The ISL78206 supports both synchronous and non-synchronous
buck operations.
Initially, the ISL78206 continually monitors the voltage at the EN
pin. When the voltage on the EN pin exceeds its rising threshold,
the internal LDO will start-up to build up VCC. After Power-On
Reset (POR) circuits detect that the VCC voltage has exceeded
the POR threshold, the soft-start will be initiated.
Soft-start
The soft-start (SS) ramp is built up in the external capacitor on
the SS pin that is charged by an internal 5µA current source.
C SS  F  = 6.5  t SS  S 
(EQ. 1)
The SS ramp starts from 0 to voltage above 0.8V. Once SS
reaches 0.8V, the bandgap reference takes over and the IC gets
into steady state operation. The soft-start time is referring to the
duration for SS pin ramps from 0 to 0.8V while output voltage
ramps up with the same rate from 0 to target regulated voltage.
The required capacitance at SS pin can be calculated from
Equation 1.
The SS plays a vital role in the hiccup mode of operation. The IC
works as cycle-by-cycle peak current limiting at over load
condition. When a harsh condition occurs and the current in the
upper side MOSFET reaches the second overcurrent threshold,
the SS pin is pulled to ground and a dummy soft-start cycle is
initiated. At dummy SS cycle, the current to charge the soft-start
cap is cut down to 1/5 of its normal value. Therefore, a dummy
SS cycle takes 5 times that of the regular SS cycle. During the
dummy SS period, the control loop is disabled and there is no
PWM output. At the end of this cycle, it will start the normal SS.
The hiccup mode persists until the second overcurrent threshold
is no longer reached.
The ISL78206 is capable of starting up with pre-biased output.
PWM Control
The ISL78206 employs the peak current mode PWM control for
fast transient response and cycle-by-cycle current limiting. See
the “Block Diagram” on page 4.
The PWM operation is initialized by the clock from the oscillator.
The upper MOSFET is turned on by the clock at the beginning of a
PWM cycle and the current in the MOSFET starts to ramp up.
When the sum of the current sense signal and the slope
compensation signal reaches the error amplifier output voltage
level, the PWM comparator is triggered to shut down the PWM
logic to turn off the high side MOSFET. The high side MOSFET
stays off until the next clock signal starts.
In synchronous buck configuration, a 5.1k or smaller value
resistor has to be added to connect LGATE to ground to avoid
falsely turn-on of LGATE caused by coupling noise.
For a non-synchronous buck operation when a power diode is
used as the low-side power device, the LGATE driver can be
disabled with LGATE connected to VCC (before IC start-up). For
non-synchronous buck, the phase node will show oscillations
after high-side turns off (as shown in Figure 18 - blue trace). This
is normal due to the oscillations among the parasitic capacitors
at phase node and output inductor. An RC snubber (suggesting
200Ω and 2.2nF as typical) at phase node can reduce this
ringing.
Input Voltage
With the part switching, the operating input voltage applied to
the VIN pins must be under 40V. This recommendation allows for
short voltage ringing spikes (within a couple of ns time range)
due to switching while not exceeding Absolute Maximum
Ratings.
The lowest IC operating input voltage (VIN pin) depends on VCC
voltage and the Rising and Falling VCC POR Threshold in the
Electrical Specifications table on page 6. At IC startup, when VCC
is just over the rising POR threshold, there is no switching yet
before the soft-start starts. Therefore, the IC minimum start-up
voltage on VIN pin is 3.05V (MAX of Rising VCC POR). When the
soft-start is initiated, the regulator is switching and the dropout
voltage across the internal LDO increases due to driving current.
Thus the IC VIN pin shutdown voltage is related to driving current
and VCC POR falling threshold. The internal upper side MOSFET
has typical 10nC gate drive. For a typical example of synchronous
buck with 4nC lower MOSFET gate drive and 500kHz switching
frequency, the driving current is 7mA total causing 70mV drop
across internal LDO under 3V VIN. Then the IC shutdown voltage
on the VIN pin is 2.87V (2.8V + 0.07V). In practical design, extra
room should be taken into account with concerns of voltage
spikes at VIN.
Output Voltage
The output voltage can be programmed down to 0.8V by a
resistor divider from VOUT to FB. For buck, the maximum
achievable voltage is (VIN*DMAX - VDROP), where VDROP is the
voltage drop in the power path, including mainly the MOSFET
rDS(ON) and inductor DCR. The maximum duty cycle DMAX is
decided by (1 - Fs * tMIN(OFF)).
The output voltage is sensed by a resistor divider from VOUT to FB
pin. The difference between the FB voltage and 0.8V reference is
amplified and compensated to generate the error voltage signal
at the COMP pin. Then the COMP pin signal is compared with the
current ramp signal to shut down the PWM.
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12
FN8618.2
March 25, 2015
ISL78206
Output Current
1200
Figures 10 and 11 on page 9 show the thermal performance of
this part operating in buck at different conditions. Figure 10
shows 2A buck applications under +25°C still air conditions.
Different VOUT (5V, 12V, 20V) applications thermal data are
shown over VIN range at +25°C and still air. The temperature rise
data in Figure 10 can be used to estimate the die temperature at
different ambient temperatures under various operating
conditions. Note that more temperature rise is expected at higher
ambient temperature due to more conduction loss caused by
rDS(ON) increase. Figure 11 shows 5V output applications'
thermal performance under various output current and input
voltage. It shows the temperature rise trend with load and VIN
changes. The part can output 2.5A under typical application
conditions (VIN 8~30V, VOUT 5V, 500kHz, still air and +85°C
ambient conditions). The output current should be derated under
any conditions, causing the die temperature to exceed +125°C.
Basically, the die temperature is equal to the sum of the ambient
temperature and the temperature rise resulting from the power
dissipated from the IC package with a certain junction to
ambient thermal impedance JA. The power dissipated in the IC
is related to the MOSFET switching loss, conduction loss and the
internal LDO loss. Besides the load, these losses are also related
to input voltage, output voltage, duty cycle, switching frequency
and temperature. With the exposed pad at the bottom, the heat
of the IC mainly goes through the bottom pad and JA is greatly
reduced. The JA is highly related to layout and air flow
conditions. In layout, multiple vias (20 recommended) are
strongly recommended in the IC bottom pad. In addition, the
bottom pad with its vias should be placed in the ground copper
plane with an area as large as possible connected through
multiple layers. The JA can be reduced further with air flow.
For applications with high output current and bad operating
conditions (compact board size, high ambient temperature, etc.),
synchronous buck is highly recommended since the external
low-side MOSFET generates smaller heat than external low-side
power diode. This helps to reduce PCB temperature rise around
the ISL78206 and less junction temperature rise.
Oscillator and Synchronization
The oscillator has a default frequency of 500kHz with the FS pin
connected to VCC, or ground, or floating. The frequency can be
programmed to any frequency between 200kHz and 2.2MHz with
a resistor from the FS pin to GND.
145000 – 16  FS  kHz 
R FS  k  = ------------------------------------------------------------FS  kHz 
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13
(EQ. 2)
1000
800
RFS (kΩ)
With the high side MOSFET integrated, the maximum current
that the ISL78206 can support is decided by the package and
many operating conditions, including input voltage, output
voltage, duty cycle, switching frequency and temperature, etc.
From the thermal perspective, the die temperature shouldn’t be
above +125°C with the power loss dissipated inside of the IC.
600
400
200
0
0
500
1000
1500
FS (kHz)
2000
2500
FIGURE 20. RFS vs FREQUENCY
The SYNC pin is bidirectional and it outputs the IC’s default or
programmed local clock signal when it’s free running. The IC
locks to an external clock injected to SYNC pin (external clock
frequency recommended to be 10% higher than the free running
frequency). The delay from the rising edge of the external clock
signal to the PHASE rising edge is half of the free running switching
period pulse 220ns, (0.5Tsw +220ns). The maximum external clock
frequency is recommended to be 1.6 of the free running frequency.
With the SYNC pins simply connected together, multiple
ISL78206s can be synchronized. The slave ICs automatically
have a 180 degree phase shift with respect to the master IC.
PGOOD
The PGOOD pin is output of an open drain transistor (refer to
“Block Diagram” on page 4). An external resistor is required to be
pulled up to VCC for proper PGOOD function. At startup, PGOOD
will be turned HIGH (internal PGOOD open drain transistor is
turned off) with 128 cycles delay after soft-start is finished
(soft-start ramp reaches 1.02V) and FB voltage is within OV/UV
window (90%REF < FB < 110%REF).
At normal operation, PGOOD will be pulled low with 1 cycle
(minimum) and 6 cycles (maximum) delay if any of the OV
(110%) or UV (90%) comparator is tripped. The PGOOD will be
released HIGH with 128 cycles delay after FB recovers to be
within OV/UV window (90%REF < FB < 110%REF). When EN is
pulled low or VCC is below POR, PGOOD is pulled low with no
delay.
In the case when the PGOOD pin is pulled up by external bias
supply instead of VCC of itself, when the part is disabled, the
internal PGOOD open drain transistor is off, the external bias
supply can charge PGOOD pin HIGH. This should be known as false
PGOOD reporting. At start-up when VCC rise from 0, PGOOD will be
pulled low when VCC reaches 1V. After EN pulled low and VCC
falling, PGOOD internal open drain transistor will open with high
impedance when VCC falls below 1V. The time between EN pulled
low and PGOOD OPEN depends on the VCC falling time to 1V.
FN8618.2
March 25, 2015
ISL78206
Fault Protection
Thermal Protection
Overcurrent Protection
The ISL78206 PWM will be disabled if the junction temperature
reaches +160°C. There is +20°C hysteresis for OTP. The part will
restart after the junction temperature drops below +140°C.
The overcurrent function protects against any overload conditions
and output shorts at worst case, by monitoring the current
flowing through the upper MOSFET.
There are 2 current limiting thresholds. The first one, IOC1, is to
limit the high-side MOSFET peak current cycle-by-cycle. The
current limit threshold is set to a default of 3.6A with the ILIMIT
pin connected to GND or VCC, or left open. The current limit
threshold can also be programmed by a resistor RLIM at ILIMIT
pin to ground. Use Equation 3 to calculate the resistor.
300000
R LIM = ---------------------------------------I OC  A  + 0.018
(EQ. 3)
RLIM (kΩ)
Note that with the lower RLIM, IOC1 is higher. The usable resistor
value range to program OC1 peak current threshold is 40kΩ
to 330kΩ. RLIM value out of this range is not recommended.
320
300
280
260
240
220
200
180
160
140
120
100
80
60
40
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0
IOC1 (A)
FIGURE 21. RLIM vs IOC1
The second current protection threshold, IOC2, is 15% higher
than IOC1 mentioned above. Upon the instant that the high-side
MOSFET current reaches IOC2, the PWM shuts off after 2 cycle
delay and the IC enters hiccup mode. In hiccup mode, the PWM
is disabled for dummy soft-start duration equaling 5 regular
soft-start periods. After this dummy soft-start cycle, the true
soft-start cycle is attempted again. The IOC2 offers a robust and
reliable protection against worst case conditions.
The frequency foldback is implemented for the ISL78206. When
overcurrent limiting, the switching frequency is reduced to be
proportional to the output voltage in order to keep the inductor
current under limit threshold during overload conditions. The low
limit of frequency under frequency foldback is 40kHz.
Component Selection
The ISL78200 iSim model (buck mode), available on the internet
(ISL78200 iSim), and can be used to simulate the ISL78206
behaviors to assist in design.
Output Capacitors
An output capacitor is required to filter the inductor current.
Output ripple voltage and transient response are 2 critical factors
when considering output capacitance choice. The current mode
control loop allows the usage of low ESR ceramic capacitors and
thus smaller board layout. Electrolytic and polymer capacitors
may also be used.
Additional consideration applies to ceramic capacitors. While they
offer excellent overall performance and reliability, the actual
in-circuit capacitance must be considered. Ceramic capacitors are
rated using large peak-to-peak voltage swings and with no DC bias.
In the DC/DC converter application, these conditions do not reflect
reality. As a result, the actual capacitance may be considerably
lower than the advertised value. Consult the manufacturers data
sheet to determine the actual in-application capacitance. Most
manufacturers publish capacitance vs DC bias so that this effect
can be easily accommodated. The effects of AC voltage are not
frequently published, but an assumption of ~20% further
reduction will generally suffice. The result of these considerations
can easily result in an effective capacitance 50% lower than the
rated value. Nonetheless, they are a very good choice in many
applications due to their reliability and extremely low ESR.
The following equations allow calculation of the required
capacitance to meet a desired ripple voltage level. Additional
capacitance may be used.
For the ceramic capacitors (low ESR):
I
V OUTripple = ------------------------------------8 f SW C OUT
(EQ. 4)
Where I is the inductor’s peak-to-peak ripple current, fSW is the
switching frequency and COUT is the output capacitor.
If using electrolytic capacitors then:
V OUTripple = I*ESR
(EQ. 5)
Overvoltage Protection
If the voltage detected on the FB pin is over 110% or 120% of
reference, the high-side and low-side driver shuts down
immediately and keep off until FB voltage drops to 0.8V. When
the FB voltage drops to 0.8V, the drivers are released ON. 110%
OVP is off at soft-start and becomes active after soft-start is
done. 120% OVP is active before and after soft-start.
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March 25, 2015
ISL78206
I OUT 2 * L
C OUT = -------------------------------------------------------------------------------------------V OUT 2 *  V OUTMAX  V OUT  2 – 1 
(EQ. 6)
Where VOUTMAX/VOUT is the relative maximum overshoot
allowed during the removal of the load.
Input Capacitors
Output Inductor
The inductor value determines the converter’s ripple current.
Choosing an inductor current requires a somewhat arbitrary
choice of ripple current, I. A reasonable starting point is 30% to
40% of total load current. The inductor value can then be
calculated using Equation 7:
Loop Compensation Design
The ISL78206 uses constant frequency peak current mode
control architecture to achieve fast loop transient response. An
accurate current sensing pilot device in parallel with the upper
MOSFET is used for peak current control signal and overcurrent
protection. The inductor is not considered as a state variable
since its peak current is constant, and the system becomes
single order system. It is much easier to design the compensator
to stabilize the loop compared with voltage mode control. Peak
current mode control has inherent input voltage feed-forward
function to achieve good line regulation. Figure 22 shows the
small signal model of a buck regulator.
^
iL
^
i in
^
Vin
ILd^
1:D
+
RT
Low Side Power MOSFET
In synchronous buck application, a power N MOSFET is needed
as the synchronous low side MOSFET and a good one should
have low Qgd, low rDS(ON) and small Rg (Rg_typ <1.5Ω
recommended). Vgth_min is recommended to be or higher than
1.2V. A good example is SQS462EN.
Rc
Ro
Co
T i(S)
d^
Fm
(EQ. 7)
Increasing the value of inductance reduces the ripple current and
thus ripple voltage. However, the larger inductance value may
reduce the converter’s response time to a load transient. The
inductor current rating should be such that it will not saturate in
overcurrent conditions.
^
vo
RLP
Vin d^
+
V IN – V OUT V OUT
L = --------------------------------  ---------------Fs  I
V IN
LP
+
Depending upon the system input power rail conditions, the
aluminum electrolytic type capacitor is normally needed to
provide the stable input voltage and restrict the switching
frequency pulse current in small areas over the input traces for
better EMC performance. The input capacitor should be able to
handle the RMS current from the switching power devices.
Ceramic capacitors must be used at the VIN pin of the IC and
multiple capacitors, including 1µF and 0.1µF, are recommended.
Place these capacitors as closely as possible to the IC.
In applications requiring the least input quiescent current, large
resistors should be used for the divider to keep its leakage
current low. Generally, a resistor value of 10k to 300k can be
used for the upper resistor.
GAIN (VLOOP (S(fi))
Regarding transient response needs, a good starting point is to
determine the allowable overshoot in VOUT if the load is suddenly
removed. In this case, energy stored in the inductor will be
transferred to COUT causing its voltage to rise. After calculating
capacitance required for both ripple and transient needs, choose
the larger of the calculated values. The following equation
determines the required output capacitor value in order to
achieve a desired overshoot relative to the regulated voltage.
Tv (S)
He(S)
v^comp
-Av(S)
FIGURE 22. SMALL SIGNAL MODEL OF BUCK REGULATOR
PWM Comparator Gain Fm
The PWM comparator gain Fm for peak current mode control is
given by Equation 9:
1
dˆ
F m = ----------------- = ------------------------------- S e + S n T s
v̂ comp
(EQ. 9)
Where Se is the slew rate of the slope compensation and Sn is
given by Equation 10.
V in – V o
S n = R t --------------------L
(EQ. 10)
In synchronous buck configuration, a 5.1k or smaller value
resistor has to be added to connect LGATE to ground to avoid
falsely turn-on of LGATE caused by coupling noise.
Where Rt is the gain of the current amplifier.
Output Voltage Feedback Resistor Divider
Current Sampling Transfer Function He(S)
The output voltage can be programmed down to 0.8V by a
resistor divider from VOUT to FB, according to Equation 8.
In current loop, the current signal is sampled every switching
cycle. It has the following transfer function in Equation 11:
R UP 

V OUT = 0.8   1 + -----------------
R LOW

P
2
(EQ. 8)
(EQ. 11)
S
S
H e  S  = ------- + --------------- + 1
2  Q
n n
n
Where Qn and n are given by Q n = – 2---  n = f s

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ISL78206
Power Stage Transfer Functions
Transfer function F1(S) from control to output voltage is:
S
1 + ----------- esr
v̂ o
- = V in --------------------------------------F 1  S  = ----2
dˆ
S
S
------- + --------------- + 1
2  Q
o p
o
(EQ. 12)
S 
S
 1 + ------------ 1 + -------------

 cz1 
 cz2
v̂ comp
1
A v  S  = ----------------- = ------------------- --------------------------------------------------------SR 1 C
S -
v̂ O
 1 + --------1

 
cp
1
1
1
 cz1 = --------------- ,  cz2 = ----------------------------------  cp = -------------- R 1 + R 3 C 3
R2 C1
R3 C3
Compensator design goal:
C
1
1
Where  esr = --------------- ,Q p  R o ------o- , o = -------------------
1 1
Loop bandwidth fc:  --- to ------ f s
Transfer function F2(S) from control to inductor current is given
by Equation 13:
Gain margin: >10dB
Rc Co
LP
(EQ. 18)
LP Co
4
S
1 + -----ˆI
V

o
in
z
F 2  S  = ---= ------------------------- --------------------------------------R o + R LP 2
dˆ
S
S
------- + --------------- + 1
2  Q
o p
o
1
Where  z = -------------Ro Co .
(EQ. 13)
10
Phase margin: 45°
The compensator design procedure is as follows:
1. Position CZ2 and CP to derive R3 and C3.
Put the compensator zero CZ2 at (1 to 3)/(RoCo)
3
 cz2 = --------------Ro Co
Current loop gain Ti(S) is expressed as Equation 14:
(EQ. 19)
The voltage loop gain with open current loop is Equation 15:
Put the compensator pole CP at ESR zero or 0.35 to 0.5 times
of switching frequency, whichever is lower. In all-ceramic-cap
design, the ESR zero is normally higher than half of the switching
frequency. R3 and C3 can be derived as following:
T v  S  = KF m F 1  S A v  S 
1
Case A: ESR zero ---------------------- less than (0.35 to 0.5) fs
2R C
(EQ. 14)
T i  S  = R t F m F 2  S H e  S 
(EQ. 15)
The Voltage loop gain with current loop closed is given by
Equation 16:
Tv  S 
L v  S  = -----------------------1 + Ti  S 
(EQ. 16)
c o
R o C o – 3R c C o
C 3 = ---------------------------------------3R 1
(EQ. 20)
3R c R 1
R 3 = -----------------------R o – 3R c
(EQ. 21)
If Ti(S)>>1, then Equation 16 can be simplified as Equation 17:
1
Case B: ESR zero ---------------------- larger than (0.35 to 0.5) fs
S
1 + -----------R o + R LP
 esr A v  S 
1
L v  S  = ------------------------- ---------------------- ---------------- ,  p  --------------Rt
Ro Co
S He  S 
1 + ------p
0.33R o C o f s – 0.46
C 3 = -------------------------------------------------fs R
(EQ. 22)
R1
R 3 = ----------------------------------------0.73R o C o f s – 1
(EQ. 23)
(EQ. 17)
1
Equation 17 shows that the system is a single order system.
Therefore, a simple type II compensator can be easily used to
stabilize the system. While type III compensator is needed to
expand the bandwidth for current mode control in some cases.
R2
C1
R3
C3
VO
VCOMP
VREF
2R c C o
R1
RBIAS
2. Derive R2 and C1.
The loop gain Lv(S) at cross over frequency of fc has unity gain.
Therefore, C1 is determined by Equation 24.
 R 1 + R 3 C 3
C 1 = ---------------------------------2f c R t R 1 C
(EQ. 24)
o
The compensator zero CZ1 can boost the phase margin and
bandwidth. To put CZ1 at 2 times of cross cover frequency fc is a
good start point. It can be adjusted according to specific design.
R1 can be derived from Equation 25.
(EQ. 25)
1
R 2 = -------------------4f c C 1
FIGURE 23. TYPE III COMPENSATOR
A compensator with 2 zeros and 1 pole is recommended for this
part as shown in Figure 23. Its transfer function is expressed as
Equation 18:
Example: VIN = 12V, VO = 5V, IO = 2A, fs = 500kHz,
Co = 60µF/3mΩ, L = 10µH, Rt = 0.20V/A, fc = 50kHz,
R1 = 105k, RBIAS = 20kΩ.
Where,
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FN8618.2
March 25, 2015
ISL78206
Select the crossover frequency to be 35kHz. Since the output
capacitors are all ceramics, use Equations 22 and 23 on page 16
to derive R3 to be 20k and C3 to be 470pF.
Then use Equations 24 and 25 on page 16 to calculate C1 to be
180pF and R2 to be 12.7k. Select 150pF for C1 and 15k for R2.
There is approximately 30pF parasitic capacitance between the
COMP to FB pins that contributes to a high frequency pole. Any
extra external capacitor is not recommended between COMP
and FB.
Figure 24 shows the simulated bode plot of the loop. It is shown
that it has 26kHz loop bandwidth with 70° phase margin and
H28dB gain margin.
LOOP GAIN
80
Layout Suggestions
1. Place the input ceramic capacitors as close as possible to the
IC VIN pin and power ground connecting to the power MOSFET
or diode. Keep this loop (input ceramic capacitor, IC VIN pin
and MOSFET/Diode) as tiny as possible to achieve the least
voltage spikes induced by the trace parasitic inductance.
2. Place the input aluminum capacitors close to the IC VIN pin.
3. Keep the phase node copper area small, but large enough to
handle the load current.
4. Place the output ceramic and aluminum capacitors also close
to the power stage components.
5. Put vias (20 recommended) in the bottom pad of the IC. The
bottom pad should be placed in the ground copper plane with
area as large as possible in multiple layers to effectively
reduce the thermal impedance.
60
dB
40
20
0
-20
-40
-60
100
1k
10k
FREQUENCY (Hz)
100k
1M
6. Place the 4.7µF ceramic decoupling capacitor at the VCC pin
and as close as possible to the IC. Put multiple vias (≥3) close
to the ground pad of this capacitor.
PHASE MARGIN
180
160
7. Keep the bootstrap capacitor close to the IC.
140
8. Keep the LGATE drive trace as short as possible and try to
avoid using via in LGATE drive path to achieve the lowest
impedance.
120
DEGREE
FIGURE 25. PCB VIA PATTERN
100
9. Place the positive voltage sense trace close to the load for
tighter regulation.
80
60
10. Put all the peripheral control components close to the IC.
40
20
0
100
1k
10k
FREQUENCY (Hz)
100k
1M
FIGURE 24. SIMULATED LOOP GAIN
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March 25, 2015
ISL78206
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make
sure you have the latest revision.
DATE
REVISION
CHANGE
March 25, 2015
FN8618.2
On page 6, updated Charged Device Model test method from “JESD22-C101E” to “AEC-Q100-11”.
February 18, 2014
FN8618.1
Initial Release
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FN8618.2
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ISL78206
Package Outline Drawing
M20.173A
20 LEAD HEAT-SINK THIN SHRINK SMALL OUTLINE PACKAGE (HTSSOP)
Rev 0, 8/13
A
1
3
6.50 ±0.10
6.40
PIN #1
I.D. MARK
4.40 ±0.10
2
4.20
SEE DETAIL "X"
10
20
3.00
3
EXPOSED
THERMAL PAD
0.20 C B A
1
9
B
0.65
0.09-0.20
TOP VIEW
END VIEW
BOTTOM VIEW
1.00 REF
H
- 0.05
C
0.90 +0.15/-0.10
1.20 MAX
SEATING
PLANE
GAUGE
PLANE
0.25 +0.05/-0.06 5
0.10 M C B A
0.10 C
0°-8°
0 MIN TO 0.15 MAX
SIDE VIEW
0.25
0.60 ±0.15
DETAIL "X"
(4.20)
(1.45)
NOTES:
1. Dimension does not include mold flash, protrusions or gate burrs.
Mold flash, protrusions or gate burrs shall not exceed 0.15 per side.
(5.65)
(3.00)
2. Dimension does not include interlead flash or protrusion. Interlead
flash or protrusion shall not exceed 0.25 per side.
3. Dimensions are measured at datum plane H.
4. Dimensioning and tolerancing per ASME Y14.5M-1994.
5. Dimension does not include dambar protrusion. Allowable protrusion
shall be 0.80mm total in excess of dimension at maximum material
condition. Minimum space between protrusion and adjacent lead
0.6500
(0.35 TYP)
TYPICAL RECOMMENDED LAND PATTERN
is 0.07mm.
6. Dimension in ( ) are for reference only.
7. Conforms to JEDEC MO-153.
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19
FN8618.2
March 25, 2015
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