AN2011: ZL2005 Component Selection Guide

ZL2005 Component Selection Guide
May 01, 2009
Application Note
AN2011.0
T
D = on
Tsw
Eq. [1]
28
29
SGND
V25
18 VSEN
30
XTEMP
17 VRTK
31
TACH
16 SS1
32
MGN
13 V1
15 SS0
EN
12 V0
CFG
DLY0
11 FC1
14 UVLO
PG
DLY1
10 FC0
33
The buck converter shown in Figure 1 is a well- known
switching power supply topology. It operates by
turning on and off the control MOSFET (QH) at a high
frequency. The amount of time the QH is on as a
fraction of the total switching period is known as the
duty cycle D, which is described by the following
equation where Ton is time on and Tsw is the switching
period.
34
Zilker Labs’ power management and conversion ICs
are synchronous voltage-mode buck converters based
on the patented Digital-DC™ technology. The buck
converter is used to convert a higher, often loosely
regulated voltage to a lower, tightly regulated voltage.
The buck converter uses a MOSFET in the
“freewheeling” or bottom switch location to increase
efficiency. The operation of a buck converter is
illustrated in Figure 1.
35
General Circuit Description
36
Introduction
Figure 1. Typical Application Circuit
1
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Application Note 2011
To simplify the analysis of this circuit, assume the
following:
•
Input voltage value (VIN) is constant
•
Output voltage value (VOUT) is constant
•
Average inductor current is not decreasing or
increasing
Under these assumptions, the net voltage across the
inductor during a switching period must equal zero.
This can be described by the following equations:
D (V IN − VOUT ) = (1 − D ) × VOUT
Eq. [2]
V
D = OUT
V IN
Eq. [3]
During D, QH is on and VIN–VOUT is applied across the
inductor. The current ramps up as shown in Figure 2.
ILpk
0
Io
ILv
-VOUT
D
Time
CURRENT (A)
VOLTAGE
(V)
VIN – VOUT
Buck Power Stage Losses
Each component in the power stage of the buck
converter dissipates power. The input and output
capacitors dissipate power in their equivalent series
resistances (ESR) proportional to the ripple current
flowing through them. The inductor dissipates power
in its ESR and in core material loss.
Core loss is proportional to the ripple current flowing
through the inductor and the frequency of the ripple.
The synchronous MOSFET dissipates power in two
ways: in its channel resistance (RDSON) as a function of
current and in the gate drive current needed to turn the
MOSFET on and off.
The gate drive current loss is proportional to
frequency. Likewise, the control MOSFET also
dissipates power in its RDSON and gate drive current as
well as in its turn-on and turn-off transitions.
The power dissipated in these transitions, called
switching loss, is proportional to frequency. Because
many of the power stage component losses are
proportional to frequency, increasing frequency
increases power loss and thus lowers efficiency.
1-D
Figure 2. Inductor Waveforms
When the top MOSFET turns off, the current flowing
in the inductor must continue to flow from the ground
up through the synchronous MOSFET (QL), after
which the current ramps down. The output capacitor
COUT is selected with a low impedance at the switching
frequency so the AC component of the inductor current
is filtered from the output voltage and the load sees a
nearly DC voltage.
2
The input capacitor CIN is likewise sized to source the
AC component of the current flowing through the top
MOSFET to prevent this AC current from being drawn
from elsewhere in the system.
Buck Converter LC Filter
The LC filter smoothes the chopped input voltage
generated by QH and QL to provide a low noise DC
output voltage. The size of this filter is inversely
proportional to the switching frequency. The inductor
core loss also increases with frequency, so there is a
trade-off between a small output filter made possible
by a higher switching frequency and getting better
power supply efficiency.
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Application Note 2011
To select the appropriate power stage components for
the desired performance goals, the power supply
requirements listed in Table 1 must be known.
Table 1. Power Supply Requirements
Range
Example
Value
3.0–14.0 V
12 V
Output voltage (VOUT)
0.6–5.0
1.2 V
Output current (IOUT)
0 to ~30 A
20 A
Output voltage ripple
(Vorip)
< 3% of VOUT
1% of VOUT
Output step load
< Io
50% of Io
Output step load rate
—
10 A/µS
Step load deviation
—
50 mV
120°C
85°C
Desired efficiency
—
85%
Desired size
—
Optimize for
small size
Parameter
Input voltage (VIN)
Maximum PCB temp.
Design Goal Trade-offs
The design of the buck power stage requires several
compromises among size, efficiency, and cost. Size
can be decreased by increasing the switching frequency
at the expense of efficiency. Cost can be minimized by
using through-hole inductors and capacitors; however
these components are physically large.
To start the design, select a frequency based on Table
2. This frequency is a starting point and may be
adjusted as the design progresses.
Table 2.
Design Considerations by Frequency
Frequency Range
Inductor Selection
When selecting an output inductor, several trade-offs
must be considered. Inductance must be sufficient to
generate a low ripple current (Iopp). Low ripple current
will allow smaller output capacitance to be used while
still achieving the desired output ripple voltage.
Because high inductance values compromise output
transient load performance, a balance must be struck
between low ripple current that allows low output
ripple and high ripple current that allows a small output
deviation during transient load steps. A good starting
point is to select the output inductor ripple equal to the
expected load transient step magnitude (Iostep):
I opp = I ostep
Eq. [4]
Now the output inductance can be calculated using the
following equation, where VINM is the maximum input
voltage:
LOUT
⎛ V
VOUT × ⎜⎜1 − OUT
⎝ VINM
=
fsw × I opp
⎞
⎟⎟
⎠
Eq. [5]
The average inductor current is equal to the maximum
output current. The peak inductor current is calculated
using the following equation where IOUT is the
maximum output current:
IL pk = I OUT +
I opp
Eq. [6]
2
Select an inductor rated for the average DC current
with a peak current rating above the peak current
computed above. Table 3 lists suggested inductor
vendors and series.
Design Considerations
200 – 400 kHz
High efficiency, larger size
400 – 800 kHz
Moderate efficiency, smaller size
800 kHz – 2 MHz
Lower efficiency, smallest size
Table 3. Suggested Inductors
Vendor
Series
Output Current
WE-HC
Up to 30 A
HC8
Up to 30 A
Vishay
IHLP5050
Up to 30 A
Vishay
IHLP2525
Up to 10 A
Wurth
Coiltronics
3
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Application Note 2011
Saturation characteristic also should be considered
when selecting an output inductor. Saturation is the
reduction of the effective inductance of the inductor as
the current through it increases.
In over-current or short-circuit conditions, the inductor
may have currents greater than 2X the normal
maximum rated output current. In such conditions, an
inductor that still provides some inductance to protect
the load and the power supply MOSFETs from
damaging currents is desirable. Consequently, gapped
ferrite inductors are not recommended because of their
rapid drop in inductance in over-current conditions.
The inductors in Table 3 are distributed gap cores that
will provide some inductance in over-current
conditions.
Once an inductor is selected, the ESR and core losses
in the inductor are calculated. Use the ESR specified in
the inductor manufacturer’s datasheet:
Power = ESR × ILrms
Eq. [6]
ILrms is given by
ILrms = I OUT 2 +
(I opp / 2)2
3
Eq. [7]
where IOUT is the maximum output current. Next,
calculate the core loss of the selected inductor. Since
this calculation is specific to each inductor and
manufacturer, refer to the chosen inductor datasheet.
Add the core loss and the ESR loss and compare the
total loss to the maximum power dissipation
recommendation in the inductor datasheet.
Output Capacitor Selection
Several trade-offs also must be considered when
selecting an output capacitor. Low ESR values are
needed to have a small output deviation during
transient load steps (Vosag) and low output voltage
ripple (Vorip). However, those capacitors with low ESR,
such as semi-stable (X5R and X7R) dielectric ceramic
capacitors, also have relatively low capacitance values.
4
For high ripple currents, a low capacitance value can
cause a significant amount of output voltage ripple.
Likewise, in high transient load steps, a relatively large
amount of capacitance is needed to minimize the
output voltage deviation while the inductor current
ramps up or down to the new steady state output
current value.
As a starting point, apportion one-half of the output
voltage ripple voltage to the capacitor ESR and the
other half to capacitance, as shown in the following
equations:
C OUT =
ESR =
I opp
8 fs ∗ VOUT ∗ Vorip / 2
VOUT ∗ Vorip / 2
I opp
Eq. [8]
Eq. [9]
Use these values to make an initial capacitor selection.
Some sample capacitor types are listed in Table 4.
Select a single capacitor or parallel several capacitors
to meet the ESR and capacitance requirement. Ceramic
capacitors provide the lowest ESR and are the smallest
size, but have relatively small capacitance values.
Organic semiconductor capacitors, also known as OSCON capacitors, have relatively low ESR with
relatively large capacitance values, but are larger than
ceramics and have more series inductance, which can
degrade their performance. Tantalum capacitors have
greater ESR, but with larger capacitance values than
ceramics. Tantalum capacitors must be surge-tested for
use in power supplies. After a capacitor has been
selected, the resulting output voltage ripple can be
calculated using
Vorip = I opp ∗ ESR +
I opp
8 fs ∗ COUT
Eq. [10]
Because each part of this equation was made to be less
than or equal to half of the allowed output ripple
voltage, the Vorip should be less than the desired
maximum output ripple.
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The performance of the power supply in response to a
transient load is quantified by
Vostep = I ostep ∗
2t nlr + tl out
+ 0.02 ∗ VOUT
2COUT
Eq. [11]
where tlout represents the time needed to ramp the
current in the output inductor, as given by
tlout =
I ostep ∗ LOUT
Eq. [12]
VINM − VOUT
and where tnlr is the delay of the non-linear response
circuit expressed as
t nlr = 1 / 16 fs
Eq. [13]
These equations highlight the ZL2005’s non-linear
response (NLR) circuit. The NLR allows the ZL2005
to respond quickly to a load transient to minimize the
amount of output capacitance needed.
Regardless of the type of output capacitors selected,
the NLR will be activated when the output deviates 2%
from the desired output voltage. Once activated, the
NLR bypasses the pulse width modulator in the
ZL2005 and immediately alters the duty cycle to
correct for the output deviation.
The capacitors selected should be checked against the
NLR equations. If the step response specification is
met by a large margin, the inductor ripple current can
be lowered by increasing the value of the inductor and
thus lowering the output capacitance required.
Conversely, if the step response requirement is not
met, either the frequency or the inductor ripple current
must be increased.
MOSFET Operation in a
Synchronous Buck Converter
The on-time of the control MOSFET QH sets the
conversion ratio from the input voltage to the output
voltage. When QH is off, the inductor current
continues to flow in the synchronous MOSFET QL.
To avoid a short circuit across the input voltage supply,
the ZL2005 must ensure that QH and QL are not on at
the same time. When QH and QL are both on, the
condition is called cross conduction. When both QH
and QL are off, the condition is called dead time.
During the dead time, the inductor current must flow in
the parasitic drain diode in QL. The voltage drop and
the resulting power loss in this diode are greater than
what would occur if the current were flowing in the
drain of QL. Therefore, the dead time should be
minimized, but not to the extent that the MOSFETs
cross-conduct.
The ZL2005 incorporates a unique algorithm that
continuously optimizes the MOSFET dead time based
on the efficiency of the power stage. Even with
optimized dead time, there is always some delay
between when either QH or QL is turned off and QH or
QL is turned on. As a result, QL is always turned on
and off with current flowing in the drain diode.
Because the voltage across the drain diode is smaller
than the supply input voltage, QL is turned on and off
with very little loss. QH, however, is turned off and on
with nearly the full input supply voltage applied from
its drain to its source. Therefore, QH experiences
losses both when it is turned on and turned off.
Table 4. Suggested Capacitors
Type
Vendor
Series
Capacitance
ESR
TDK
C3216X5R
47 µF
2.5 mΩ
Organic Semiconductor
Vishay
255D
220 µF
40 mΩ
Tantalum
Vishay
593D
100 µF
100 mΩ
Ceramic
5
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Application Note 2011
using the equation for QL above. As was done with
QL, calculate the RMS current as follows:
QL Selection
The bottom MOSFET should be selected primarily
based on the device’s RDSON and secondarily based on
its gate charge. To choose QL, use the following
equation and allow 2–5% of the output power to be
dissipated in the RDSON of QL (lower output voltages
and higher step-down ratios will be closer to 5%):
PQL = (0.02 ~ 0.05) × VOUT × I OUT
Eq. [14]
Calculate the RMS current in QL as follows:
I botrms =
(1 − D)
1
∗ (3I OUT 2 + I opp 2 )
3
4
Eq. [15]
Calculate the desired RDSON as follows:
RDSON = PQL / Ibotrms 2
Eq. [17]
Keep in mind that the total allowed gate drive current
for both QH and QL is 80 mA.
MOSFETs with lower RDSONs tend to have higher gate
charge requirements, which increases the current and
resulting power required to turn them on and off. Since
the MOSFET gate drive circuits are contained in the
ZL2005, this power is dissipated in the ZL2005
according to the following equation:
Eq. [18]
QH Selection
In addition to the RDSON loss and gate charge loss, QH
also has switching loss. The procedure to select QH is
similar to the procedure for QL. First, assign 2–5% of
the output power to be dissipated in the RDSON of QH
6
Eq. [19]
PQH = (0.02 ~ 0.05) × VOUT × I OUT
Eq. [20]
RDSON = PQH / Itoprms 2
Eq. [21]
Select a candidate MOSFET, and calculate the
resulting gate drive current. Verify that the combined
gate drive current from QL and QH does not exceed 80
mA.
Next, calculate the switching time using
Select a candidate MOSFET, and calculate the required
gate drive current as follows:
Power = fs ∗ Q g ∗ V IN
D
1
∗ (3IOUT 2 + I opp 2 )
3
4
Calculate a starting RDSON as follows:
Eq. [16]
Note that the RDSON given in the manufacturer’s
datasheet is measured at 25°C. The actual RDSON in the
end-use application will be much higher. For example,
a Vishay Si7114 MOSFET with a junction temperature
of 125°C has an RDSON 1.4X higher than the value at
25°C.
I g = fs ∗ Q g
I toprms =
tsw =
VINM ∗ C gd
Eq. [22]
I gdr
where Cgd is the gate to drain capacitance of the
selected QH and Igdr is the peak gate drive current
available from the ZL2005.
Although the ZL2005 has a typical gate drive current
of 4 A, use the minimum guaranteed current of 2 A for
a conservative design. Using the calculated switching
time, calculate the switching power loss in QH using
Pswtop = VINM ∗ t sw ∗ I OUT ∗ fs
Eq. [23]
The total power dissipated by QH is given by the
following equation:
PQHtot = PQH + Pswtop
Eq. [24]
MOSFET Thermal Check
Once the power dissipations for QH and QL have been
calculated, the MOSFETs junction temperature can be
estimated. Using the junction-to-case thermal
resistance (Rth) given in the MOSFET manufacturer’s
datasheet and the expected maximum printed circuit
board temperature, calculate the junction temperature
as follows:
Tj max = T pcb + PQ ∗ Rth
Eq. [25]
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Application Note 2011
Several examples of MOSFETs that can be used for
QH and QL are listed in
For further details of thermal analysis and design see
Zilker Labs Application Note 10 [1].
Table 5. MOSFETS for QH and QL
The input capacitors should be rated at 1.4X the ripple
current calculated above to assure a 50% power
derating. Ceramic capacitors with X7R or X5R
dielectric with low ESR and 1.1X the maximum
expected input voltage are recommended.
.
Input Capacitor
It is highly recommended that dedicated input
capacitors be used in any point-of-load design, even
when the supply is powered from a heavily filtered 5 or
12 V “bulk” supply from an off-line, computer-grade
power supply. This is because of the high RMS ripple
current that is drawn by the buck converter topology.
This ripple can be determined from the following
equation, where η is the converter efficiency, usually
around 90%:
⎛V
I rms = I OUT ∗ ⎜⎜ OUT
⎝ VINM
⎞ ⎡ ⎛ VOUT
⎟⎟ ⎢1 + ⎜⎜
⎠ ⎣ ⎝ VINM
CB Selection
This capacitor is the filter for the bootstrap bias voltage
that is used to drive the top MOSFET. It should be
large enough to minimize voltage drop while providing
drive current between charging pulses as a function of
the drop and gate charge of the top FET. This is done
by sizing CB such that the charge needed to turn on the
top MOSFET is less than 1/100 the charge stored in
CB. Using the gate charge specified in the MOSFET
manufacturer’s datasheet and the relationship Q = C ×
V,
⎞⎛ 1 − 2η ⎞⎤
⎟⎟⎜⎜ 2 ⎟⎟⎥
⎠⎝ η ⎠⎦
Eq. [26]
CB =
Without capacitive filtering near the power supply
circuit, this current would flow through the supply bus
and return planes, coupling noise into other system
circuitry. Furthermore, filter capacitors on the input
bus may not be rated to handle this much ripple
current, and heating of these devices could lead to
premature failure.
100Q g
Eq. [27]
4.5 V
Table 5. MOSFETS for QH and QL
Part Number
Mfg
RDSON (mΩ)
VDS
Qg (nC)
Case
IRF6609
IR
2.6
20
46
MT
IRF6620
IR
3.6
20
28
MX
IRF6637
Vishay
7.7
20
11
MP
IRF6617
IR
8.1
20
11
ST
Vishay
5.3
20
40
DPAK
IR
5.5
20
29
SO8
Si7106DN
Vishay
6.2
20
18
PP1212
Si7406DH
Vishay
65
20
5
SC70-6
Si5404DC
Vishay
30
20
12
S1206-8
Si2312DS
Vishay
31
20
8
SOT23
Si6404DQ
Vishay
10
30
32
TSSOP8
ON
3.2
30
47
PPSO8
SUD70N02_03P
IRF7834
NTMFS4108N
7
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Application Note 2011
As an example, a 10 nC MOSFET would need CB =
0.2 µF. CB should be a stable dielectric such as X7R
and rated 6.3 V or more.
CV25 Selection
This capacitor is used to both stabilize and provide
noise filtering for the 2.5 V internal power supply. It
should be between 4.7 and 10 µF, should use a semistable X5R or X7R dielectric ceramic with a low (less
than 10 mΩ) ESR, and should have a rating of 4 V or
more.
CVR Selection
This capacitor is used to both stabilize and provide
noise filtering for the 5 V reference supply (VR). It
should be between 2.2 and 10 µF, be a semi-stable
X5R or X7R dielectric ceramic capacitor with a low
ESR less than 10 mΩ, and be rated 6.3 V or more.
Because the current for the bootstrap supply is drawn
from this capacitor, CVR should be sized at least 10X
the value of CB so that a discharged CB does not cause
the voltage on it to drop excessively during a CB
recharge pulse.
DB Selection
This Schottky diode is used to provide the bootstrap
voltage. The bootstrap voltage is a floating bias supply
used to drive the top MOSFET. It can be any Schottky
diode rated for 200 mA average, 500 mA peak, with
less than a 0.5 V forward drop and reverse voltage
rating of at least 30 V. The BAT54C in the SOT23 or
SC75 package is recommended. The anodes of this
device can be connected together as shown in Figure 1.
8
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Application Note 2011
Revision History
References
[1]
AN2010 – ZL2005 Thermal and
Guidelines, Zilker Labs, Inc., 2005.
9
Layout
Date
Rev. #
9/30/05
0.8
Initial release
9/25/06
1.0
Changed Eq. [22], Eq.
[26], and CV25 Selection
5/01/09
AN2011.0
Assigned file number
AN2011 to app note as
this will be the first release
with an Intersil file
number. Replaced header
and footer with Intersil
header and footer.
Updated disclaimer
information to read
“Intersil and it’s
subsidiaries including
Zilker Labs, Inc.” No
changes to application
note content.
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Application Note 2011
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