AN6668: Applications of the CA3080 and CA3080A High-Performance Operational Transconductance Amplifiers

Applications of the CA3080 High-Performance
Operational Transconductance Amplifiers
Application Note
November 1996
AN6668.1
Introduction
The CA3080 and CA3080A are similar in generic form to
conventional operational amplifiers, but differ sufficiently to
justify an explanation of their unique characteristics. This
new class of operational amplifier not only includes the usual
differential input terminals, but also contains an additional
control terminal which enhances the device's flexibility for
use in a broad spectrum of applications. The amplifier
incorporated in these devices is referred to as an
Operational Transconductance Amplifier (OTA), because its
output signal is best described in terms of the output-current
that it can supply:
Circuit Description
A simplified block diagram of the OTA is shown in Figure 2.
Transistors Q1 and Q2 comprise the differential input amplifier
found in most operational amplifiers, while the lettered-circles
(with arrows leading either into or out of the circles) denote
“current-mirrors”. Figure 3A shows the basic type of currentmirror which is comprised of two transistors, one of which is
diode-connected. In a current-mirror with similar geometries
for QA and QB, the current I’ establishes a second current I
whose value is essentially equal to that of I’.
∆i OUT
Transconductance g M = ----------------∆e IN
V+
The amplifier's output-current is proportional to the voltage
difference at its differential input terminals.
This Application Note describes the operation of the OTA
and features various circuits using the OTA. For example,
communications and industrial applications including modulators, multiplexers, sample-and-hold-circuits, gain control
circuits and micropower comparators are shown and discussed. In addition, circuits have been included to show the
operation of the OTA being used in conjunction with CMOS
devices as post-amplifiers.
Figure 1 shows the equivalent circuit for the OTA. The output
signal is a current which is proportional to the transconductance (gM) of the OTA established by the amplifier bias current (IABC) and the differential input voltage (eIN). The OTA
can either source or sink current at the output terminal,
depending on the polarity of the input signal.
V+
7
2
OTA
RIN
eIN
2RO
6
gM x eIN
IOUT = gM(±eIN)
2RO
3
7
Y
INVERTING
INPUT
2
AMPLIFIER
BIAS CURRENT
5
IABC
Z
Q1 Q2
3
OUTPUT
NON-INVERTING
INPUT
6
W
X
4
V-
FIGURE 2. SIMPLIFIED DIAGRAM OF THE OTA
This basic current-mirror configuration is sensitive to the
transistor beta (β). The addition of another active transistor,
shown in Figure 3B, greatly diminishes the circuit sensitivity
to transistor beta and increases the current-source output
impedance in direct proportion to the transistor beta.
Current-mirror W (Figure 2) uses the configuration shown in
Figure 3A, while mirrors X, Y, and Z are basically the version
shown in Figure 3B. Mirrors Y and Z employ PNP transistors,
as depicted by the arrows pointing outward from the mirrors.
Appendix 1 describes current-mirrors in more detail.
+
gM (mS) = 19.2 IABC (mA)
RO (MΩ) ≈ 7.5/IABC (mA)
4
5 VIABC
FIGURE 1. BASIC EQUIVALENT CIRCUIT OF THE OTA
The availability of the amplifier bias current (IABC) terminal
significantly increases the flexibility of the OTA and permits
the circuit designer to exercise his creativity in the utilization
of this device in many unique applications not possible with
the conventional operational amplifier.
1
I’
I
QA
I’
I
QB
QB
QA
V-
V-
FIGURE 3A. DIODE-CONNECTED TRANSISTOR PAIRED WITH
TRANSISTOR
1-888-INTERSIL or 321-724-7143 | Copyright
© Intersil Corporation 1999
Application Note 6668
I’
I
I’
Transistors Q10, Q11, and diode D6 of Figure 4 comprise the
current-mirror “X” of Figure 2. Diodes D2 and D4 are connected
across the base-emitter junctions of Q5 and Q8, respectively, to
improve the circuit speed. The amplifier output signal is derived
from the collectors of the “Z” and “X” current-mirror of Figure 2,
providing a push-pull Class A output stage that produces full
differential gM. This circuit description applies to both the
CA3080 and CA3080A. The CA3080A offers tighter control of
gM and input offset voltage, less variation of input offset voltage
with variation of IABC and controlled cut-off leakage current. In
the CA3080A, both the output and the input cut-off leakage
resistances are greater than 1,000MΩ.
I
I
I
V-
V-
FIGURE 3B. IMPROVED VERSION: EMPLOYS AN EXTRA
TRANSISTOR
FIGURE 3. BASIC TYPES OF CURRENT MIRRORS
10kΩ
Figure 4 is the complete schematic diagram of the OTA. The
OTA employs only active devices (transistors and diodes).
Current applied to the amplifier-bias-current terminal, IABC,
establishes the emitter current of the input differential
amplifier Q1 and Q2 . Hence, effective control of the
differential transconductance (gM) is achieved.
V+
2
10kΩ
CHANNEL #1
INPUT
V+
7
D3
3
-
OTA
AMP 1
CA3080
+
4
5
V-
D5
150pF
Q6
INVERTING
INPUT
2
Q1
D4
Q9
10kΩ
CHANNEL #2
INPUT
3
Q2
OUTPUT
6
NONINVERTING
INPUT
3
TO TERM 5
AMP 1
Q10
AMPLIFIER
BIAS-CURRENT
-
7
OTA
AMP 2
CA3080
+
4
5
V-
6
V+ = 5V
V- = -5V
(NO SUPPLY BYPASSING
SHOWN)
IABC
V+
Q3
5
VABC
620Ω
V+
2
Q8
MULTIPLEXED
OUTPUT
IABC
Q7
Q5
6
10kΩ
Q4
D2
7
Q11
D1
D6
V4
FIGURE 4. SCHEMATIC DIAGRAM OF OTA TYPES CA3080
AND CA3080A
The gM of a differential amplifier is equal to:
qαI C
------------2KT
(see Reference 2 for derivation) where q is the charge on an
electron, α is the ratio of collector current to emitter current
of the differential amplifier transistors, (assumed to be 0.99
in this case), IC is the collector current of the constantcurrent source (IABC in this case), K is Boltzman's constant,
and T is the ambient temperature in degrees Kelvin. At room
temperature, gM = 19.2 x IABC, where gM is in mS and IABC
is in milliamperes. The temperature coefficient of gM is
approximately -0.33%/oC (at room temperature).
Transistor Q3 and diode D1 (shown in Figure 4) comprise the
current mirror “W” of Figure 2. Similarly, transistors Q7, Q8 and
Q9 and diode D5 of Figure 4 comprise the generic current mirror “Z” of Figure 2. Darlington-connected transistors are
employed in mirrors “Y” and “Z” to reduce the voltage sensitivity
of the mirror, by the increase of the mirror output impedance.
2
CLOCK
INPUT
Q 36kΩ
DTL OR T2L
FLIP-FLOP
2N4037
Q 36kΩ
FIGURE 5. SCHEMATIC DIAGRAM OF OTAs IN A TWOCHANNEL LINEAR TIME-SHARED MULTIPLEXER
CIRCUIT
Applications
Multiplexing
The availability of the bias current terminal, IABC, allows the
device to be gated for multiplexer applications. Figure 5 shows
a simple two-channel multiplexer system using two CA3080
OTA devices. The maximum level-shift from input to output is
low (approximately 2mV for the CA3080A and 5mV for the
CA3080). This shift is determined by the amplifier input offset
voltage of the particular device used, because the open-loop
gain of the system is typically 100dB when the loading on the
output of the CA3080A is low. To further increase the gain and
reduce the effects of loading, an additional buffer and/or gainstage may be added. Methods will be shown to successfully
perform these functions.
Application Note 6668
In this example ±5V power-supplies were used, with the IC flipflop powered by the positive supply. The negative supply-voltage may be increased to -15V, with the positive-supply at 5V to
satisfy the logic supply voltage requirements. Outputs from the
clocked flip-flop are applied through PNP transistors to gate the
CA3080 amplifier-bias-current terminals. The grounded-base
configuration is used to minimize capacitive feed-through coupling via the base-collector junction of the PNP transistor.
Another multiplexer system using the OTAs clocked by a CMOS
flip-flop is shown in Figure 6. The high output voltage capability
of the CMOS flip-flop permits the circuit to be driven directly
without the need for PNP level-shifting transistors.
V+
10kΩ
7
2
10kΩ
3
10kΩ
OTA
AMP 1
+
4
5
V-
6
MULTIPLEXED
OUTPUT
150pF
7 V+
2
10kΩ
3
V+ = 10V, V- = -10V
(NO SUPPLY BYPASSING
SHOWN)
14
0V
CLOCK
INPUT
-10V
2kΩ
7 6
6
TO TERM 5
AMP 1
4
VDD VSS
CL
Q
1/2 CD4013A
5
D
Q
3
620Ω
FIGURE 7. VOLTAGE WAVEFORMS FOR CIRCUIT OF FIGURE 6
OTA
AMP 2
+
4
5
V-
V-
Top Trace: Multiplexed Output; 1V/Div., 100µs/Div.
Bottom Trace: Time Expansion of Switching Between
Inputs; 2V/Div., 5µs/Div.
1
IABC
82kΩ
TO TERM 5
AMP 2
IABC
82kΩ
2
FIGURE 6. SCHEMATIC DIAGRAM OF A TWO-CHANNEL
LINEAR MULTIPLEXER SYSTEM USING A CMOS
FLIP-FLOP TO GATE TWO OTAs
A simple RC phase-compensation network is used on the
output of the OTA in the circuits shown in Figures 5 and 6.
The values of the RC-network are chosen so that:
Top Trace: Output; 1V/Div., 100µs/Div.
Bottom Trace: Voltage Expansion of Output; 1mV/Div., 100µs/Div.
FIGURE 8. VOLTAGE WAVEFORMS FOR CIRCUIT OF FIGURE 6
1
---------------- ≅ 2MHz.
2πRC
Sample-and-Hold Circuits
This RC network is connected to the point shown because the
lowest-frequency pole for the system is usually found at this
point. Figure 7 shows an oscilloscope photograph of the multiplexer circuit functioning with two input signals. Figure 8 shows
an oscilloscope photograph of the output of the multiplexer with
a 6VP-P, sine wave signal (22kHz) applied to one amplifier and
the input to the other amplifier grounded. This photograph demonstrates an isolation of at least 80dB between channels.
3
An extension of the multiplex system application is a sampleand-hold circuit (Figure 9), using the strobing characteristics
of the OTA amplifier bias-current (ABC) terminal as a means
of control. Figure 9 shows the basic system using the
CA3080A as an OTA in a simple voltage-follower configuration
with the phase-compensation capacitor serving the additional
function of sampled-signal storage. The major consideration
for the use of this method to “hold” charge is that neither the
charging amplifier nor the signal readout device significantly
alter the charge stored on the capacitor. The CA3080A is a
particularly suitable capacitor-charging amplifier because its
output resistance is more than 1000MΩ under cut-off conditions, and the loading on the storage capacitor during the
Application Note 6668
hold-mode is minimized. An effective solution to the read-out
requirement involves the use of a 3N138 insulated-gate fieldeffect transistor (MOSFET) in the feedback loop. This transistor has a maximum gate-leakage current of 10pA; its loading
on the charge “holding” capacitor is negligible. The open-loop
voltage-gain of the system (Figure 9) is approximately 100dB
if the MOSFET is used in the source-follower mode with the
CA3080A as the input amplifier. The open-loop output impedance (1/gM) of the 3N138 is approximately 220Ω because its
transconductance is about 4,600µS at an operating current of
5mA. When the CA3080A drives the 3N138, the closed loop
operational-amplifier output impedance characteristic is:
Z O ( OPEN-LOOP )
Z OUT ≅ ------------------------------------------------------------------------------------------A ( OPEN-LOOP VOLTAGE-GAIN )
220Ω 220Ω
≅ ----------------- ≅ --------------- ≅ 0.0022Ω
5
100dB
10
Top Trace: Sampled Signal 1V/Div., 20µs/Div.
Center Trace: Top Portion of Upper Signal; 1V/Div., 2µs/Div.
Bottom Trace: Sampling Signal; 20V/Div., 20µs/Div.
2.0kΩ
2
OTA
CA3080
INPUT
2.0kΩ
0.01µF
7
-
3
+
4
30kΩ
-15V
HOLD
3N138
OUTPUT
6
R
5
SAMPLE
0V
FIGURE 10. WAVEFORMS FOR CIRCUIT OF FIGURE 9
V+ =15V
120pF
0.01µF
220Ω
300pF
C
3kΩ
IABC
480µA
STORAGE AND PHASE
COMPENSATION
NETWORK
V- = -15V
Figure 12 shows a dual-trace photograph of a triangular
signal being “sampled-and-held” for approximately 14ms
with a 300pF storage capacitor. The center trace (expanded
to 20mV/Div.) shows the worst-case “tilt” for all the steps
shown in the upper trace. The total equivalent leakage
current in this case is only 170pA (I = C dv/dt).
Figure 13 is an oscilloscope photograph of a ramp voltage
being sampled by the “sample-and-hold” circuit of Figure 9. The
input signal and sampled-output signal are superimposed. The
lower trace shows the sampling signal. Data shown in Figure
13 were recorded with supply voltages of ±10V and the series
input resistor at terminal 5 was 22kΩ.
FIGURE 9. SCHEMATIC DIAGRAM OF OTA IN A SAMPLEAND-HOLD CIRCUIT
Once the signal is acquired, variation in the stored-signal level
during the hold-period is of concern. This variation is primarily a
function of the cutoff leakage current of the CA3080A (a maximum limit of 5nA), the leakage of the storage element, and
other extraneous paths. These leakage currents may be either
“positive” or “negative” and, consequently, the stored-signal
may rise or fall during the “hold” interval. The term “tilt” is used
to describe this condition. Figure 11 shows the expected pulse
“tilt” in microvolts versus time for various values of the compensation/storage capacitor. The horizontal axis shows three
scales representing leakage currents of 50nA, 5nA, 500pA.
4
100K
HOLD PERIOD (µs)
Figure 10 shows a “sampled” triangular signal. The lower
trace in the photograph is the sampling signal. When this
signal goes negative, the CA3080A is cutoff and the signal is
“held” on the storage capacitor, as shown by the plateaus on
the triangular waveform. The center trace is a time
expansion of the top-most transition (in the upper trace) with
a time scale of 2µs/Div.
1000K
10K
1K
100
10
1
1
10
100
C=
µF
10
F
3µ
F
1µ F
µ
0.3 F
µ
0.1
F
3µ
.
0 0 µF
1
0
0.
pF
00
30 0pF
0
10
F
0p
30 pF
0
0
1
pF
30 F
p
10
10
100
1K
100
1K
10K
1K
10K
100K
THIS SCALE FOR
500pA
5nA
50nA
LEAKAGE
10K
100K
1000K
PULSE TILT (µV)
FIGURE 11. “TILT” IN “HELD” VOLTAGE vs HOLD TIME
Application Note 6668
Top Trace: Input and Sampled Output Superimposed;
100mV/Div., 100ns/Div.
Bottom Trace: Sampling Signal; 20V/Div., 100ns/Div.
Top Trace: Sampled Signal; 1V/Div., 20ms/Div.
Center Trace: Worse Case Tilt; 20mV/Div., 20ms/Div.
FIGURE 12. “TRIANGULAR-VOLTAGE” BEING SAMPLED BY
CIRCUIT OF FIGURE 9
FIGURE 14. “TRIANGULAR-VOLTAGE” BEING SAMPLED BY
CIRCUIT OF FIGURE 9
Figure 15 shows the basic circuit of Figure 9 implemented with
a 2N4037 PNP transistor to minimize capacitive feedthrough.
Figure 16 shows oscilloscope photographs taken with the circuit of Figure 15 operating in the sampling mode at supply voltage of ±15V. The 9.1kΩ resistor in series with the PNP
transistor emitter establishes amplifier-bias-current (IABC) conditions similar to those used in the circuit of Figure 9.
2.0kΩ
V+ = 15V
120pF
2
OTA
CA3080
INPUT
2.0kΩ
7
-
3
+
Top Trace: Input and Output Superimposed; 1V/Div., 2µs/Div.
Bottom Trace: Sampling Signal; 20V/Div., 2µs/Div.
FIGURE 13. “RAMP-VOLTAGE” BEING SAMPLED BY CIRCUIT
OF FIGURE 9
DTL/TTL CONTROL
LOGIC
5V
2N4037
OUTPUT
6
4
C
0.01µF
5
3N138
0.01µF
68Ω
300pF
3kΩ
IABC
STORAGE AND PHASE
COMPENSATION
NETWORK
0V
9.1kΩ
V- = -15V
In Figure 14, the trace of Figure 13 has been expanded
(100mV/Div. and 100ns/Div.) to show the response of the
sample-and-hold circuit with respect to the sampling signal.
After the sampling interval, the amplifier overshoots the
signal level and settles (within the amplifier offset voltage) in
approximately 1µs. The resistor in series with the 300pF
phase-compensation capacitor was adjusted to 68Ω for minimum recovery time.
5
FIGURE 15. SCHEMATIC DIAGRAM OF THE OTA IN A SAMPLEAND-HOLD CONFIGURATION (DTL/TTL CONTROL
LOGIC)
Application Note 6668
Top Trace: Input and Sampled Output Superimposed;
100mV/Div., 100ns/Div.
Bottom Trace: Sampling Signal; 5V/Div., 100ns/Div.
Top Trace: Output; 5V/Div., 2µs/Div.
Center Trace: Differential Comparsion of Input and
Output; 2mV/Div., 0V thru Center; 2µs/Div.
Bottom Trace: Input; 5V/Div., 2µs/Div.
FIGURE 16. CIRCUIT OF FIGURE 15 OPERATING IN SAMPLING
MODE
Figure 17 shows a multi-trace oscilloscope photograph of
input and output signals for the circuit of Figure 9, operating
in the linear mode. The lower portion of the photograph
shows the input signal, and the upper portion shows the
output signal. The amplifier slew-rate is determined by the
output current and the capacitive loading: in this case the
slew rate (dv/dt) = 1.8V/µs.
The center trace in Figure 17 shows the difference between
the input and output signals as displayed on a Tektronix 7A13
differential amplifier at 2mV/Div. The output of the amplifier
system settles to within 2mV (the offset voltage specification
for the CA3080A) of the input level in 1µs after slewing.
Figure 18 is a curve of slew-rate versus amplifier-bias-current (IABC) for various storage/compensation capacitors. The
magnitude of the current being supplied to the storage/compensation capacitor is equal to the amplifier-bias-current
(IABC) when the OTA is supplying its maximum output current.
Gain Control - Amplitude Modulation
Effective gain control of a signal may be obtained by
controlled variation of the amplifier-bias-current (IABC) in the
OTA because its gM is directly proportional to the amplifierbias-current (IABC). For a specified value of amplifier-bias-current, the output current (IO) is equal to the product of gM and
the input signal magnitude. The output voltage swing is the
product of output current (IO) and the load resistance (RL).
6
FIGURE 17. CIRCUIT OF FIGURE 9 OPERATING IN THE
LINEAR SAMPLE MODE
Figure 19 shows the configuration for this form of basic gain
control (a modulation system). The output signal current (IO)
is equal to -gM x VX; the sign of the output signal is negative
because the input signal is applied to the inverting input
terminal of the OTA. The transconductance of the OTA is controlled by adjustment of the amplifier bias current, IABC. In this
circuit the level of the unmodulated carrier output is established by a particular amplifier-bias-current (IABC) through
resistor RM. Amplitude modulation of the carrier frequency
occurs because variation of the voltage VM forces a change in
the amplifier-bias-current (IABC) supplied via resistor RM.
When VM goes positive, the bias current increases which
causes a corresponding increase in the gM of the OTA. When
the VM goes in the negative direction (toward the amplifierbias-current terminal potential), the amplifier-bias-current
decreases, and reduces the gM of the OTA.
100
10
SLEW RATE (V/µs)
Considerations of circuit stability and signal retention require
the use of the largest possible phase-compensation capacitor,
compatible with the required slew rate. In most systems the
capacitor is chosen for the maximum allowable “tilt” in the storage mode and the resistor is chosen so that 1/2πRC ≅ 2MHz,
corresponding to the first pole in the amplifier at an output current level of 500µA. It is frequently desirable to optimize the system response by the placement of a small variable resistor in
series with the capacitor, as is shown in Figures 9 and 15. The
120pF capacitor shunting the 2kΩ resistor improves the amplifier transient response.
1.0
C
0.1
0.01
0.001
0.1
F
0p
=1 F
p
30 pF
0
10 0pF
30 0pF
0
pF
10
00
F
30
1µ
0.0 03µF
0.
µF
0.1
1
10
100
AMPLIFIER BIAS CURRENT (IABC µA)
1000
FIGURE 18. SLEW RATE vs AMPLIFIER-BIAS-CURRENT (IABC)
Application Note 6668
CARRIER
FREQUENCY
VX
+6V
2
51Ω
7
OTA
CA3080A
51Ω
3
47kΩ
MODULATING VFREQUENCY
VM
+
4
5
IABC
-6V
IO
IO = gM VX RL
AMPLITUDE
MODULATED
OUTPUT
NPN transistor-array as an input emitter-follower, with the
three remaining transistors of the transistor-array connected
as a current-source for the emitter followers.
VX
+6V
6
3
47kΩ
RM
7
-
AM
OUTPUT
OTA
CA3080A
51Ω
V+
100kΩ
2
51Ω
5.1kΩ
+
6
5.1kΩ
4
5
-6V
-6V
+6V
IABC
100kΩ
VM
2N4037
FIGURE 19. AMPLITUDE MODULATOR CIRCUIT USING THE OTA
5.1kΩ
24kΩ
As discussed earlier, gM = 19.2 x IABC, where gM is in
millisiemens when IABC is in milliamperes. In this case, IABC
is approximately equal to:
+6V
FIGURE 20. AMPLITUDE MODULATOR USING OTA
CONTROLLED BY PNP TRANSISTOR
V M – ( V- )
------------------------- = I ABC
RM
VX
IO = –gM VX
+15V
2
51Ω
g M V X = ( 19.2 ) ( I ABC ) ( V X )
– 19.2 [ V M – ( V- ) ]V X
I O = -----------------------------------------------------RM
100kΩ
19.2 ( V X ) ( V- ) 19.2 ( V X ) ( V M )
I O = ------------------------------------ – -------------------------------------RM
RM
There are two terms in the modulation equation: the first
term represents the fixed carrier input, independent of VM
and the second term represents the modulation, which either
adds to or subtracts from the first term. When VM is equal to
the V- term, the output is reduced to zero.
3
+
4
6
5.1kΩ
5
47kΩ
-15V
+15V
IABC
+15V
VM
AM
OUTPUT
OTA
CA3080A
51Ω
-15V
7
-
CA3018A
10kΩ
2N4037
+15V
75kΩ
1.3MΩ
+6V
In the preceding modulation equations the term,
-15V
V ABC
( 19.2 ) ( V X ) --------------RM
FIGURE 21. AMPLITUDE MODULATOR USING OTA
CONTROLLED BY PNP AND NPN TRANSISTORS
involving the amplifier-bias-current terminal voltage (VABC)
(see Figure 4 for VABC) was neglected. This term was
assumed to be small because VABC is small compared with Vin the equation. If the amplifier-bias-current terminal is driven by
a current-source (such as from the collector of a PNP transistor), the effect of VABC variation is eliminated and transferred to
the involvement of the PNP transistor base-emitter junction
characteristics. Figure 20 shows a method of driving the amplifier-bias-current terminal to effectively remove this latter variation.
The 100kΩ potentiometer shown in these schematics is used
to null the effects of amplifier input offset voltage. This potentiometer is adjusted to set the output voltage symmetrically
about zero. Figures 22A and 22B show oscilloscope photographs of the output voltages obtained when the circuit of Figure 19 is used as a modulator for both sinusoidal and triangular
modulating signals. This method of modulation permits a range
exceeding 1000:1 in the gain, and thus provides modulation of
the carrier input in excess of 99%. The photo in Figure 22C
shows the excellent isolation (>80dB at f = 100kHz) achieved in
this modulator during the “gated-off” condition.
If an NPN transistor is added to the circuit of Figure 20 as an
emitter-follower to drive the PNP transistor, variations due to
base-emitter characteristics are considerably reduced due to
the complementary nature of the NPN base-emitter junctions. Moreover, the temperature coefficients of the two
base-emitter junctions tend to cancel one another. Figure 21
shows a configuration using one transistor in the CA3018A
7
Four-Quadrant Multipliers
A single CA3080A is especially suited for many lowfrequency, low-power four-quadrant multiplier applications.
The basic multiplier circuit of Figure 23 is particularly useful
for waveform generation, doubly balanced modulation, and
other signal processing applications, in portable equipment,
Application Note 6668
where low-power consumption is essential and accuracy
requirements are moderate. The multiplier configuration is
basically an extension of the previously discussed gaincontrolled configuration (Figure 19).
To obtain a four-quadrant multiplier, the first term of the
modulation equation (which represents the fixed carrier)
must be reduced to zero. This term is reduced to zero by the
placement of a feedback resistor (R) between the output and
the inverting input terminal of the CA3080A, with the value of
the feedback resistor (R) equal to 1/gM. The output current is
IO = gM (-VX) because the input is applied to the inverting
terminal of the OTA. The output current due to the resistor
(R) is VX/R. Hence, the two signals cancel when R = 1/gM.
The current for this configuration is:
-19.2 V X V M
I O = --------------------------------, and V M = V Y
RM
The output signal for these configurations is a current which
is best terminated by a short-circuit. This condition can be
satisfied by making the load resistance for the multiplier output very small. Alternatively, the output can be applied to a
current-to-voltage converter as shown in Figure 24.
TIME (50µs/DIV.)
Top Trace: Modulation Input (20V)
Bottom Trace: Amplitude Modulated Output; 500mV/Div.
FIGURE 22B. RESPONSE FOR TRIANGLE WAVE MODULATION
In Figure 23, the current “cancellation” in the resistor R is a
direct function of the OTA differential amplifier linearity. In the
following example, the signal excursion is limited to ±10mV to
preserve this linearity. Greater signal-excursions on the input
terminal will result in a significant departure from linear operation (which may be entirely satisfactory in many applications).
TIME (50µs/DIV.)
Top Trace: Gated Output; 1V/Div.
Bottom Trace: Voltage Expansion Of Above Signal
Showing No Residual; 1mv/Div.
FIGURE 22C. RESPONSE FOR SQUARE WAVE MODULATION
FIGURE 22. AMPLITUDE MODULATOR CIRCUIT OF FIGURE 19
WITH RM = 40kΩ, VS = ±10V
TIME (50µs/DIV.)
Top Trace: Modulation Input (≅ 20VP-P)
Center Trace: Amplitude Modulated Output; 500mV/Div.
Bottom Trace: Expanded Output to Show
Depth of Modulation; 20mV/Div.
2
VX
R = 1/gM
OTA
CA3080A
FIGURE 22A. RESPONSE FOR SINE WAVE MODULATION
3
RM
6
IO ≅ -K VX VY
+
5
IABC
VY
FIGURE 23. BASIC FOUR QUADRANT ANALOG MULTIPLIER
USING AN OTA
8
Application Note 6668
ANALOG MULTIPLIER
CA3080A
RF
IO
X
Y
VX
OP AMP
CA3741CT
OTA
EOUT = - IO RF
2
5.1Ω
7
-
OUTPUT
OTA
CA3080A
5.1Ω
3
24kΩ
100Ω
150kΩ
+15V
250kΩ
5.1kΩ
+
6
4.7MΩ
4
5
-15V
V+
-15V
V-
VY
200kΩ
FIGURE 24. OTA ANALOG MULTIPLIER DRIVING A CURRENTTO-VOLTAGE CONVERTER
Figure 25 shows a schematic diagram of the basic multiplier with
the adjustments set-up to give the multiplier an accuracy of
approximately ±7 percent full-scale. There are only three adjustments: 1) one is on the output, to compensate for slight variations
in the current-transfer ratio of the current-mirrors (which would
otherwise result in a symmetrical output about some current level
other than zero); 2) the adjustment of the 20kΩ potentiometer
establishes the gM of the system equal to the value of the fixed
resistor shunting the system when the Y-input is zero; 3) compensates for error due to input offset voltage.
Procedure for adjustment of the circuit:
1. a) Set the 1MΩ output-current balancing potentiometer
to the center of its range
b) Ground the X- and Y- inputs
c) Adjust the 100kΩ potentiometer until a 0V reading is
obtained at the output.
2. a) Ground the Y-input and apply a signal to the Xinput through a low source-impedance generator (it is
essential that a low impedance source be used; this
minimizes any change in the gM balance or zero-point
due to the 50µA Y-input bias current).
b) Adjust the 20kΩ potentiometer in series with Y-input
until a reading of 0V is obtained at the output. This
adjustment establishes the gM of the CA3080A at the
proper level to cancel the output signal. The output
current is diverted through the 510kΩ resistor.
3. a) Ground the X-input and apply a signal to the Y-input
through a low source-impedance generator.
b) Adjust the 1MΩ resistor for an output voltage of 0V.
-15V
+15V
100kΩ
+15V
1MΩ
2N4037
2.2kΩ
62kΩ
20kΩ
+15V
FIGURE 26. SCHEMATIC DIAGRAM OF ANALOG MULTIPLIER
USING OTA CONTROLLED BY A PNP TRANSISTOR
Figure 26 shows the schematic of an analog multiplier circuit
with a 2N4037 PNP transistor replacing the Y-input “current”
resistor. The advantage of this system is the higher input
resistance resulting from the current-gain of the PNP
transistor. The addition of another emitter-follower preceeding the PNP transistor (shown in Figure 21) will further
increase the current gain while markedly reducing the effect
of the Vbe temperature-dependent characteristic and the
offset voltage of the two base-emitter junctions.
Figures 27A and 27B show oscilloscope photographs of the
output signals delivered by the circuit of Figure 26 which is connected as a suppressed-carrier generator. Figures 28A and
28B contain photos of the outputs obtained in signal “squaring”
circuits, i.e. “squaring” sine-wave and triangular- wave inputs.
If ±15V power supplies are used (shown in Figure 26), both
inputs can accept ±10V input signals. Adjustment of this
multiplier circuit is similar to that already described above.
There will be some interaction among the adjustments and
the procedure should be repeated to optimize the circuit
performance.
VX
+6V
510kΩ
5.1kΩ
10Ω
2
7
-
OTA
CA3080A
10Ω
3
24kΩ
+
OUTPUT
6
3.3MΩ
4
5
-6V
-6V
VY
+6V
-6V
100kΩ
20kΩ
IABC
+6V
1MΩ
91kΩ
FIGURE 25. SCHEMATIC DIAGRAM OF ANALOG MULTIPLIER
USING OTA
9
500mV/Div., 200µs/Div.,
Triangular Input: 700Hz; 5VP-P to VY Input
Carrier Input: 30kHz; 13.5VP-P to VX Input
FIGURE 27A.
Application Note 6668
0V
0V
500mV/Div., 200µs/Div.,
Modulating Frequency: 700Hz; 5VP-P to VY Input
Carrier Input: 21kHz; 13.5VP-P to VX Input
Top Trace: Input to X And Y; 2V/Div., 1ms/Div. (200Hz)
Bottom Trace: Output; 500mV/Div., 1ms/Div. (400Hz)
FIGURE 27B.
FIGURE 28A.
FIGURE 27. WAVEFORMS OBSERVED WITH OTA ANALOG
MULTIPLIER USED AS A SUPPRESSED CARRIER
GENERATOR
The accuracy and stability of these multipliers are a direct
function of the power supply-voltage stability because the Yinput is referred to the negative supply-voltage. Tracking of
the positive and negative supply is also important because
the balance adjustments for both the offset voltage and output current are also referenced to these supplies.
0V
Linear Multiplexer - Decoder
A simple, but effective system for multiplexing and decoding
can be assembled with the CA3080 shown in Figure 29.
Only two channels are shown in this schematic, but the
number of channels may be extended as desired. Figure 30
shows oscilloscope photos taken during operation of the
multiplexer and decoder. A CA3080 is used as a 10µs delay“one-shot” multivibrator in the decoder to insure that the
sample-and-hold circuit can sample only after the input signal has settled. Thus, the trailing edge of the “one-shot” output-signal is used to sample the input at the sample-andhold circuit for approximately 1µs. Figure 31 shows
oscilloscope photos of various waveforms observed during
operation of the multiplexer/decoder circuit. Either the Q or Q
output from the flip-flop may be used to trigger the 10µs
“one-shot” to decode a signal.
10
0V
Top Trace: Input to X And Y; 2V/Div., 1ms/Div. (200Hz)
Bottom Trace: Output; 500mV/Div., 1ms/Div. (400Hz)
FIGURE 28B.
FIGURE 28. WAVEFORMS OBSERVED WITH OTA ANALOG
MULTIPLIER USED IN SIGNAL-SQUARING
CIRCUITS
Application Note 6668
MULTIPLEXER
3
CHANNEL #1
INPUT
10kΩ
2
+5V
DECODER
7
+
OTA
CA3080A
-
6
TRANSMISSION
MEDIA
7
3
OTA
CA3080
2kΩ
4
2
5
-5V
10kΩ
-
CL
Q
1/2 CD4013A
FLIP FLOP
2
D
Q
-5V
7 6
3kΩ
1
DECODED
OUTPUT
-5V
5
620Ω
+5V
150pF
4
2N4037
Q1
500kΩ
-5V
IABC
62kΩ
82kΩ
82kΩ
20pF
18pF
2
10kΩ
3
1N914
68kΩ
+5V
+5V
5
10kΩ
CHANNEL #2
INPUT
300pF
IABC
14
3
3N138
270Ω
0.01µF
5
82kΩ
0V
6
4
2kΩ
IABC
2.2kΩ
0.01µF
+
+5V
7
3
7
-
OTA
CA3080A
1N914
6
+
4
-5V
51pF
FROM
Q OR Q
2
1kΩ
10ms ONE - SHOT
5
OTA
CA3080A
+
6
4
10µs
-5V
FIGURE 29. TWO-CHANNEL MULTIPLEXER AND DECODER USING OTAs
Top Trace: Input Signal; 1V/Div., 20ms/Div.
Center Trace: Recovered Output; 1V/Div., 20ms/Div.
Bottom Trace: Multiplexed Signals; 2V/Div., 20ms/Div.
Top Trace: Input Signal; 1V/Div., 20ms/Div.
Center Trace: Recovered Output; 1V/Div., 20ms/Div.
Bottom Trace: Multiplexed Signals; 1V/Div., 20ms/Div.
FIGURE 30. WAVEFORMS SHOWING OPERATION OF LINEAR MULTIPLEXER/SAMPLE-AND-HOLD DECODE CIRCUITRY (FIGURE 29)
11
Application Note 6668
Top Trace: Flip-flop Output; 5V/Div., 20µs/Div.
Center Trace: “One-shot” Output; 5V/Div., 20µs/Div.
Bottom Trace: Strobe Pulse At The Collector of Q1;
0.1V/Div., 20µs/Div.
FIGURE 31A. WAVEFORMS CONTROLLING DECODER ENABLE
500µs/Div.
FIGURE 31C. SAME AS FIGURE 31B BUT WITH EXPANDED
TIME SCALE
FIGURE 31. VARIOUS WAVEFORMS SHOWING THE
OPERATION OF LINEAR MULTIPLEXER
High-Gain, High-Current Output Stages
Top Trace: Strobe Pulse at Q1; 0.5V/Div., 5ms/Div.
Center Trace: Multiplexed Output With One
Input at GND; 0.5V/Div., 5ms/Div.
Bottom Trace: Decoded Output; 0.5V/Div., 5ms/Div.
FIGURE 31B. WAVEFORMS SHOWING DECODER OPERATION
12
In the previously discussed examples, the OTA has been
buffered by a single insulated-gate field-effect-transistor
(MOSFET) shown in Figure 9. This configuration yields a
voltage gain equal to the (gM) (RO) product of the CA3080,
which is typically 142,000 (103dB). The output voltage and
current-swing of the operational amplifier formed by this
configuration (Figure 9) are limited by the 3N138 MOSFET
performance and its source-terminal load. In the positive
direction, the MOSFET may be driven into saturation; the
source-load resistance and the MOSFET characteristics
become the factors limiting the output-voltage swing in the
negative direction. The available negative-going load current
may be kept constant by the return of the source-terminal to
a constant-current transistor. Phase compensation is applied
at the interface of the CA3080 and the 3N138 MOSFET
shown in Figure 9.
Another variation of this generic form of amplifier utilizes the
CD4007A (CMOS) inverter as an amplifier driven by the
CA3080. Each of the three inverter/amplifiers in the
CD4007A has a typical voltage gain of 30dB. The gain of a
single CMOS inverter/amplifier coupled with the 100dB gain
of the CA3080 yields a total forward-gain of about 130dB.
Use of a two-stage CMOS amplifier configuration will
increase the total open-loop gain of the system to about
160dB (100,000,000). Figures 32 through 35 show examples
of these configurations. Each CMOS inverter/amplifier can
sink or source a current of 6mA (Typ). In Figures 34 and 35,
two CMOS inverter/amplifiers have been connected in parallel to provide additional output current.
Application Note 6668
+6V
+6V
14
5
INVERTING
INPUT
2
7
-
13
OUTPUT
CA3080
NON-INVERTING
INPUT
3
6
4
+
6
8
7
-6V
CD4007A. For greater output current capability, the remaining amplifiers in the CD4007A may be connected in parallel
with the single stage shown. Precise timing and thresholds
are assured by the stable characteristics of the input differential amplifier in the CA3080. Moreover, speed vs power
consumption trade-offs may be made by adjustment of the
IABC current to the CA3080. The quiescent power consumption of the circuits shown in Figure 37 is typically 6mW, but
can be made to operate in the micropower region by suitable
circuit modifications.
V+
-6V
V+
1/3 CD4007A
14
FIGURE 32. OTA DRIVING CMOS INVERTER/AMPLIFIER IN
OPEN-LOOP MODE
The open-loop slew-rate of the circuit in Figure 32 is
approximately 65V/µs. When compensated for the unity-gain
voltage-follower mode, the slew-rate is about 1V/µs (shown
in Figure 33). Even when the three inverter/amplifiers in the
CD4007A are connected as shown in Figure 34, the openloop slew-rate remains at 65V/µs. A slew-rate of about 1V/µs
is maintained with this circuit connected in the unity-gain
voltage-follower mode, as shown in Figure 35. Figure 36
contains oscilloscope photos of input-output waveforms
under small-signal and large-signal conditions for the circuits
of Figures 33 and 35. These photos illustrate the inherent
stability of the OTA and CMOS circuits operating in concert.
INVERTING
INPUT
2
3
NONINVERTING
INPUT
13
-
5
2kΩ
+6V
OTA
CA3080
+
6 6
VIN
2kΩ
2
OTA
CA3080
-
V-
7
50Ω
2.7Ω
0.1µF
+
8
0.25µF
OUTPUT
9
4
+6V
14
50µF
6
4
2kΩ
+6V
13
6
OUTPUT
FIGURE 34. OTA DRIVING TWO-STAGE CMOS
INVERTER/AMPLIFIER IN OPEN-LOOP MODE
7
+
N
8
V-
2
3
12
10
N 5
N
4
7
14
5
1
3
+6V
24kΩ
11
P
P
P
7
2
2kΩ
-
11
P
P
P
24kΩ
2
13
5
OTA
CA3080
3
+
0.1µF
4
1
3
6
6
N 5
N
12
10
N
8
OUTPUT
0.33Ω
7
INPUT
50µF
0.1µF
7
4
9
-6V
40µF
-6V
1/3 CD4007A
FIGURE 33. OTA DRIVING CMOS INVERTER/AMPLIFIER IN
UNITY-GAIN CLOSED-LOOP MODE
Precision Multistable Circuits
The micropower capabilities of the CA3080, when combined
with the characteristics of the CD4007A CMOS
inverter/amplifiers, are ideally suited for use in connection
with precision multistable circuits. In the circuits of Figures
32, 33, 34, and 35, for example, power-supply current drawn
by the CMOS inverter/amplifier approaches zero as the output voltage swings either positive or negative, while the
CA3080 current-drain remains constant.
Figure 37 shows a variety of circuits that can be assembled
using the CA3080 to drive one inverter/amplifier in the
13
-6V
-6V
FIGURE 35. OTA DRIVING TWO-STAGE CMOS
INVERTER/AMPLIFIER IN UNITY GAIN CLOSEDLOOP MODE
Application Note 6668
Top Trace: Input; 5V/Div., 100µs/Div.
Bottom Trace: Output; 5V/Div., 100µs/Div.
FIGURE 36A. LARGE SIGNAL RESPONSE FOR CIRCUIT IN
FIGURE 33
Top Trace: Input; 5V/Div., 100µs/Div.
Bottom Trace: Output; 5V/Div., 100µs/Div.
FIGURE 36C. LARGE SIGNAL RESPONSE FOR CIRCUIT IN
FIGURE 35
Top Trace: Input; 50mV/Div., 1µs/Div.
Bottom Trace: Output; 50mV/Div., 1µs/Div.
FIGURE 36B. SMALL SIGNAL RESPONSE FOR CIRCUIT IN
FIGURE 33
Top Trace: Input; 50mV/Div., 1µs/Div.
Bottom Trace: Output; 50mV/Div., 1µs/Div.
FIGURE 36D. SMALL SIGNAL RESPONSE FOR CIRCUIT IN
FIGURE 35
FIGURE 36. PERFORMANCE OF OTA DRIVING CMOS INVERTER/AMPLIFIER
14
Application Note 6668
1
f ≈ ------------------------------------------- 2R 1

2RCln  ----------- + 1
R
 2

R
V+
5kΩ
V+
100kΩ
14
7
3
C
0.01µF
13
OTA
CA3080
2
V+
V-
5
+
6 6
-
8
1/3 CD4007A
7
V-
10kΩ
R1
V-
The schematic diagram of a micropower comparator is
shown in Figure 38. Quiescent power consumption of this
circuit is about 10µW (Typ). When the comparator is strobed
“ON”, the CA3080A becomes active and consumes 420µW.
Under these conditions, the circuit responds to a differential
input signal in about 8µs. By suitably biasing the CA3080A,
the circuit response time can be decreased to about 150ns,
but the power consumption rises to 21mW.
The differential amplifier input common-mode range for the
circuit of Figure 38 is -1V to +10.5V. Voltage gain of the
micropower comparator is typically 130dB. For example, a
5µV input signal will switch the output.
4
R2
Micropower Comparator
V+
10kΩ
V+
FIGURE 37A. ASTABLE MULTIVIBRATOR
2
R1
--------------------- (V+ - V-) + V+ - V D
R1 + R2
= RCln ------------------------------------------------------------------------V+
10MΩ
R
V+
R1
V+
100kΩ
100kΩ
R2
T
7
2
56pF
V-
13
OTA
CA3080
+
3
4
VD
1N914
6 6
FIGURE 37B. MONOSTABLE MULTIVIBRATOR
 R1 
Threshold = ± V S  ---------------------
 R 1 + R 2
V+
V+
100kΩ
5.1kΩ
1/3 CD4007A
2
+
-
14
7
OTA
CA3080
V-
OUTPUT
14
V+
P
13
9
VSTROBE
FIGURE 38. SCHEMATIC DIAGRAM OF MICROPOWER COMPARATOR USING THE CA3080A AND CMOS
CD4007A
Current Mirrors
7
V-
3
12
Appendix I
V-
R = R1||R2
N
8
8
1/3 CD4007A
5
7
6
V+
5
-
CD4007A
6
V+ = 12V
V- = -2V
10kΩ
14
5
V-
1.3MΩ
10kΩ
1000pF
OTA
CA3080
3
+
4
IABC
10µA
V+
C
10
7
-
11
13
6
6
8
7
R2
V-
10kΩ
The basic current-mirror, described in the beginning of this
note, in its rudimentary form, is a transistor with a second transistor connected as a diode. Figure A shows this basic configuration of the current-mirror. Q2 is a diode connected transistor.
Because this diode-connected transistor is not in saturation and
is “active”, the “diode” formed by this connection may be considered as a transistor with 100% feedback. Therefore, the
base current still controls the collector current as is the case in
normal transistor action, i.e., IC = β IB. If a current I1 is forced
into the diode-connected transistor, the base-to-emitter voltage
will rise until equilibrium is reached and the total current being
supplied is divided between the collector and base regions.
Thus, a base-to-emitter voltage is established in Q2 such that
Q2 “sinks” the applied current I1.
I1
I2
R1
10kΩ
Q2
Q1
FIGURE 37C. THRESHOLD DETECTOR
FIGURE 37D. MULTISTABLE CIRCUITS USING THE OTA AND
CMOS INVERTER/AMPLIFIERS
15
FIGURE 39A. DIODE - TRANSISTOR CURRENT SOURCE
Application Note 6668
If the base of a second transistor (Q1) is connected to the baseto-collector junction of Q2, shown in Figure 39A, Q1 will also be
able to “sink” a current approximately equal to that flowing in the
collector lead of the diode-connected transistor Q2. This
assumes that both transistors have identical characteristics, a
prerequisite established by the IC fabrication technique. The
difference in current between the input current (I1) and the collector current (I2) of transistor Q1 , is due to the fact that the
base-current for both transistors is supplied from I1. Figure 39B
shows this current division, using a “unit” of base current (1) to
each transistor base. This base current causes a collector current to flow in direct proportion to the β of each transistor. The
ratio of the “sinking” current I2 to the input current I1 is therefore:
Figure 39D shows a curve-tracer photograph of characteristics
for the circuit of Figure 39B. No consideration in this discussion is
given to the variation of the transistor (Q1) collector current as a
function of its collector-to-emitter voltage. The output resistance
characteristic of Q1 retains its similarity to that of a single transistor operating under similar conditions. An improvement in its output resistance characteristic can be made by the insertion of a
diode-connected transistor in series with the emitter of Q1.
I2
---- = β/ ( β +˙ 2 ) .
I1
Thus, as β increases, the output current (I2) approaches the
input current (I1). The curves in Figure 39C show this ratio as a
function of the transistor β. When the transistor β is equal to
100, for example, the difference between the two currents is
only two percent.
I1
I2
β+2
I2
β
----- = ------------I1
β+2
β
β
2
Q2
Scale: Horizontal = 2V/Div.
Vertical = 1mA/Div.
Steps = 1mA/STEP
Q1
1
1
FIGURE 39D. PHOTO SHOWING RESULTS OF FIGURE 39B
CURRENT TRANSFER RATIO I2/I1
FIGURE 39B. DIODE - TRANSISTOR CURRENT SOURCE.
ANALYSIS OF CURRENT FLOW
1.5
1.4
1.3
1.2
1.1
1.0
0.9
This diode-connected transistor (Q3 in Figure 39E) may be
considered as a current-sampling diode that senses the emitter-current of Q1 and adjusts the base current Q1 (via Q2) to
maintain a constant-current in I2. Because all controlling transistors are operated at relatively fixed voltages, the previously
discussed effects due to voltage coefficients do not exist. The
curve-tracer photograph of Figure 39F shows the improved
output resistance characteristics of the circuit of Figure 39E.
(Compare Figure 39D and 39F).
2
I2
β + 2β
---- = ----------------------------2
I1
β + 2β + 2
0.8
0.7
0.6
0.5
0.4
I2
β
---- = ------------I1
β+2
I1
I2
Q1
0.3
0.2
0.1
0
1
10
100
1000
Q2
Q3
TRANSISTOR BETA
FIGURE 39C. CURRENT TRANSFER RATIO I2/I1 vs
TRANSISTOR BETA
16
FIGURE 39E. DIODE - 2 TRANSISTOR CURRENT SOURCE
Application Note 6668
Conclusions
The Operational Transconductance Amplifier (OTA) is a
unique device with characteristics particularly suited to
applications in multiplexing, amplitude modulation, analog
multiplication, gain control, switching circuitry, multivibrators,
comparators, and a broad spectrum of micropower circuitry.
The CA3080 is ideal for use in conjunction with CMOS ICs
being operated in the linear mode.
Acknowledgments
The author is indebted to C. F. Wheatly for many helpful
discussions. Valued contributions in circuit evaluation were
made by A. J. Visioli Jr. and J. H. Klinger.
Scale: Horizontal = 2V/Div.
Vertical = 1mA/Div.
Steps = 1mA/STEP
FIGURE 39F. PHOTO SHOWING RESULTS OF FIGURE 39E
Figure 39G shows the current-division within the mirror
assuming a “unit” (1) of current in transistors (Q2 and Q3).
The resulting current transfer ratio
2
β + 2β
I 2 /I 1 = -----------------------------.
2
β + 2β + 2
Figure 39C shows this equation plotted as a function of beta.
It is significant that the current transfer ratio (I2/I1) is
improved by the β2 term, and reduces the significance of the
2β + 2 term in the denominator.
I1
β+2
------------β+1
β+2
β + ------------β+1
I2
β+2
β  -------------
 β + 1
Q1
β+2
β
2
Q2
β
Q3
1
1
β+2
β  -------------
2
I2
 β + 1
β + 2β
---- = ---------------------------- = ----------------------------2
I1
β+2
β +  ------------- β + 2β + 2
 β + 1
FIGURE 39G. CURRENT FLOW ANALYSIS OF FIGURE 39E
All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification.
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reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
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17
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