AN9873: HC5503PRC SLIC with NeWave NW1034 Quad Combo Device

HC5503PRC SLIC and the NeWave NW1034
Combo Device
TM
Application Note
September 2000
AN9873
Author: Chris Ludeman
Introduction
the subscriber line. The desired source and termination
impedance at V2W is Z0 as shown in the diagram. The
purpose of the feedback block f(Z0) is to measure the loop
current IL that flows due to the signal source VS , and operate
on it such that the 2*RP+2*RS turns into a source and
termination impedance of Z0.
The network requirements of many countries require that an
analog subscriber line circuit terminate the subscriber line
with an impedance for voiceband frequencies which is
complex, rather than 600Ω. This requires that the physical
resistance that is situated between the SLIC and the
subscriber line, comprised of protection and/or sensing
resistors, and the output resistance of the SLIC itself, be
adapted to present an impedance to the subscriber line that
varies with frequency. This is accomplished using feedback
around the SLIC circuitry itself and the purpose of this
application note is to show a means of accomplishing this
task for the HC5503PRC, low cost SLIC for long loops with
the minimum amount of added circuitry.
This section therefore will develop an expression for VA/VTX ,
which is equivalent to f(Z0).
(EQ. 1)
V TX = 2 × ( 2 × R S ) × I L
Where:
V TR
I L = ----------------------------------------------------------( Z0 – 2 × R P – 2 × R S )
(EQ. 2)
a matching substituting for IL in (Equation 1)
The solution will accomplish the following:
• 2-wire complex impedance matching
2 × ( 2 × R S ) × V TR
V TX = ----------------------------------------------------------( Z0 – 2 × R P – 2 × R S )
• Flat gain versus frequency in both transmit and receive
direction in the presence of a frequency dependent
(complex) load
(EQ. 3)
Set inside SLIC
(EQ. 4)
V TR = 2 × V A
• Flexibility to accommodate other values of protection
and feed resistors
therefore,
• User selectable transmit and receive gains.
Impedance Matching
Impedance matching of the HC5503PRC to the subscriber
load is important for optimization of 2-wire return loss, which
in turn cuts down on echoes in the end to end voice
communication path. It is also important for maintaining
voice signal levels on long loops. Consider the equivalent
circuit shown in Figure 1.
Z0
2*RP
FROM
COMBO
X2
2*RS
FEED
AMPS VA
VRX
F(Z0)
XMIT
+
V2W
VS
TO
COMBO
X2
VTX
(EQ. 6)
2 × ( RP + RS )
1 Z0
f ( Z0 ) = --- ----------------- – ------------------------------------2 4 × RS
4 × RS
(EQ. 7)
1st term: Z0/4*RS requires gain and has no phase inversion.
The VTX signal therefore needs to pass through 2 inversions.
The circuitry inside the dotted box is representative of the
SLIC feed and transmit amplifiers, that pass the voice signals
in the receive and transmit directions respectively. Without the
feedback block f(Z0), the signal on the subscriber loop, V2W,
would see a source or termination resistance of 2*RP+2*RS ,
as the feed amplifiers present a very low output impedance to
1-888-INTERSIL or 321-724-7143
VA
1 ( Z0 – 2 × R P – 2 × R S )
f ( Z0 ) = ----------- = --- × ----------------------------------------------------------2
4 × RS
V TX
Note also that the form of the solution for impedance
matching can be deduced from the terms in equation 7.
FIGURE 1. IMPEDANCE MATCHING BLOCK DIAGRAM
4-1
(EQ. 5)
Note: In equation 6 above it would seem logical to simplify
the numerator by trying to combine Z0 and the two
subsequent terms together. In practice however, the network
Z0 cannot easily have 2*RP and 2*RS subtracted from it
since the sum of these resistors is often larger than the value
of the series resistance of the complex Z0 network. Also, as
will be seen, there is a need to identify a separate term
(Z0/4*RS) for equalization in the transmit path without
adding more reactive components to the application circuit.
VTR
IL
ZL
2 × ( 2 × RS ) × 2 × VA
V TX = ----------------------------------------------------------( Z0 – 2 × R P – 2 × R S )
2nd term: This term will be ≤1.0 but needs to be operated on
by the 1/2 outside the parentheses, resulting in attenuation
with phase inversion. This requires an op amp stage.
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Copyright
© Intersil Corporation 2000
Application Note 9873
RP
RS
0.47µF
TF
ZL
-
VA
IL
VFRO
+
RX
T
V2W
SLIC
GSR
R
VS
RP
VFRO-
R
RF
RS
NEWAVE
NW1034
2R
R*4RS
RP+RS
INTERSIL
HC5503PRC
+
VFXI-
0.2µF
GSX
TX
VTX
0.47µF
4RS
Z0
FIGURE 2. IMPEDANCE MATCHING
Receive Gain G(4-2)
See Figure 3.
Around SLIC op amp:
– ( V TX – V RX )
( 0 – V RX )
V A = ------------------------------------- × 3R – -------------------------- × R
8R
2R
–1
V A = ------ × ( 3V TX – 7V RX )
8
(EQ. 9)
(EQ. 10)
but
V2w
V TX = – 400 × -----------ZL
(EQ. 11)
So substituting for VTX in (Equation 10):
1 ( 1200 × V2w + 7V RX × Z L )
V A = --- × -----------------------------------------------------------------------8
ZL
(EQ. 12)
To express VA in terms of V2w:
( Z L + 300 )
V TR = V2w × ---------------------------ZL
(EQ. 15)
 ( Z L + 300 ) 300 7
V2w ×  ---------------------------- – ---------- = --- × V RX
ZL
ZL  4

(EQ. 16)
(EQ. 8)
Note that the term 0 above results from the fact that this
signal is cancelled by the echo cancellation circuitry
connected from VFRO to the VFXI - input of the Combo
transmit op amp. (See later.) Also, to simplify the equations,
some specific values have been selected for RP (50) and RS
(100), such that the term (4 * RS)/(RP + RS) simplifies to 8/3,
–1
V A = ------ × ( 3V TX – 3V RX – 4V RX )
8
V2w ( Z L + 300 ) 1 ( 1200 × V2w + 7V RX × Z L )
------------ × ---------------------------- = --- × -----------------------------------------------------------------------8
2
ZL
ZL
(EQ. 13)
V2w/VRX = 7/4 and this would normally be preceded by an
attenuator to adjust it to a gain of 1.0dB or 0dB (See Figure
6). Note that G(4-2) is not a function of ZL , and therefore is
flat over frequency.
Transmit Gain G(2-4)
See Figure 4.
From (Equation 11):
2 × 2 × RS
V TX = --------------------------- × V2W
ZL
Note that the sign is now changed from -VE to +VE
compared to the expression for VTX derived in the G(4-2)
derivation. A VTX signal that is in fact a transhybrid signal,
i.e., is derived from a source on the VRX input to the circuit,
undergoes inversion through the SLIC. A VTX signal that
results from a source on the 2-wire loop does not undergo
inversion through the SLIC. This can be deduced from the
direction of IL , the signal current in the 2-wire loop in
Figure 4.
The output of the Combo transmit op amp GSX is as follows:
Z0 × V TX
GSX = – k × ------------------------4 × RS
and VA = VTR/2, so
V2w ( Z L + 300 )
V A = ------------ × ---------------------------2
ZL
(EQ. 14)
Equating expressions for VA ; (Equation 12) and (Equation 14).
4-2
(EQ. 17)
(EQ. 18)
Application Note 9873
RS
RP
ZL
+
RX
T
V2w
SLIC
0.47µF
TF
-
VA
IL
VRX
R
R
RF
RP
RS
INTERSIL
HC5503PRC
2R
R*4RS
RP+RS
0.2µF
TX
VTX
0.47µF
0
FIGURE 3. RECEIVE GAIN G(4-2)
Specific Implementation for China
Substituting for VTX
– k × Z0 × 4 × Rs
GSX = ------------------------------------------4 × Rs × Z L
(EQ. 19)
GSX = -k x Z0/ZL . If Z0 is made equal to ZL as it would be
for correct impedance matching, then the transmit gain or
G(2-4) can be altered by adjusting the factor k associated
with the input resistor 4 x RS .
The design criteria for a China specific solution are as
follows:
• Desired line circuit impedance is 200 + 680//0.1µF.
• Receive gain is -3.5dB.
• Transmit gain is 0dB.
Transhybrid or Echo Cancellation G(4-4)
• 0dBm across the 2W load is defined as 1mW into the
complex impedance at 1020Hz.
See Figure 5.
• RP = 50, RS = 100
Since it was established earlier, that the signals VRX and
VTX of the application circuit are of opposite phase, if they
are summed together in the correct magnitudes at the input
to the Combo transmit op amp, they will cancel at the output
GSX, and this is necessary for hybrid or echo cancellation.
Assuming that the circuit has been set up so that the SLIC
matches the load impedance and that both G(4-2) and G(24) are adjusted to be 1.0 and flat over frequency as derived
above, then the gain from VFRO to GSX is +1.0 or 0dB, as
the VFRO signal goes through 2 inversions to GSX.
In order to achieve echo cancellation therefore, VFRO must
be added to the VTX signal at the input to the Combo
transmit op amp such that the gain from VFRO to GSX is 0.
Since the gain before echo cancellation is -1.0, VFRO must
be summed in with a gain of -1.0, and this can be done by
using an input resistor equal to the feedback resistor Z0.The
general solution is shown in Figure 5 with an input resistor
ZL , to match the load impedance value.
This then gives the general line circuit solution using the
Intersil HC5503PRC SLIC and the NeWave NW1034 Quad
CODEC devices. This is shown in Figure 6.
4-3
Impedance Matching
There is a one to one relationship between the SLIC
impedance setting components and the impedance it should
present to the 2w loop. The network most responsible for this
matching, is the Z0 network in the feedback circuit of the
Combo transmit op amp. It is usual to scale this network up
by a factor so that the load on the CMOS op amp does not
lead to distortion and possible instability effects. In this
example the scaling factor will be chosen to be 100.
200 + 680//0.1µF becomes 20K + 68K1//1µF.
The other components tied to impedance matching are
around the SLIC op amp. These three components are all
factors or R and so R can be a common scaling factor for
these components. In this example we chose to make
R = 100K. 2R becomes therefore 200K and the expression:
R × ( 4R S )
--------------------------RP + RS
becomes 267kΩ.
(EQ. 20)
Application Note 9873
RS
RP
0.47µF
TF
ZL
V2W
VFRO
RX
T
IL
VS
R
RF
RP
RS
NEWAVE
NW1034
INTERSIL
HC5503PRC
+
VFXI-
GSX
0.2µF
TX
0.47µF
VTX
Z0
4RS/K
FIGURE 4. TRANSMIT GAIN G(2-4)
RP
RS
VFRO-
0.47µF
TF
RX
T
0.47µF
ZL
R
RF
RP
RS
ZL
NEWAVE
NW1034
INTERSIL
HC5503PRC
ECHO
+
VFXI-
GSX
TX
0
VTX
0.47µF
4RS/K
Z0
FIGURE 5. ECHO CANCELLATION G(4-4)
These component values however are not the final values
because they have to be altered to account for the difference
between 0dBm on the 2w load and the 0dBm reference level
of the Combo.
-1.4dB is a ratio of 0.85 or 1/1.175.
Transmit Half Path Gain
Since this component change also affects the loop gain of
the feedback circuit that accomplishes impedance matching,
this component change must be compensated for elsewhere
in the circuit. The most obvious component to adjust is the
200K resistor that was calculated in the previous section.
This must be adjusted in the opposite direction by the same
factor. So 200K becomes 169K.
0dBm (complex load) = 0dBm (600) + 1.4dB at 1020Hz
reference frequency.
So to ensure that a 0dBm signal on the 2w loop is
represented by the digital milliwatt on the PCM backplane or
DX output of the Combo, a -1.4dB correction needs to be
added to the transmit half path gain.
4-4
The most convenient place to make this adjustment is on the
input resistor to the combo transmit op amp. So that 40K
becomes 47k5.
Application Note 9873
Receive Half Path Gain
We already know from the prior analysis of the receive path
transfer function that G(4-2) is 7/4 or +4.86dB but the
desired G(4-2) is;
-3.5dB + 1.4dB to convert 0dBm (600) into 0dBm (complex
load) = -2.1dB.
The gain without adjustment is +4.86dB.
The required adjustment is therefore (2.1 - 4.86)dB which is a
ratio of 0.448 or approximately 9/20 which can be implemented
with a potential divider of 90k1 and 110K resistors or, since the
input to the SLIC op amp is very high in the non-inverting
configuration, a 182K and 221K resistor divider.
the following analysis of the DC levels in the circuit will show
whether signal clipping is likely to occur.
The DC voltage at the non-inverting terminal of the SLIC op
amp is determined by VFRO which is about 2.5VDC. This
voltage is;
2.5 × 182K
----------------------------- = 1.129V
403K
(EQ. 21)
which also appears at the inverting terminal.
The only other contributor to the DC circuit conditions is
GSX, which also sits at near +2.5V. The DC voltage at the
output lead of the SLIC op amp is;
Transhybrid or Echo Cancellation
– ( 2.5 – 1.129 ) × 100K
---------------------------------------------------------- = 0.81VDC
169K
Referring to Figure 6, the only component involved in the
echo cancellation function is the network shown as ZL , one
of the input resistors to the Combo transmit op amp.
To calculate the maximum positive signal swing on the input
or output of the SLIC op amp we have to add the voice signal
to the DC signal.
Having calculated the half path gains, the transhybrid gain
from VFRO to GSX is now known.
Assuming that the maximum signal out of the Combo without
distortion is 0dBm0, and knowing that the potential divider
drops this signal in the receive path by a factor of 0.45, the
voice signal swing at this input is;
G(4-4) = (-3.5 + 1.4 - 1.4 + 0)dB,
G(4-4) = -3.5dB.
Since the gain from VFRO to GSX is Z0/ZL , and it should be
flat over frequency as VFRO is, then ZL has to be the same
network as Z0. This then has to be scaled to give the -3.5dB
loss from VFRO to GSX, such that the transhybrid signal is
cancelled at the output GSX of the combo transmit op amp.
1.129 + 0.775 x 1.414 x 0.45 = 1.62VPEAK .
Assuming again that the maximum signal delivered by the
combo to the 2w loop without distortion is 0dBm (complex)
or 0dBm(600) + 1.4dB, then the maximum voice signal swing
on the SLIC op amp output is calculated as follows;
-3.5dB is a ratio of 0.668 or 2/3.
From before:
Since Z0 was originally scaled by a factor of 100, we need to
scale ZL by a factor of 150.
( Z L + 300 )
V A = 0.5 × V2w × ---------------------------ZL
So 200 + 680//0.1µF becomes 300K + 102k//0.67µF or the
closest values. In practice this is the last adjustment to be
made to the line circuit after all the other standard
component values have been chosen. The transhybrid gain
or G(4-4) should then be measured and the ZL network
adjusted for optimum cancellation. The values calculated
here represent a good starting point.
where ZL is complex.
Considerations for Single Supply Combos
The NeWave NW1034 is a single supply Combo and has DC
present on the VFRO , GSX and VFRI- terminals. The
HC5503PRC is also a single 5V supply device and has DC
present on the RX and TX terminals. It is very important to
ensure that a direct connection is not present between the
SLIC and the Combo, otherwise latch-up of the Combo may
occur during circuit operation.
The DC blocking capacitors in Figure 6 are designed to block
the DC from the SLIC and the Combo, but still allows a DC
signal to be present at the input and output of the SLIC op
amp. This was done to allow a reduction in component
count. Since there are likely to be other circuit designs that
have different component values in the application circuit,
4-5
(EQ. 22)
(EQ. 23)
This simplifies to:
( 600 + 300 )
V A = 0.5 × 0.775 × 1.414 × ------------------------------- + 1.4dB
600
(EQ. 24)
VA = 0.82 + 1.4dB,
VA = 0.82 x 1.17,
VA = 0.96VPEAK .
The peak signal swing on the output lead of the SLIC op
amp is therefore this voice signal added to the DC already
present.
VA = +/-0.96VPEAK - 1.81VDC,
VA = +0.15VPEAK/-1.62VPEAK .
These signal swings are well within capabilities of this op amp.
For reference, a circuit designed for -7.0dB receive gain
would only increase the negative signal swing on the output
lead by 0.2VPEAK to -1.84VPEAK , while the +IN lead would
sit even closer to ground.
Application Note 9873
RP
RS
4R
SLIC
0.47µF
TF
RX
T
-
VA
ZL
3R
+
VFRO
0.47µF
R
R
RF
RP
RS
2R
ZL
NEWAVE
NW1034
R*RS
INTERSIL
HC5503PRC
RP+RS
VFXI -
0.2µF
+
GSX
TX
0.47µF
4RS/K
Z0
FIGURE 6. GENERIC LINE CIRCUIT SOLUTION
50
100
0.47µF
TF
RX
T
ZL
SLIC
163K
+
-
150*Z0
100K
267K
RF
100
VFRO
0.47µF
R
50
200K
NEWAVE
NW1034
170K
INTERSIL
HC5503PRC
+
VFXI -
0.2µF
GSX
TX
0.47µF
47K
100*Z0
FIGURE 7. LINE CIRCUIT DESIGN FOR CHINA NATIONAL NETWORK
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4-6
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