HC55185 and the Texas Instruments TCM38C17 Quad Combo TM Application Note May 2001 AN9933 Author: Don LaFontaine Reference Design using the HC55185 and the Texas Instruments TCM38C17 Quad Combo equal to the desired terminating impedance ZL, minus the value of the protection resistors (RP). The formula to calculate the proper RS for matching the 2-wire impedance is shown in Equation 1. The purpose of this application note is to provide a reference design for the HC55185 and Texas Instruments TCM38C17 Quad Combo. R S = 133.3 • ( Z L – 2R P ) (EQ. 1) Equation 1 can be used to match the impedance of the SLIC and the protection resistors (ZTR) to any known line impedance (ZL). Figure 1 shows the calculations of RS to match a resistive and 2 complex loads. The network requirements of many countries require the analog subscriber line circuit (SLIC) to terminate the subscriber line with an impedance for voiceband frequencies which is complex, rather than resistive (e.g. 600Ω). The HC55185 accomplishes this impedance matching with a single network connected between the VTX pin and the -IN pin. EXAMPLE 1: Calculate RS to make ZTR = 600Ω in series with 2.16 µF. RP = 49Ω. The TCM38C17 Quad Combo has a programmable receive output amplifier to adjust the output gain into the SLIC. The output amplifier gain is programmed with two simple resistors. Transhybrid balance is achieved via the TCM38C17 GSX amplifier. 1 - – ( 2 ) ( 49 ) R S = 133.3 600 + --------------------------------- –6 jω 2.16X10 (EQ. 2) RS = 66.9kΩ in series with 16.2nF. Note: Some impedance models, with a series capacitor, will cause the op-amp feedback to behave as an open circuit DC. A resistor with a value of about 10 times the reactance of the RS capacitor (2.16µF/ 133.3 = 16.2nF) at the low frequency of interest (200Hz for example) can be placed in parallel with the capacitor in order to solve the problem (491kΩ for a 16.2nF capacitor). Discussed in this application note are the following: • 2-wire 600Ω impedance matching. • Receive gain (4-wire to 2-wire) and transmit gain (2wire to 4-wire) calculations. • Transhybrid balance calculations. • Reference design for 600Ω 2-wire load. • Reference design for China complex 2-wire load. EXAMPLE 2: Impedance Matching Calculate RS to make ZTR = 200 + 680//0.1µF RP = 49Ω. Impedance matching of the HC55185 to the subscriber load is important for optimization of 2 wire return loss, which in turn cuts down on echoes in the end to end voice communication path. Impedance matching of the HC55185 is accomplished by making the SLIC’s impedance (ZO, Figure 1) 680 ZT = 133.3 200 + --------------------------------------------------------- – ( 2 ) ( 49 ) –6 1 + jω680 ( 0.1 )X10 (EQ. 3) RS = 13.6kΩ in series with the parallel combination of 90.6kΩ and 750pF. RS RESISTIVE INTERSIL HC55185 RP 49Ω VRX RS + VTR - ZL EG COMPLEX RS = 133.3(600 - 2*49) TIP + V2W - RS ZL = ZTR = 600Ω 66.9kΩ Std value 66.5kΩ ZL = ZTR = 600Ω + 2.16µF RS RS = 133.3(600 - 2*49) + 2.16µF/133.3 COMPLEX ZL = ZTR = 200Ω + 680//0.1µF RS RING RP 49Ω ZTR 66.9kΩ RS = 133.3(200 - 2*49)+ 133.3(680) // 0.1µF/133.3 ZT VTX RS ZO 491kΩ CTX 16.2nF -IN 13.6kΩ 750pF 90.6kΩ CFB ZO = ZL - 2RP VFB FIGURE 1. IMPEDANCE MATCHING 1 1-888-INTERSIL or 321-724-7143 | Intersil and Design is a trademark of Intersil Americas Inc. | Copyright © Intersil Americas Inc. 2001 Application Note 9933 SLIC in the Active Mode Loop Equation at HC55185 feed amplifier and load Figure 2 shows a simplified AC transmission model of the HC55185 and the connection of the TCM38C17 to the SLIC. Circuit analysis of the HC55185 yields the following design equations: The Sense Amplifier is configured as a 4 input differential amplifier with a gain of 3/4. The voltage at the output of the sense amplifier (VSA) is calculated using superposition. VSA1 is the voltage resulting from V1, VSA2 is the voltage resulting from V2 and so on (reference Figure 2). 3 VSA 1 = – --- ( V 1 ) 4 (EQ. 4) IX R - V TR + I X R = 0 (EQ. 16) Substitute Equation 15 into Equation 16 R S ∆I M 60 VTR = 2V RX – ------------------------- 8K (EQ. 17) Substitute Equation 12 for RS and -V2w /ZL for ∆IM into Equation 17. Z O V 2W VTR = 2V RX + -------------------Z (EQ. 18) L Loop Equation at Tip/Ring interface (EQ. 19) V2W -IM 2RP + VTR = 0 Substitute Equation 18 into Equation 19 and combine terms. 3 VSA 2 = --- ( V 2 ) 4 (EQ. 5) 3 VSA 3 = – --- ( V 3 ) 4 (EQ. 6) 3 VSA 4 = --- ( V 4 ) 4 (EQ. 7) 3 3 VSA = [ ( V 2 – V 1 ) + ( V 4 – V 3 ) ] --- = [ ∆V + ∆V ] --4 4 (EQ. 8) IX = Internal current in the SLIC that is the difference between the input receive current and the feedback current. (EQ. 9) IM = The AC metallic current. Z L + ZO + 2R P V2W -------------------------------------- = – 2V RX ZL (EQ. 20) where: VRX = The input voltage at the VRX pin. VSA = An internal node voltage that is a function of the loop current and the output of the Sense Amplifier. Where ∆V is equal to IMRSENSE (RSENSE = 20Ω) 3 V SA = 2 ( ∆I M × 20 ) --- = ∆I M 30 4 RP = A protection resistor (typical 49.9Ω). The voltage at VTX is equal to: RS RS V TX = – VSA -------- = – -------- ∆I M 30 8K 8K (EQ. 10) VTR is defined in Figure 2, note polarity assigned to VTR: V TR = 2 ( VRX + V TX ) (EQ. 11) Setting VRX equal to zero, substituting EQ. 10 into EQ. 11 and defining Z O = -VTR/∆IM will enable the user to determine the require feedback to match the line impedance at V2W. 1 ZO = ------------------ R S 133.33 (EQ. 12) ZO is the source impedance of the device and is defined as ZO = ZL - 2Rp. ZL is the line impedance. R S is defined as: R S = 133.33 ( Z L – 2R P ) (EQ. 13) Node Equation at HC55185 VRX input V RX V TX - + ----------IX = ---------R R (EQ. 14) Substitute Equation 10 into Equation 14 R S ∆I M 30 V RX - – -----------------------IX = ---------- R8K R (EQ. 15) 2 RS = An external resistor/network for matching the line impedance. VTR = The tip to ring voltage at the output pins of the SLIC. V2W = The tip to ring voltage including the voltage across the protection resistors. ZL = The line impedance. ZO = The source impedance of the device. Receive Gain (VIN to V2W) 4-wire to 2-wire gain is equal to the V2W divided by the input voltage VIN , reference Figure 2. The gain through the TCM38C17 is programmed to be 1.0 using Equation 21. R1 + R2 G ( PCMI N – PWRO + ) = ----------------------------R1 4 R 2 + ------ 4 (EQ. 21) The input and output gain adjustments are discussed in detail in PCM CODEC / Filter Combo Family: Device Designin and Application Data [1]. The maximum output (Gain =1) can be obtained by maximizing R1 and minimizing R2 (Figure 2). This can be done by letting R1 = infinity and R2 = 0, as shown in Figure 3. Application Note 9933 IX TIP + R IM - V2 RSENSE IM IX R 1:1 - Z0 VRX PWRO+ + VRX - R1 GSR + Ra1 VTX E - G + RING RP VTX - RSENSE V3 V4 + - 20Ω - IM + + IX + - + ANLGIN- PCMOUT + ANLGIN+ 100nF - IM + PCMIN + PWRO- Ra2 VTR + VIN R2 R - + + V2W - I + M ZL - + 20Ω RP RECEIVE BLOCK IX V1 TEXAS INSTRUMENTS TCM38C17 INTERSIL HC55185 + - AREF GSX VGSX - R TA FEEDBACK AMPLIFIER Rf RS + 4R 3R -IN 4R 4R 4R 3R CFB 8k + VFB VSA = ∆IM30 SENSE AMPLIFIER FIGURE 2. HC55185 SIMPLIFIED AC TRANSMISSION CIRCUIT AND TCM38C17 The receive gain is calculated using Equation 20 and the relationship RS = 133.33(ZL-2RP). Equation 22 expresses the receive gain (VIN to V2W) in terms of network impedances, where VIN = VPCMIN = VPWRO+ = VRX. ZL V 2W - = -2 --------------------------------------G 4-2 = ----------V IN Z L + Z O + 2 RP V2W Z L – 2R P - -----------------------V TX = ----------2 ZL (EQ. 26) Combining Equations 25 and 26 results in Equation 27. V TX ZO Z L – 2R P (EQ. 27) - = – ----------------------------------------------G 2-4 = ---------= – ----------------------------------------------EG 2 ( Z L + 2R P + ZO ) 2 ( ZL + 2R P + Z O ) (EQ. 22) Notice that the phase of the 4-wire to 2-wire signal is 180o out of phase with the input signal. A more useful form of the equation is rewritten in terms of VTX /V 2W. A voltage divider equation is written to convert from EG to V2W as shown in Equation 28. Transmit Gain Across HC55185 (EG to VTX) ZO + 2 RP V 2W = --------------------------------------- E G Z L + Z O + 2 R P The 2-wire to 4-wire gain is equal to VTX/ EG with VRX = 0, reference Figure 2. Substituting ZL = ZO + 2RP and rearranging Equation 28 in terms of E G results in Equation 29. Loop Equation E G = 2V 2W (EQ. 23) (EQ. 28) (EQ. 29) – E G + ZL I M + 2R P IM – VTR = 0 From Equation 18 with VRX = 0 Z O V 2W V TR = -------------------ZL (EQ. 24) Substituting Equation 24 into Equation 23 and simplifying. Z L + 2R P + ZO E G = – V 2W -------------------------------------ZL (EQ. 25) Substituting Equation 12 into Equation 10 and defining ∆IM = -V2W/ZL results in Equation 26 for VTX. 3 Substituting Equation 29 into Equation 27 results in an equation for 2-wire to 4-wire gain that’s a function of the synthesized input impedance of the SLIC and the protection resistors. ZO V TX - = – ------------------------------------------G 2-4 = ----------V 2W ( Z L + 2R P + ZO ) (EQ. 30) Notice that the phase of the 2-wire to 4-wire signal is out-of-phase with the input signal and when the protection resistors are set to zero, the transmit gain is -6dB. Application Note 9933 TEXAS INSTRUMENTS TCM38C17 RP 49Ω CRX 0.47µ µF TIP VRX + VTR - ZL EG GSR RING RP 49Ω ZTR PWRO+ PCMIN + + Ra2 PWRO- INTERSIL HC55185 ZO + 0.47µ µF VTX ANLGINCTX RS ZO = ZL - 2 RP -IN VFB CFB 0.47µ µF + ANLGIN+ 100nF + V2W - VIN Ra1 GSX PCMOUT AREF VGSX Rf FIGURE 3. RECEIVE GAIN G(4-2), TRANSMIT GAIN (2-4) AND TRANSHYBRID BALANCE Transmit Gain Across the System (V2W to VPCMOUT) 2-wire to 4-wire gain is equal to the VPCMOUT voltage divided by the 2-wire voltage V2W, reference Figure 3. V PC MOUT G 2 – 4 = ---------------------------V 2W (EQ. 31) VPCMOUT is only a function of VTX and the feedback resistors Ra1 and R f Equation 32. This is because VIN is considered ground for this analysis, thereby effectively grounding the VPWRO+ output. Transhybrid balance is achieved by summing the PWRO+ signal with the output signal from the HC55185 when proper gain adjustments are made to match VPWRO+ and VTX magnitudes. For discussion purpose, the GSX amplifier is redrawn with the external resistors in Figure 4. Ra2 ANLGIN+ Ra1 ANLGIN- VPWRO+ VTX + VGSX GSX Rf Rf V PCMOUT = – VTX ---------R a1 (EQ. 32) An equation for the system transmit gain is achieved by substituting Equation 30 into Equation 32. Rf ZO VTX R f V PC MOUT (EQ. 33) - ---------- ---------- = ------------------------------------------- = – ----------G 2 – 4 = ---------------------------V 2W V2W R a1 ( Z L + 2R P + Z O ) R a1 To achieve aTransmit Gain of one (VPCMOUT/ V 2W), make Rf = (ZL+2RP+Z0) and R a1 = Z0. Actual values of Ra1 and Rf were multiplied by 100 to reduce loading effects on the GSX opamp. Transhybrid Balance G(4-4) Transhybrid balance is a measure of how well the input signal is canceled (that being received by the SLIC) from the transmit signal (that being transmitted from the SLIC to the CODEC). Without this function, voice communication would be difficult because of the echo. The signals at VPWRO+ and VTX (Figure 3) are opposite in phase. Transhybrid balance is achieved by summing two signals that are equal in magnitude and opposite in phase into the GSX amplifier. 4 FIGURE 4. TRANSHYBRID BALANCE CIRCUIT The gain through the GSX amplifier from VPWRO+ is set by resistors Ra2 and R f. The gain through the GSX amplifier from VTX is set by resistors Ra1 and Rf . Transhybrid balance is achieved by adjusting the magnitude from both VPWRO+ and V TX so their equal to each other. Reference Design of the HC55185 and the TCM38C17 With a 600Ω Load The design criteria is as follows: • 4-wire to 2-wire gain (VPCMIN to V2W) equal 0dB • 2-wire to 4-wire gain (V2W to VPCMOUT) equal 0dB • Two Wire Return Loss greater than -30dB (200Hz to 4kHz) • Rp = 49.9Ω Figure 5 gives the reference design using the Intersil HC55185 and the Texas Instruments TCM38C17 Quad Combo. Also shown in Figure 5 are the voltage levels at specific points in the circuit. Application Note 9933 Impedance Matching Actual values of Ra1 and Rf were multiplied by 100 to reduce loading effects on the GSX op-amp. The 2-wire impedance is matched to the line impedance Z0 using Equation 1, repeated here in Equation 34. R S = 133.3 • ( Z L – 2R P ) Closest standard value for Rf is 121.0kΩ Closest standard value for Ra1 is 49.9kΩ (EQ. 34) The TCM38C17 receive gain is programmed to 1.0 by maximizing R1 and minimizing R2 resistor values (Figure 2). For a line impedance of 600Ω, RS equals: R S = 133.3 • ( 600 – 98 ) = 66.9kΩ The gain from PWRO+/VRX through the SLIC at VTX is 0.418 (Eq. 10 in the Intersil HC55185 data sheet, repeated here in Equation 39). (EQ. 35 V TX ZO = – ------------------------------------------G 4-4 = ----------VRX ( ZL + 2R P + Z O ) (EQ. 39) To achieve transhybrid balance from the PWRO+ pin to PCMOUT set Ra2 = Ra1 / 0.418. The closest standard value for R S is 66.5kΩ. Transhybrid Balance (ZL = 600Ω) 49.9 kΩ R a2 = -------------------- = 119.37kΩ 0.418 The internal GSX amplifier of the TCM38C17 is used to perform the transhybrid balance function.Transhybrid balance is achieved by summing two signals that are equal in magnitude and opposite in phase into the GSX amplifier. From Equation 33, repeated here in Equation 36, aTransmit Gain (VPCMOUT/ V 2W) of one is achieved if we make Rf = (ZL + 2RP + Z0) and Ra1 = Z0. (EQ. 40) Closest standard value for Ra2 = 118kΩ. Specific Implementation for China The design criteria for a China specific solution are as follows: • • • • Desired line circuit impedance is 200 + 680//0.1µF Receive gain (V2W/VPCMIN) is -3.5dB Transmit gain (VPCMOUT/V2W) is 0dB 0dBm0 is defined as 1mW into the complex impedance at 1020Hz • Rp = 49.9Ω Rf ZO V TX R f VPCMOU T - --------- ---------- = ------------------------------------------- (EQ. 36) G 2 – 4 = ---------------------------- = – ----------V2W V 2W R a1 ( Z L + 2R P + ZO ) R a1 R f = ( Z L + 2R P + Z O ) = ( 600 + 98 + 502 ) ( 100 ) = 120kΩ (EQ. 37) (EQ. 38) R a1 = ( Z O ) = 502Ω ( 100 ) = 50.2kΩ G4-2 0dBm0(600Ω) 0.7745VRMS CRX 0.47µ µF TIP PWRO+ VRX + ZL VTR - GSR RING RP 49Ω ZTR Ra2 VIN PCMIN + PCM Bus PWRO+ INTERSIL HC55185 ZO + 0.7745VRMS 118kΩ 0.47µ µF ANLGIN- VTX ZO = ZL - 2RP Ra1 -IN VFB + ANLGIN+ CTX RS 66.5kΩ CFB 0.47µ µF 49.9kΩ 0dBm0(600Ω) -7.5769dBm0 (600Ω) 0.7745VRMS 0.3239VRMS 100nF EG TEXAS INSTRUMENTS TCM38C17 0.7745VRMS RP 49Ω + V2W - 0dBm0(600Ω) 0dBm0(600Ω) GSX PCMOUT AREF Rf = 121kΩ VGSX 0dBm0(600Ω) 0.7745VRMS G2-4 FIGURE 5. REFERENCE DESIGN OF THE HC55185 AND THE TCM38C17 WITH A 600Ω LOAD IMPEDANCE 5 Application Note 9933 Figure 6 gives the reference design using the Intersil HC55185 and theTexas Instruments TCM38C17 Quad Combo. Also shown in Figure 6 are the voltage levels at specific points in the circuit. These voltages will be used to adjust the gains of the network. Adjustment to Get -3.5dBm0 at the Load Referenced to 600Ω The voltage equivalent to 0dBm0 into 811Ω (0dBm0(811Ω)) is calculated using Equation 41 (811Ω is the impedance of complex China load at 1020Hz). 2 V 0d Bm ( 811Ω ) = 10 log ------------------------------ = 0.90055V RMS 811 ( 0.001 ) (EQ. 41) The gain referenced back to 0dBm0 (600Ω) is equal to: 0.90055VRMS = 1.309dB G AIN = 20 log -------------------------------------0.7745V RMS (EQ. 42) (EQ. 43) The voltage at the load (referenced to 600Ω) is given in Equation 44: 2 V – 2.19 dBm ( 600Ω ) = 10 log ------------------------------ = 0.60196VRMS (EQ. 44) 600 ( 0.001 ) Impedance Matching The 2-wire impedance is matched to the line impedance ZL using Equation 1, repeated here in Equation 45. (EQ. 45) R S = 133.3 • ( Z L – 2R P ) 680 R S = 133.3 • 200 + --------------------------------------------------------- – ( 2 ) ( 49.9 ) –6 1 + jω680 ( 0.1 )X10 (EQ. 46) 0.1 µF R S = 133.3 • ( 102Ω ) + 133.3 • 680 Ω || ---------------133.3 (EQ. 47) RS = 13.6kΩ in series with the parallel combination of 90.6kΩ and 750pF (closest standard values are: RS = 13.7kΩ, RP = 90.9kΩ and CP = 680pF). To achieve a 4-wire to 2-wire gain (V PCMIN to V2W) that is equivalent to 0dBm(600Ω) at the complex load, the gain through the TCM38C17 (VPCMIN to VPWRO+) must equal 2.19dBm (0.60196VRMS). The gain through the TCM38C17 will then equal -2.19dBm (0.60196VRMS) divided by the input voltage 0dBm (0.7745V RMS). This gain is equal to 0.777. The gain through the TCM38C17 (VPCMIN to VPWRO+) is given in Equation 21 and repeated here in Equation 48. The adjustment to get -3.5dBm0 at the load referenced to 600Ω is: Adjus tm ent = – 3.5dBm0 + 1.309dBm0 = – 2.19 dB For a line impedance of 200 + 680//0.1µF, RS equals: R1 + R2 G ( PCMIN – PWRO ) = -----------------------------R 4 R 2 + ------1- 4 (EQ. 48) Setting the gain equal to 0.777 we can now determine the value of the gain setting resistors R1 and R2. Selecting the value of R 1 to be 49.9kΩ, R2 is calculated to 5.27kΩ. (Note: the value of R1 + R 2 should be greater than 10kΩ but less than 100kΩ.) R1 + R2 0.777 = -----------------------------R 4 R 2 + ------1- 4 (EQ. 49) G4-2 -2.19dBm0(600Ω) -2.19dBm0(600Ω) 0.60196VRMS 0.60196VRMS RP 49Ω VTR - EG R1 49.9kΩ VTX CTX 0.47µ µF RING RP 49Ω ZTR PWRO+ VRX + ZL VPWRO+ CRX 0.47µ µF TIP + V2W - ZO Ra2 57.6kΩ 13.7kΩ INTERSIL HC55185 680pF 680pF TEXAS INSTRUMENTS TCM38C17 GSR R2 5.23kΩ PWRO- 90.9kΩ + VFB R a1 CFB 0.47µ µF 68.1kΩ 100nF -IN - GSX + -2.19dBm0 (600Ω) -9.3294dBm0(600Ω) 0.60196VRMS 0.26461 RMS PCMOUT AREF Rf 10kΩ 1nF PCMIN + ANLGIN+ ZO = ZL - 2 RP VIN + ANLGIN- 90.9kΩ 0dBm0(600Ω) 0.7745VRMS 137kΩ 40.2kΩ 470pF VPCMOUT -3.5dBm0(600Ω) 0.51769VRMS G2-4 FIGURE 6. REFERENCE DESIGN OF THE HC55185 AND THE TCM38C17 WITH CHINA COMPLEX LOAD IMPEDANCE 6 Application Note 9933 0.222 R 2 = R 1 --------------- = 49.9k Ω ( 0.105 ) = 5.27k Ω 2.108 (EQ. 50) The closest standard value for R 2 is 5.23kΩ. Transhybrid Balance (ZL= 200 + 680//0.1µF) The internal GSX amplifier of the TCM38C17 is used to perform the transhybrid balance function.Transhybrid balance is achieved by summing two signals that are equal in magnitude and opposite in phase into the GSX amplifier. From Equation 33, repeated here in Equation 51, aTransmit Gain (VPCMOUT/ V 2W) of one is achieved if we make Rf = (ZL+2RP+Z0) and Ra1 = Z0. Rf V TX R f ZO VPCMOU T - --------- ---------- = ------------------------------------------- (EQ. 51) G 2 – 4 = ---------------------------- = – ----------V2W V 2W R a1 ( Z L + 2R P + ZO ) R a1 Closest standard values for Ra1 are: RS = 10kΩ, RP = 68.1kΩ and CP = 1nF. The TCM38C17 receive gain is programmed to 1.0 by maximizing R1 and minimizing R2 resistor values (Figure 2). The gain from PWRO+/VRX through the SLIC at VTX is 0.442 @ 1kHz (ZL = 813, ZO = 719) (Eq. 10 in the Intersil HC55185 data sheet, repeated here in Equation 56). V TX ZO G 4-4 = ----------= – ------------------------------------------VRX ( ZL + 2R P + Z O ) (EQ. 56) To achieve transhybrid balance from the PWRO+ pin to PCMOUT set Ra2 = Ra1 / 0.442. 10kΩ + 68kΩ || 1 nF R a2 = ----------------------------------------------------- = ( 22.62kΩ + 153.8kΩ || 442pF ) 0.442 (EQ. 57) Note: 2R P + ZO = ZL R f = ( Z L + 2R P + Z O ) = ( 2Z L ) = 2 ( 200 + 680 || 0.1µF ) (EQ. 52) R f = ( 400 Ω + 1.36kΩ || 50nF )100 = 40kΩ + 136k Ω || 500pF (EQ. 53) R a1 = ( Z O ) = ( Z L – 2R P ) = ( 100.2 + 680 || 0.1µF ) Closest standard values for Ra2 are: RS = 22.6kΩ, RP = 154kΩ and CP = 470pF. Reference [1] Website www.ti.com/sc/docs/psheets/abstract/apps/slwa006.htm (EQ. 54) R a1 = ( 100.2Ω + 680Ω || 0.1µF )100 = 10kΩ + 68kΩ || 1 nF (EQ. 55) Actual values of Ra1 and Rf were multiplied by 100 to reduce loading effects on the GSX op-amp. Closest standard values for Rf are: R S = 40.2kΩ, RP = 137kΩ and CP = 470pF. All Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at website www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see web sit ewww.intersil.com Sales Office Headquarters EUROPE Intersil SA Mercure Center 100, Rue de la Fusee 1130 Brussels, Belgium TEL: (32) 2.724.2111 FAX: (32) 2.724.22.05 NORTH AMERICA Intersil Corporation 2401 Palm Bay Road Palm Bay, FL 32905 TEL: (321) 724-7000 FAX: (321) 724-7240 7 ASIA Intersil Ltd. 8F-2, 96, Sec. 1, Chien-kuo North, Taipei, Taiwan 104 Republic of China TEL: 886-2-2515-8508 FAX: 886-2-2515-8369

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