AN9933: HC55185 and the Texas Instruments TCM38C17 Quad Combo

HC55185 and the Texas Instruments
TCM38C17 Quad Combo
TM
Application Note
May 2001
AN9933
Author: Don LaFontaine
Reference Design using the HC55185 and
the Texas Instruments TCM38C17 Quad
Combo
equal to the desired terminating impedance ZL, minus the
value of the protection resistors (RP). The formula to
calculate the proper RS for matching the
2-wire impedance is shown in Equation 1.
The purpose of this application note is to provide a reference
design for the HC55185 and Texas Instruments TCM38C17
Quad Combo.
R S = 133.3 • ( Z L – 2R P )
(EQ. 1)
Equation 1 can be used to match the impedance of the SLIC
and the protection resistors (ZTR) to any known line
impedance (ZL). Figure 1 shows the calculations of RS to
match a resistive and 2 complex loads.
The network requirements of many countries require the analog
subscriber line circuit (SLIC) to terminate the subscriber line
with an impedance for voiceband frequencies which is complex,
rather than resistive (e.g. 600Ω). The HC55185 accomplishes
this impedance matching with a single network connected
between the VTX pin and the -IN pin.
EXAMPLE 1:
Calculate RS to make ZTR = 600Ω in series with 2.16 µF.
RP = 49Ω.
The TCM38C17 Quad Combo has a programmable receive
output amplifier to adjust the output gain into the SLIC. The
output amplifier gain is programmed with two simple
resistors. Transhybrid balance is achieved via the
TCM38C17 GSX amplifier.
1
- – ( 2 ) ( 49 )
R S = 133.3  600 + ---------------------------------

–6
jω 2.16X10
(EQ. 2)
RS = 66.9kΩ in series with 16.2nF.
Note: Some impedance models, with a series capacitor, will
cause the op-amp feedback to behave as an open circuit
DC. A resistor with a value of about 10 times the reactance
of the RS capacitor (2.16µF/ 133.3 = 16.2nF) at the low
frequency of interest (200Hz for example) can be placed in
parallel with the capacitor in order to solve the problem
(491kΩ for a 16.2nF capacitor).
Discussed in this application note are the following:
• 2-wire 600Ω impedance matching.
• Receive gain (4-wire to 2-wire) and transmit gain (2wire to 4-wire) calculations.
• Transhybrid balance calculations.
• Reference design for 600Ω 2-wire load.
• Reference design for China complex 2-wire load.
EXAMPLE 2:
Impedance Matching
Calculate RS to make ZTR = 200 + 680//0.1µF
RP = 49Ω.
Impedance matching of the HC55185 to the subscriber load
is important for optimization of 2 wire return loss, which in
turn cuts down on echoes in the end to end voice
communication path. Impedance matching of the HC55185 is
accomplished by making the SLIC’s impedance (ZO, Figure 1)
680
ZT = 133.3  200 + --------------------------------------------------------- – ( 2 ) ( 49 )


–6
1 + jω680 ( 0.1 )X10
(EQ. 3)
RS = 13.6kΩ in series with the parallel combination of
90.6kΩ and 750pF.
RS
RESISTIVE
INTERSIL
HC55185
RP
49Ω
VRX
RS
+
VTR
-
ZL
EG
COMPLEX
RS = 133.3(600 - 2*49)
TIP
+
V2W
-
RS
ZL = ZTR = 600Ω
66.9kΩ
Std value
66.5kΩ
ZL = ZTR = 600Ω + 2.16µF
RS
RS = 133.3(600 - 2*49) +
2.16µF/133.3
COMPLEX
ZL = ZTR = 200Ω + 680//0.1µF
RS
RING
RP
49Ω
ZTR
66.9kΩ
RS = 133.3(200 - 2*49)+
133.3(680) // 0.1µF/133.3
ZT
VTX
RS
ZO
491kΩ
CTX
16.2nF
-IN
13.6kΩ
750pF
90.6kΩ
CFB
ZO = ZL - 2RP
VFB
FIGURE 1. IMPEDANCE MATCHING
1
1-888-INTERSIL or 321-724-7143
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Copyright © Intersil Americas Inc. 2001
Application Note 9933
SLIC in the Active Mode
Loop Equation at HC55185 feed amplifier and load
Figure 2 shows a simplified AC transmission model of the
HC55185 and the connection of the TCM38C17 to the SLIC.
Circuit analysis of the HC55185 yields the following design
equations:
The Sense Amplifier is configured as a 4 input differential
amplifier with a gain of 3/4. The voltage at the output of the
sense amplifier (VSA) is calculated using superposition.
VSA1 is the voltage resulting from V1, VSA2 is the voltage
resulting from V2 and so on (reference Figure 2).
3
VSA 1 = – --- ( V 1 )
4
(EQ. 4)
IX R - V TR + I X R = 0
(EQ. 16)
Substitute Equation 15 into Equation 16
R S ∆I M 60
VTR = 2V RX –  -------------------------
8K
(EQ. 17)
Substitute Equation 12 for RS and -V2w /ZL for ∆IM into
Equation 17.
Z O V 2W
VTR = 2V RX + -------------------Z
(EQ. 18)
L
Loop Equation at Tip/Ring interface
(EQ. 19)
V2W -IM 2RP + VTR = 0
Substitute Equation 18 into Equation 19 and combine terms.
3
VSA 2 = --- ( V 2 )
4
(EQ. 5)
3
VSA 3 = – --- ( V 3 )
4
(EQ. 6)
3
VSA 4 = --- ( V 4 )
4
(EQ. 7)
3
3
VSA = [ ( V 2 – V 1 ) + ( V 4 – V 3 ) ] --- = [ ∆V + ∆V ] --4
4
(EQ. 8)
IX = Internal current in the SLIC that is the difference
between the input receive current and the feedback current.
(EQ. 9)
IM = The AC metallic current.
Z L + ZO + 2R P
V2W -------------------------------------- = – 2V RX
ZL
(EQ. 20)
where:
VRX = The input voltage at the VRX pin.
VSA = An internal node voltage that is a function of the loop
current and the output of the Sense Amplifier.
Where ∆V is equal to IMRSENSE (RSENSE = 20Ω)
3
V SA = 2 ( ∆I M × 20 ) --- = ∆I M 30
4
RP = A protection resistor (typical 49.9Ω).
The voltage at VTX is equal to:
RS
RS
V TX = – VSA  -------- = –  -------- ∆I M 30
8K
8K
(EQ. 10)
VTR is defined in Figure 2, note polarity assigned to VTR:
V TR = 2 ( VRX + V TX )
(EQ. 11)
Setting VRX equal to zero, substituting EQ. 10 into EQ. 11
and defining Z O = -VTR/∆IM will enable the user to determine
the require feedback to match the line impedance at V2W.
1
ZO = ------------------ R S
133.33
(EQ. 12)
ZO is the source impedance of the device and is defined as
ZO = ZL - 2Rp. ZL is the line impedance. R S is defined as:
R S = 133.33 ( Z L – 2R P )
(EQ. 13)
Node Equation at HC55185 VRX input
V RX V TX
- + ----------IX = ---------R
R
(EQ. 14)
Substitute Equation 10 into Equation 14
R S ∆I M 30
V RX
- – -----------------------IX = ----------
 R8K 
R
(EQ. 15)
2
RS = An external resistor/network for matching the line
impedance.
VTR = The tip to ring voltage at the output pins of the SLIC.
V2W = The tip to ring voltage including the voltage across
the protection resistors.
ZL = The line impedance.
ZO = The source impedance of the device.
Receive Gain (VIN to V2W)
4-wire to 2-wire gain is equal to the V2W divided by the input
voltage VIN , reference Figure 2. The gain through the
TCM38C17 is programmed to be 1.0 using Equation 21.
R1 + R2
G ( PCMI N – PWRO + ) = ----------------------------R1


4 R 2 + ------
4
(EQ. 21)
The input and output gain adjustments are discussed in
detail in PCM CODEC / Filter Combo Family: Device Designin and Application Data [1]. The maximum output (Gain =1)
can be obtained by maximizing R1 and minimizing R2
(Figure 2). This can be done by letting R1 = infinity and
R2 = 0, as shown in Figure 3.
Application Note 9933
IX
TIP
+
R
IM
-
V2
RSENSE
IM
IX
R
1:1
-
Z0
VRX
PWRO+
+
VRX
-
R1
GSR
+
Ra1
VTX
E
- G
+
RING
RP
VTX
-
RSENSE
V3
V4
+
-
20Ω
- IM +
+
IX
+
-
+
ANLGIN-
PCMOUT
+
ANLGIN+
100nF
- IM +
PCMIN
+
PWRO-
Ra2
VTR
+
VIN
R2
R
-
+
+
V2W
-
I
+ M
ZL
-
+
20Ω
RP
RECEIVE
BLOCK
IX
V1
TEXAS
INSTRUMENTS
TCM38C17
INTERSIL
HC55185
+
-
AREF
GSX
VGSX
-
R
TA
FEEDBACK AMPLIFIER
Rf
RS
+
4R
3R
-IN
4R
4R
4R
3R
CFB
8k
+
VFB
VSA = ∆IM30
SENSE
AMPLIFIER
FIGURE 2. HC55185 SIMPLIFIED AC TRANSMISSION CIRCUIT AND TCM38C17
The receive gain is calculated using Equation 20 and the
relationship RS = 133.33(ZL-2RP).
Equation 22 expresses the receive gain (VIN to V2W) in
terms of network impedances, where VIN = VPCMIN =
VPWRO+ = VRX.
ZL
V 2W
- = -2 --------------------------------------G 4-2 = ----------V IN
Z L + Z O + 2 RP
V2W Z L – 2R P
- -----------------------V TX = ----------2
ZL
(EQ. 26)
Combining Equations 25 and 26 results in Equation 27.
V TX
ZO
Z L – 2R P
(EQ. 27)
- = – ----------------------------------------------G 2-4 = ---------= – ----------------------------------------------EG
2 ( Z L + 2R P + ZO )
2 ( ZL + 2R P + Z O )
(EQ. 22)
Notice that the phase of the 4-wire to 2-wire signal is 180o
out of phase with the input signal.
A more useful form of the equation is rewritten in terms of
VTX /V 2W. A voltage divider equation is written to convert
from EG to V2W as shown in Equation 28.
Transmit Gain Across HC55185
(EG to VTX)
 ZO + 2 RP 
V 2W =  --------------------------------------- E G
 Z L + Z O + 2 R P
The 2-wire to 4-wire gain is equal to VTX/ EG with VRX = 0,
reference Figure 2.
Substituting ZL = ZO + 2RP and rearranging Equation 28 in
terms of E G results in Equation 29.
Loop Equation
E G = 2V 2W
(EQ. 23)
(EQ. 28)
(EQ. 29)
– E G + ZL I M + 2R P IM – VTR = 0
From Equation 18 with VRX = 0
Z O V 2W
V TR = -------------------ZL
(EQ. 24)
Substituting Equation 24 into Equation 23 and simplifying.
Z L + 2R P + ZO
E G = – V 2W -------------------------------------ZL
(EQ. 25)
Substituting Equation 12 into Equation 10 and defining
∆IM = -V2W/ZL results in Equation 26 for VTX.
3
Substituting Equation 29 into Equation 27 results in an
equation for 2-wire to 4-wire gain that’s a function of the
synthesized input impedance of the SLIC and the protection
resistors.
ZO
V TX
- = – ------------------------------------------G 2-4 = ----------V 2W
( Z L + 2R P + ZO )
(EQ. 30)
Notice that the phase of the 2-wire to 4-wire signal is
out-of-phase with the input signal and when the protection
resistors are set to zero, the transmit gain is -6dB.
Application Note 9933
TEXAS
INSTRUMENTS
TCM38C17
RP
49Ω
CRX
0.47µ
µF
TIP
VRX
+
VTR
-
ZL
EG
GSR
RING
RP
49Ω
ZTR
PWRO+
PCMIN
+
+
Ra2
PWRO-
INTERSIL
HC55185
ZO
+
0.47µ
µF
VTX
ANLGINCTX
RS
ZO = ZL - 2 RP
-IN
VFB
CFB
0.47µ
µF
+
ANLGIN+
100nF
+
V2W
-
VIN
Ra1
GSX
PCMOUT
AREF
VGSX
Rf
FIGURE 3. RECEIVE GAIN G(4-2), TRANSMIT GAIN (2-4) AND TRANSHYBRID BALANCE
Transmit Gain Across the System
(V2W to VPCMOUT)
2-wire to 4-wire gain is equal to the VPCMOUT voltage
divided by the 2-wire voltage V2W, reference Figure 3.
V PC MOUT
G 2 – 4 = ---------------------------V 2W
(EQ. 31)
VPCMOUT is only a function of VTX and the feedback
resistors Ra1 and R f Equation 32. This is because VIN is
considered ground for this analysis, thereby effectively
grounding the VPWRO+ output.
Transhybrid balance is achieved by summing the PWRO+
signal with the output signal from the HC55185 when proper
gain adjustments are made to match VPWRO+ and VTX
magnitudes.
For discussion purpose, the GSX amplifier is redrawn with
the external resistors in Figure 4.
Ra2
ANLGIN+
Ra1
ANLGIN-
VPWRO+
VTX
+
VGSX
GSX
Rf
Rf
V PCMOUT = – VTX ---------R a1
(EQ. 32)
An equation for the system transmit gain is achieved by
substituting Equation 30 into Equation 32.
Rf
ZO
VTX R f
V PC MOUT
(EQ. 33)
- ---------- ---------- = ------------------------------------------- = – ----------G 2 – 4 = ---------------------------V 2W
V2W R a1
( Z L + 2R P + Z O ) R a1
To achieve aTransmit Gain of one (VPCMOUT/ V 2W), make
Rf = (ZL+2RP+Z0) and R a1 = Z0. Actual values of Ra1 and Rf
were multiplied by 100 to reduce loading effects on the GSX
opamp.
Transhybrid Balance G(4-4)
Transhybrid balance is a measure of how well the input
signal is canceled (that being received by the SLIC) from the
transmit signal (that being transmitted from the SLIC to the
CODEC). Without this function, voice communication would
be difficult because of the echo.
The signals at VPWRO+ and VTX (Figure 3) are opposite in
phase. Transhybrid balance is achieved by summing two
signals that are equal in magnitude and opposite in phase
into the GSX amplifier.
4
FIGURE 4. TRANSHYBRID BALANCE CIRCUIT
The gain through the GSX amplifier from VPWRO+ is set by
resistors Ra2 and R f. The gain through the GSX amplifier
from VTX is set by resistors Ra1 and Rf .
Transhybrid balance is achieved by adjusting the magnitude
from both VPWRO+ and V TX so their equal to each other.
Reference Design of the HC55185 and the
TCM38C17 With a 600Ω Load
The design criteria is as follows:
• 4-wire to 2-wire gain (VPCMIN to V2W) equal 0dB
• 2-wire to 4-wire gain (V2W to VPCMOUT) equal 0dB
• Two Wire Return Loss greater than -30dB (200Hz to
4kHz)
• Rp = 49.9Ω
Figure 5 gives the reference design using the Intersil
HC55185 and the Texas Instruments TCM38C17 Quad
Combo. Also shown in Figure 5 are the voltage levels at
specific points in the circuit.
Application Note 9933
Impedance Matching
Actual values of Ra1 and Rf were multiplied by 100 to reduce
loading effects on the GSX op-amp.
The 2-wire impedance is matched to the line impedance Z0
using Equation 1, repeated here in Equation 34.
R S = 133.3 • ( Z L – 2R P )
Closest standard value for Rf is 121.0kΩ
Closest standard value for Ra1 is 49.9kΩ
(EQ. 34)
The TCM38C17 receive gain is programmed to 1.0 by
maximizing R1 and minimizing R2 resistor values (Figure 2).
For a line impedance of 600Ω, RS equals:
R S = 133.3 • ( 600 – 98 ) = 66.9kΩ
The gain from PWRO+/VRX through the SLIC at VTX is
0.418 (Eq. 10 in the Intersil HC55185 data sheet, repeated
here in Equation 39).
(EQ. 35
V TX
ZO
= – ------------------------------------------G 4-4 = ----------VRX
( ZL + 2R P + Z O )
(EQ. 39)
To achieve transhybrid balance from the PWRO+ pin to
PCMOUT set Ra2 = Ra1 / 0.418.
The closest standard value for R S is 66.5kΩ.
Transhybrid Balance (ZL = 600Ω)
49.9 kΩ
R a2 =  --------------------  = 119.37kΩ
 0.418 
The internal GSX amplifier of the TCM38C17 is used to
perform the transhybrid balance function.Transhybrid
balance is achieved by summing two signals that are equal
in magnitude and opposite in phase into the GSX amplifier.
From Equation 33, repeated here in Equation 36, aTransmit
Gain (VPCMOUT/ V 2W) of one is achieved if we make
Rf = (ZL + 2RP + Z0) and Ra1 = Z0.
(EQ. 40)
Closest standard value for Ra2 = 118kΩ.
Specific Implementation for China
The design criteria for a China specific solution are as
follows:
•
•
•
•
Desired line circuit impedance is 200 + 680//0.1µF
Receive gain (V2W/VPCMIN) is -3.5dB
Transmit gain (VPCMOUT/V2W) is 0dB
0dBm0 is defined as 1mW into the complex impedance
at 1020Hz
• Rp = 49.9Ω
Rf
ZO
V TX R f
VPCMOU T
- --------- ---------- = ------------------------------------------- (EQ. 36)
G 2 – 4 = ---------------------------- = – ----------V2W
V 2W R a1
( Z L + 2R P + ZO ) R a1
R f = ( Z L + 2R P + Z O ) = ( 600 + 98 + 502 ) ( 100 ) = 120kΩ
(EQ. 37)
(EQ. 38)
R a1 = ( Z O ) = 502Ω ( 100 ) = 50.2kΩ
G4-2
0dBm0(600Ω)
0.7745VRMS
CRX
0.47µ
µF
TIP
PWRO+
VRX
+
ZL
VTR
-
GSR
RING
RP
49Ω
ZTR
Ra2
VIN
PCMIN
+
PCM
Bus
PWRO+
INTERSIL
HC55185
ZO
+
0.7745VRMS
118kΩ
0.47µ
µF
ANLGIN-
VTX
ZO = ZL - 2RP
Ra1
-IN
VFB
+
ANLGIN+
CTX
RS
66.5kΩ
CFB
0.47µ
µF
49.9kΩ
0dBm0(600Ω)
-7.5769dBm0 (600Ω)
0.7745VRMS
0.3239VRMS
100nF
EG
TEXAS
INSTRUMENTS
TCM38C17
0.7745VRMS
RP
49Ω
+
V2W
-
0dBm0(600Ω)
0dBm0(600Ω)
GSX
PCMOUT
AREF
Rf = 121kΩ
VGSX
0dBm0(600Ω)
0.7745VRMS
G2-4
FIGURE 5. REFERENCE DESIGN OF THE HC55185 AND THE TCM38C17 WITH A 600Ω LOAD IMPEDANCE
5
Application Note 9933
Figure 6 gives the reference design using the Intersil
HC55185 and theTexas Instruments TCM38C17 Quad
Combo. Also shown in Figure 6 are the voltage levels at
specific points in the circuit. These voltages will be used to
adjust the gains of the network.
Adjustment to Get -3.5dBm0 at the Load
Referenced to 600Ω
The voltage equivalent to 0dBm0 into 811Ω (0dBm0(811Ω))
is calculated using Equation 41 (811Ω is the impedance of
complex China load at 1020Hz).
2
V
0d Bm ( 811Ω ) = 10 log ------------------------------ = 0.90055V RMS
811 ( 0.001 )
(EQ. 41)
The gain referenced back to 0dBm0 (600Ω) is equal to:
0.90055VRMS
= 1.309dB
G AIN = 20 log -------------------------------------0.7745V RMS
(EQ. 42)
(EQ. 43)
The voltage at the load (referenced to 600Ω) is given in
Equation 44:
2
V
– 2.19 dBm ( 600Ω ) = 10 log ------------------------------ = 0.60196VRMS (EQ. 44)
600 ( 0.001 )
Impedance Matching
The 2-wire impedance is matched to the line impedance ZL
using Equation 1, repeated here in Equation 45.
(EQ. 45)
R S = 133.3 • ( Z L – 2R P )
680
R S = 133.3 •  200 + --------------------------------------------------------- – ( 2 ) ( 49.9 )
–6
1 + jω680 ( 0.1 )X10
(EQ. 46)
0.1 µF
R S = 133.3 • ( 102Ω ) + 133.3 • 680 Ω || ---------------133.3
(EQ. 47)
RS = 13.6kΩ in series with the parallel combination of
90.6kΩ and 750pF (closest standard values are: RS =
13.7kΩ, RP = 90.9kΩ and CP = 680pF).
To achieve a 4-wire to 2-wire gain (V PCMIN to V2W) that is
equivalent to 0dBm(600Ω) at the complex load, the gain
through the TCM38C17 (VPCMIN to VPWRO+) must equal 2.19dBm (0.60196VRMS). The gain through the TCM38C17
will then equal -2.19dBm (0.60196VRMS) divided by the
input voltage 0dBm (0.7745V RMS). This gain is equal to
0.777.
The gain through the TCM38C17 (VPCMIN to VPWRO+) is
given in Equation 21 and repeated here in Equation 48.
The adjustment to get -3.5dBm0 at the load referenced to
600Ω is:
Adjus tm ent = – 3.5dBm0 + 1.309dBm0 = – 2.19 dB
For a line impedance of 200 + 680//0.1µF, RS equals:
R1 + R2
G ( PCMIN – PWRO ) = -----------------------------R
4  R 2 + ------1-

4
(EQ. 48)
Setting the gain equal to 0.777 we can now determine the
value of the gain setting resistors R1 and R2. Selecting the
value of R 1 to be 49.9kΩ, R2 is calculated to 5.27kΩ.
(Note: the value of R1 + R 2 should be greater than 10kΩ but
less than 100kΩ.)
R1 + R2
0.777 = -----------------------------R
4  R 2 + ------1-

4
(EQ. 49)
G4-2
-2.19dBm0(600Ω)
-2.19dBm0(600Ω)
0.60196VRMS
0.60196VRMS
RP
49Ω
VTR
-
EG
R1
49.9kΩ
VTX
CTX
0.47µ
µF
RING
RP
49Ω
ZTR
PWRO+
VRX
+
ZL
VPWRO+
CRX
0.47µ
µF
TIP
+
V2W
-
ZO
Ra2
57.6kΩ
13.7kΩ
INTERSIL
HC55185
680pF
680pF
TEXAS
INSTRUMENTS
TCM38C17
GSR
R2
5.23kΩ
PWRO-
90.9kΩ
+
VFB
R a1
CFB
0.47µ
µF
68.1kΩ
100nF
-IN
- GSX
+
-2.19dBm0 (600Ω)
-9.3294dBm0(600Ω)
0.60196VRMS
0.26461 RMS
PCMOUT
AREF
Rf
10kΩ
1nF
PCMIN
+
ANLGIN+
ZO = ZL - 2 RP
VIN
+
ANLGIN-
90.9kΩ
0dBm0(600Ω)
0.7745VRMS
137kΩ
40.2kΩ
470pF
VPCMOUT
-3.5dBm0(600Ω)
0.51769VRMS
G2-4
FIGURE 6. REFERENCE DESIGN OF THE HC55185 AND THE TCM38C17 WITH CHINA COMPLEX LOAD IMPEDANCE
6
Application Note 9933
0.222
R 2 = R 1  --------------- = 49.9k Ω ( 0.105 ) = 5.27k Ω
 2.108 
(EQ. 50)
The closest standard value for R 2 is 5.23kΩ.
Transhybrid Balance (ZL= 200 + 680//0.1µF)
The internal GSX amplifier of the TCM38C17 is used to
perform the transhybrid balance function.Transhybrid
balance is achieved by summing two signals that are equal
in magnitude and opposite in phase into the GSX amplifier.
From Equation 33, repeated here in Equation 51, aTransmit
Gain (VPCMOUT/ V 2W) of one is achieved if we make
Rf = (ZL+2RP+Z0) and Ra1 = Z0.
Rf
V TX R f
ZO
VPCMOU T
- --------- ---------- = ------------------------------------------- (EQ. 51)
G 2 – 4 = ---------------------------- = – ----------V2W
V 2W R a1
( Z L + 2R P + ZO ) R a1
Closest standard values for Ra1 are: RS = 10kΩ,
RP = 68.1kΩ and CP = 1nF.
The TCM38C17 receive gain is programmed to 1.0 by
maximizing R1 and minimizing R2 resistor values (Figure 2).
The gain from PWRO+/VRX through the SLIC at VTX is
0.442 @ 1kHz (ZL = 813, ZO = 719) (Eq. 10 in the Intersil
HC55185 data sheet, repeated here in Equation 56).
V TX
ZO
G 4-4 = ----------= – ------------------------------------------VRX
( ZL + 2R P + Z O )
(EQ. 56)
To achieve transhybrid balance from the PWRO+ pin to
PCMOUT set Ra2 = Ra1 / 0.442.
10kΩ + 68kΩ || 1 nF
R a2 =  -----------------------------------------------------  = ( 22.62kΩ + 153.8kΩ || 442pF )


0.442
(EQ. 57)
Note: 2R P + ZO = ZL
R f = ( Z L + 2R P + Z O ) = ( 2Z L ) = 2 ( 200 + 680 || 0.1µF ) (EQ. 52)
R f = ( 400 Ω + 1.36kΩ || 50nF )100 = 40kΩ + 136k Ω || 500pF
(EQ. 53)
R a1 = ( Z O ) = ( Z L – 2R P ) = ( 100.2 + 680 || 0.1µF )
Closest standard values for Ra2 are: RS = 22.6kΩ, RP =
154kΩ and CP = 470pF.
Reference
[1] Website
www.ti.com/sc/docs/psheets/abstract/apps/slwa006.htm
(EQ. 54)
R a1 = ( 100.2Ω + 680Ω || 0.1µF )100 = 10kΩ + 68kΩ || 1 nF
(EQ. 55)
Actual values of Ra1 and Rf were multiplied by 100 to reduce
loading effects on the GSX op-amp.
Closest standard values for Rf are: R S = 40.2kΩ,
RP = 137kΩ and CP = 470pF.
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