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Non-Intrusive Hall-Effect Current-Sensing Techniques
Provide Safe, Reliable Detection and Protection for Power Electronics
By Paul Emerald
Abstract
As systems extend and expand the exploitation of the
latest power semiconductors (IGBTs, MCTs, etc.) that
manifest the very relentless advance in power output
limits, a prerequisite (and parallel) demand for sensing
these escalating current levels is (increasingly) very
apparent. Hall-effect ICs provide ‘non-intrusive’ current
sensing techniques and safe, isolated detection of high
current levels without dissipating the sizable amounts of
wasted power (and the resultant heating) associated with
resistive current-sensing methods. Further, Hall-effect
current sensing provides electrical isolation of the currentcarrying conductor; hence, a safe environment for circuitry,
operators, etc.
The proliferating current-sensing applications for Halleffect sensor ICs continue; become even more diverse; plus
expand and grow as other designers endeavor to protect
systems, create more reliable ‘bulletproof’ equipment, and
reconcile any safety issues. The prime applications for costeffective Hall-effect sensor ICs for current sensing include:
• Current Imbalance
• Current Monitoring
• Operator/User Safety and Security
• Overcurrent Detection/System Protection
• System Diagnosis and Fault Detection
• Test and Measurement
Background and Introduction
The discovery of the Hall-effect originated back in 1879;
however, any meaningful application of this Edwin H.
Hall finding awaited semiconductor integration that first
occurred in the late 1960s. Subsequently, further advances
(particularly those of the 1990s) have evolved further, more
fully functional integration plus an expanding series of
application-specific Hall sensor IC types. Yet the relentless
progress of magnetic sensor electronics continues to proliferate an increasing demand for low-cost, reliable, and ‘non-
STP98-1-AN, Rev. 2
contact’ Hall-effect circuitry for sensing/detecting motion,
direction, position, and measuring/monitoring current.
Hall-effect sensor ICs (especially the ratiometric linear
types) are superb devices for ‘open-loop’ current-sensing
designs. However, there are limits to the operational range,
accuracy and precision, frequency response, etc. that may
be realized. Because many prospective users are ignorant
of and/or oblivious to either the benefits or shortcomings of
current-sensing techniques using Hall-effect ICs, this paper
endeavors to provide a comprehensive discussion of the
essential, basic techniques of ‘non-intrusive’ current sensing
with silicon Hall-effect devices (HEDs) now available.
Most Hall-effect current-sensing requirements do not
develop adequate magnetic fields without the use of a slotted toroid to concentrate (and focus) the induced flux field.
Low-to-modest currents (<≈ 15 amperes) require winding
sufficient turns on the slotted toroid (core) to induce usable
flux strength and develop a suitable signal voltage. A higher
current level (>15 to 20 amperes) induces field intensities
that allow passing the current-carrying conductor straight
through the center of the toroid (no turns necessary at these
higher currents).
Designs requiring a broad (or continuous) current range
mandate utilizing linear Hall-effect sensor ICs. However,
overcurrent protection and/or fault detection designs can
be accommodated by digital HEDs. Examples and particulars of the essentials of current-sensing techniques,
device parameters, temperature stability, and other relevant
concerns of Hall-effect current sensing are covered in this
treatise on HEDs for sensing AC and DC currents.
Rival, Competing Technologies
Although there are many current-sensing methods, only
three are commonplace in low-cost, volume applications.
The others are expensive laboratory systems, emerging
technologies (magnetoresistive is an example), or seldom
used. The commonly used techniques include: (1) resistive,
(2) Hall effect, and (3) current transformers.
Hall-effect sensor ICs (open- and closed-loop) represent the next
tier of commonplace solutions. Insertion loss (and related heating, etc.) are not an obstacle. However, frequency range, cost, DC
offset, and external power represent the potential disadvantages
of Hall-effect IC technology when compared to the resistivesensing methods.
Current transformers close out the last low-cost technology, and
(as the term transformer should imply) are only useful with alternating currents. Most low-cost current transformers are designed
for narrow frequency ranges, are more expensive than resistive or
Hall-effect, and cannot be used for DC currents. However, current transformers avoid insertion loss, offer electrical isolation,
do not require external power, and exhibit no offset voltage at the
zero (null) current level.
Because this treatise focuses upon Hall-effect ICs, understanding
the elements of linear, ratiometric HEDs is imperative to openloop current sensing.
‘classic’ transfer curve for a ratiometric linear is illustrated in figure 1. Note that, at each extreme of its range, the output saturates.
Most recent linear Hall ICs provide a ratiometric output voltage. The quiescent (i.e., null) voltage is (nominally) 50% of
the applied, stable supply. This quiescent output voltage signal
equates to no applied magnetic field and, for current sensing, is
equivalent to zero current flow. A south polarity field induces
a positive voltage transition (toward VCC), and a north polarity
results in a transition toward ground (0 V). Output saturation
voltages are (typically) 0.3 V (high/sourcing) and 0.2 V (low/
sinking) and are measured at ±1 mA. [Ed. Note. output voltages
are now in the multivolt range.]
VCC
Saturation
Output Voltage (V)
Resistive sensing is very widely used, low-cost, and easily
understood. However, the shortcomings are its insertion loss
(heating and wasted power) and lack of isolation. Also, the series
inductance of many power resistors constrains the frequency
range with low-cost components. Low inductance, high-power
resistors for high frequency are more expensive, but allow operation beyond 500 kHz. Further, signal amplification is (usually)
required with resistive current-sensing techniques (either a comparator or operational amplifier is needed).
Quiescent
Output Voltage
Saturation
Linear Hall-Effect Sensor ICs
As the term implies, linear Hall sensor ICs develop an output signal that is proportional to the applied magnetic field. Normally,
in any current-sensing application, this flux field is focused by
a ‘slotted’ toroid to develop an adequate field intensity, and this
magnetic field is induced by current flowing in a conductor. A
0
B–
0
B+
Magnetic Flux Density, B (G)
Figure 1. Linear Hall Sensor IC Transfer Curve
Table 1. Commonplace, Inexpensive Current-Sensing Techniques
Power Consumption
Widely Used Sensors
Insertion
Loss
External
Power
Circuit
Isolation
Frequency
Range
Size
Accuracy
Relative
Cost
Sense Resistor + Op-Amp
High
Low
Low
DC to 10 MHz
Medium
±3 to 5%
Low
Standard Open Loop Hall-Effect
Low
Low
High
DC to 50 kHz
Small
±5 to 10%
Medium
Hall-Effect Closed-Loop
Low
Medium
High
DC to 1 MHz
Medium to
Large
< ±1%
High
Allegro Open-Loop Hall-Effect
Current Sensor ICs
Low
Low
High
DC to 120 kHz
Small
±2 to 3%
Medium
Current Transformers
Medium (AC)
None
High
60 Hz to 1 MHz*
Medium to
Large
±3 to 5%
High
* Current transformers usually operate over a limited frequency range but can be designed for use from low to high frequencies.
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Each linear Hall-effect IC integrates a sensitive Hall element
(also called a ‘plate’), a low-noise (bipolar) amplifier, and sink/
source output stage. Any systems problems associated with lowlevel signals and noise are minimized by the monolithic integration of magnetic Hall element, amplifier, output, and allied signal
processing circuitry.
Existing very stable, linear HEDs exploit dynamic quadrature
offset cancellation circuitry and utilize electronic switching to
change the current path in the Hall element. Switching the current
paths, from 0° to 90°, at a high repetition rate offers a new answer
to the (intrinsic) DC offset that has long plagued linear sensor IC
operation and stability.
Sample-and-hold circuitry and a low-pass filter are exploited to
properly ‘recondition’ the internal dynamic signals of these innovative linear HEDs.
Linear Hall-effect ICs can detect small changes in flux intensity,
and are (generally) more useful than digital Hall ICs for current
sensing. Linear HEDs are often capacitively coupled to op amps,
or DC connected to comparators, to attain system design objectives. Also, microcontrollers (µCs) and microprocessors (µPs) are
being exploited to detect small signal changes from linear Hall
ICs, and are very suitable (with proper software) for sensing/measuring either AC or DC currents.
Inducing a Magnetic Field
As mentioned, Hall-effect current sensing usually necessitates
the use of a slotted toroid (made of ferrous materials). The toroid
both concentrates and focuses an induced magnetic field toward
the location of the Hall-effect element within the IC package.
Figure 2 typifies a classic example of ‘non-intrusive’ current
sensing exploiting a slotted toroid. The conductor current flows
through the turns wound upon the toroid, and the induced flux
field is concentrated on the sensor IC in the gap (or slot) in
the toroid. Usually, this gap is made to closely match the Hall
IC package thickness ( approx. 0.060” or 1.52 mm), and this
provides optimal magnetic coupling. The current flow (with this
‘tight’ magnetic coupling) induces a flux intensity per the formula
B (gauss) ≈ N (turns) × 6.9 gauss/ampere
[Ed. Note: 6.9 gauss/ampere is updated from the earlier 6 gauss/
ampere.]
Widening the slot (gap) reduces the flux coupling and can
increase the upper current limit, which is predicated upon the
Hall sensor IC sensitivity (more to follow). However, decoupling the induced field to extend the maximum current limit may
affect linearity, usable range, etc. This ‘loose’ coupling is under
evaluation, but not yet complete; hence, no new formulas for
magnetic flux and conductor current (and larger gaps) have been
documented.
‘Calibrated’ Ratiometric Linear HEDs
The two newest [Ed. Note: Article originally presented in 1997.]
linear Hall sensors, with dynamic DC offset cancellation, provide
a cornerstone for a discourse on linear ratiometric HEDs and current sensing. The A3515 plot (figure 3) and related data (table 2)
Figure 2. Current Sensing with Gapped Toroid
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Figure 3. Linear, Ratiometric Hall-Effect Device Characteristics (A3515 Output)
Table 2. Linear, Ratiometric Hall-Effect Device Characteristics
Measurement Data (A3515), Measured Over ±250 Gauss
Marker
VCC
(Volts)
VOQ
(Volts)
Sensitivity
(mV/G)
Non-Linearity
(%)
Symmetry
(%)
Circle
4.500
2.217
4.450
≤0.1
99.9
Filled Square
5.000
2.463
5.014
≤0.2
99.9
Triangle
5.500
2.710
5.704
≤0.1
99.7
Figure 4. Linear, Ratiometric Hall-Effect Device Characteristics (A3516 Output)
Table 3. Linear, Ratiometric Hall-Effect Device Characteristics
Measurement Data (A3516), Measured Over ±500 Gauss
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Marker
VCC
(Volts)
VOQ
(Volts)
Sensitivity
(mV/G)
Non-Linearity
(%)
Symmetry
(%)
Circle
4.500
2.232
2.149
≤0.1
99.9
Filled Square
5.000
2.475
2.481
≤0.1
99.6
Triangle
5.500
2.723
2.820
≤0.1
99.9
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record the vital characteristics of the most sensitive linear HED;
its counterpart, the A3516, properties are in figure 4 and table 3.
Presently, though seldom sold, ‘calibrated’ linear Hall-effect ICs
are superb circuits for setting up and measuring system magnetic
parameters, and represent an excellent entry to the performance,
characteristics, and limitations of ratiometric ICs.
Linear Current Range(s)
The practical current limit (maximum with ‘tight’ coupling) is
derived using the range and flux per turn in the prior formula per
the approximation:
• A3515: ≥ ±400 G ÷6.9 G/A ≈ ±58 A
• A3516: ≥ ±800 G ÷6.9 G/A ≈ ±116 A
Sensor Sensitivity
The elemental distinction between the A3515 and A3516 is magnetic sensitivity. The nominal data for the two specific sensor ICs
depicted in figures 3 and 4 is listed in tables 2 and 3. Sensitivity
is specified in millivolts per gauss (mV/G). Three voltages are
listed; however, most designs utilize fixed, low-cost 5 V regulator
ICs for stability. The nominal sensitivity (and usable range) of the
two linear HEDs is as follows (VCC = 5 V):
• A3515
▫ Sensitivity: 5.0 mV/G
▫ Range: ≥ ±400 G (≥ ±2.0 V)
• A3516
▫ Sensitivity: 2.5 mV/G
Per a prior mention, current values beyond ≈ 115 amperes mandate reducing the magnetic coupling, shunting higher current
levels (i.e., pass a portion of the total through the toroid), or other
methods that effectively ‘desensitize’ the circuitry. There are
many, growing and expanding applications for ‘non-intrusive’
current sensing, especially at high currents (>100 A). An ultralow value resistor (<1 mΩ) dissipates considerable power and
heating at these currents, and the ‘non-inductive’ resistors needed
raise costs. I2R losses cannot be avoided; a sense resistor of
500 mΩ and 200 A produces 20 watts. Obviously, this is a situation that a designer would prefer to avoid. However, low-cost
options are scarce (or non-existent). [Ed. Note: The Allegro™
ACS75x current sensor IC series is now available, which can
accommodate currents up to the ±200 A range.]
Linear, Ratiometric Hall-Effect ICs
▫ Range: ≥ ±800 G (≥ ±2.0 V)
Linearity and Symmetry
From these plots (figures 3 and 4) it is apparent that neither
linearity nor symmetry (the deviation in the slope from the quiescent (or null) voltage) is a vital design consequence as neither
surpasses 0.3% for the A3515. The plots record ±400 G for the
A3515, and ±800 G for the A3516, and output voltage swings of
≥ ±2.0 V for both types.
1
The latest linear HEDs incorporating the dynamic quadrature DC
offset cancellation are illustrated in figure 5. The Hall element is a
‘single-plate,’ and designated by its symbol (Χ). Sensor IC current
is switched from a 0° orientation (downward) to a 90° path (across
the Hall plate) at ≈ 170 kHz. This precludes most of the earlier
offset related factors (DC imbalances due to resistive gradients,
geometrical dissimilarities, piezoresistive effects, etc.). A low-pass
filter and a sample-and-hold circuit are employed to recondition
the signal fed to the linear, ratiometric Hall sensor IC output.
SUPPLY
+
DYNAMIC
–
OFFSET CANCELLATION –
+
LOW PASS
FILTER
X
DYNAMIC OFFSET
CANCELLATION
Vcc
–
LOW-PASS
FILTER
+
Vcc/2
3
OUTPUT
2
GROUND
Dwg. FH-016A
Figure 5. Linear Hall-Effect Sensor with Dynamic Quadrature Offset Cancellation
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Powering Linear Hall-Effect ICs
Although the power requirements for linear HEDs are small,
external power is needed. The source must be stable and well
regulated; and with fixed voltage IC regulators (usually 5 V) this
design issue is easily (and inexpensively) resolved. The linear
sensor ICs specify a maximum supply current of ≤10 mA with
5 V (typical value ≈ 7 mA). Easy, on-board, ‘down’ regulation
from a system supply is simple with low-cost IC regulators.
A listing of absolute maximum limits for the new linear, ratiometric sensor ICs follows in table 4.
Operation beyond the above specified limits may affect device
operation, performance, or result in compromising (sacrificing)
circuit and/or system reliability and is (absolutely) not recommended.
Maximum Supply Voltage The recent linear HEDs, with
offset cancellation, permit operation at a higher supply than the
prior generation (A3506, etc.). These new linear ICs boost the
maximum limit to that of table 4.
Maximum Output Voltage Also itemized in table 4; however,
it should be noted that the output must not be connected to a
voltage either beyond the supply or below the IC ground. Either
might compromise the Hall sensor IC reliability and/or affect
system dependability.
Maximum Output Current The newest linear HEDs specify
a higher current than prior devices. However, typical applications rarely involve more than a trivial percentage of the 10 mA
maximum listed in table 4. The high-impedance inputs of today’s
analog or conversion circuitry (usually) necessitates microamperes not milliamperes of Hall sensor IC output current.
Maximum Flux Density Magnetic fields that exceed the linear
range of these Hall-effect ICs neither damage nor destroy the
device. However, magnetic fields beyond the usable range force
the output into saturation (and non-linear operation) without harm
to the HED.
Package Power Dissipation The maximum package power
dissipation limit is based upon operating with safe, reliable junction temperatures. The two package types in use are specified
Table 4. Absolute Maximum Limits (TA= 25°)
Characteristic
Supply Voltage, VCC
Rating
8.0 V
Output Voltage, VOUT
8.0 V
Output Sink Current, IOUT
10 mA
Magnetic Flux Density, B
Unlimited
Package Power Dissipation, PD
600 mW*
* ‘UA’ package rating of 183°C/W. [Ed. Note: Rating is 184.]
STP98-1-AN, Rev. 2
below for their thermal resistance (and maximum power with
TA = +25°C).
• ‘U’ Package: RθJA = 183°C/W (PD = 683 mW)
[Ed. Note: RθJA rating of 184 is correct.]
•‘UA’ Package: RθJA = 206°C/W (PD = 606 mW)
[Ed. Note: RθJA rating of 165 is correct.]
The maximum recommended junction temperature is 150° [Ed.
Note: Now up to 165°C.] and the dissipation should equal zero at
this temperature. However, the newest linears permit infrequent
(i.e., transient) excursions up to 200°C (ambient temperature,
TA ≤ 170°C).
The internal power (PD) consists of two factors: (a) the HED
supply power (ICC × VCC) and (b) the IC output power (IOUT ×
VOUT(SAT) ). Normally, supply power (a) smothers output dissipation (b), and for 5 V operation typical power dissipation is
≤40 mW. With ≤40 mW, the device junction temperature might
rise ≈ 8°C above the ambient (TJ ≤ TA + [PD × RθJA] ).
Internal power is (usually) not a HED limitation, but designers
should comprehend the basic results of device power dissipation
and its relationship to elevating the sensor IC junction temperature. IC (and system) reliability is inversely correlated to the
temperature of all system components. Higher ambient and junction temperatures reduce the life expectancy and dependability of
any system.
Distinctive Linear HED Parameters
Various, numerous linear-HED characteristics are of concern in
current-sensing applications, and brief descriptions of these follow. Subsequently, many of these characteristics and parameters
will be embodied in a focus on accuracy, temperature effects,
linearity, symmetry, etc.
Voltage Output As mentioned, the ratiometric, linear Hall
sensor ICs provide an output voltage that is proportional to
the applied magnetic field induced by current as illustrated in
figure 2. The output is specified to sink and source ±1 mA at
guaranteed limits. Per figures 2, 3, and 4, the usable range is
≥ ±2.0 V with a 5 V supply. Also previously mentioned, the
quiescent output voltage is 1⁄2 the supply when no magnetic field
is present (or current induced). A stable, well-regulated supply is
very necessary for proper operation, otherwise the output voltage
will fluctuate and follow any variations in supply. [Ed. Note: For
latest performance characteristics, refer to the selection guide on
the Allegro website.]
Circuit Loading with Hall-Effect Sensor ICs The linear
HEDs present no load to the conductor being sensed. A ‘nodisconnect,’ ‘non-intrusive’ technique is based upon forming a
‘toroid’ around the conductor being sensed. Rather than pass the
wire through the toroid (figures 6A and 6B), a soft iron piece is
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formed around the conducting wire. This permits sensing currents
without the need for disconnecting any conductors in the power
system (‘no-disconnect’ formed toroid is shown in figure 6C).
Tolerance to Current Overloads As mentioned, a conductor
current that exceeds the range of the linear Hall IC forces the output into a non-linear, saturated condition. Excessive current does
not impair or damage the sensor IC. However, extreme, sustained
overcurrent could be a fire or safety hazard if the conductor overheats and creates a dangerous situation.
Response Time of Hall-Effect Current Sensors A review
of some of the current sensing devices utilizing Hall-effect-based
techniques and toroids reveals a rather broad range of sensor
IC response times. A majority of these (those including amplifiers) fall within a range of ≈ 7 ms to ≈ 15 ms, though others are
below and above these limits. Testing is (normally) specified with
di/dt = 100 A/ms; and the specified linear current ranges vary
from rather low (<5 A) to the extreme (>20,000 A). Obviously,
the 20 kA variety are expensive and do not exploit any low-cost
toroid techniques.
Hall-Effect Sensor IC Bandwidth Today, the usable bandwidth of most linear Hall ICs is ≥ 20 kHz. Signal voltage changes
little up to this frequency. However, noticeable phase shift
Figure 6B. Toroidal Current Sensing Application(>15 A)
Figure 6A. Toroidal Current Sensing Application(<15 A)
Figure 6C. ‘No-Disconnect’ Current Sensing Application
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becomes distinct at somewhat lower frequencies. Some variation
is apparent amongst different ICs and vendors, but the rolloff is
quite steep beyond ≈ 20 kHz. Although the cutoff frequency for
the −3 dB rolloff of all linear HEDs is inconsistent, 20 kHz to
25 kHz is a valid approximation.
Representative oscilloscope plots show the effects of frequency
on the Hall sensor IC signal. From DC to 500 Hz (figure 7) no
discernible phase shift materializes. The top signal is the HED
voltage, and the lower trace is the winding (coil) current.
The phase shift becomes quite noticeable with a 10 kHz input rate
(figure 8), and very apparent at 20 kHz (figure 9). Note: Testing performed with 20 turns on a gapped toroid; and the voltage
scales of the three plots are not identical. Other intermediatefrequency plots exhibit similar phase shifts, but were not included
due to space limits. [Ed. Note: Limitations refer to the strictures
of the original publication.]
Also, it should be mentioned that this bandwidth limitation is
correlated with the linear sensor IC. The magnetics (and induced
coupling) is definitely not a restricting factor to bandwidth within
this range of operating frequencies.
Obviously, with such bandwidth limitations, Hall sensor ICs cannot sense high-power PWM circuitry exploiting power MOSFETs
Figure 8. VOUT (upper) vs. IIN (lower) at 10 kHz
Figure 7. VOUT (upper) vs. IIN (lower) at 500 Hz
Figure 9. VOUT (upper) vs. IIN (lower) at 20 kHz
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or IGBTs at normal, inaudible operating frequencies (>20 kHz),
but a linear HED is viable for DC and ‘mains’ power.
Linear HED Response to Application of Power Increasingly, systems designers confront stringent power ‘budgets,’ and
seek techniques to conserve current and power. Battery-powered
and battery ‘backup’ designs are particular concerns, and any
method capable of curtailing power is scrutinized.
A recurring technique is to (periodically) activate the sensor IC
circuitry by switching the power supply on for brief intervals,
and then off for longer periods. Average power is related to duty
cycle. Thus, for low duty-cycle applications, the power consumed
can be decreased substantially. Fixed-voltage IC regulators (with
an ENABLE input) are one very viable circuit technique to
switch the HED supply and reduce average power.
Clearly, the time required for a linear Hall IC to provide a stable,
usable signal is very important, and two different linear HEDs
were evaluated to ascertain their power-up response characteristics. The devices exhibit dissimilar properties, and the oscilloscope plots portray their dynamic operation upon applying power
to the linears. These plots include a 5% window to compare the
settling of the signals as the voltage attains its final value.
Figure 11. A3506 Power-Up (2.0 µs/div.)
Figure 10. A3506 Power-Up (0.2 µs/div.)
Figure 12. A3515 Power-Up (5.0 µs/div.)
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The latest linear HEDs (with dynamic quadrature offset cancellation) have a slower response than an earlier generation that
exploits the orthogonal Hall element. The previous series (A3506,
etc.) settles to above 95% of final voltage in less than 1 ms (per
figure 10), and takes approximately 15 ms (per figure 11) to reach
its final value. The very obvious tradeoff: speed vs. accuracy and
resolution of the signal voltage at power-up.
Linear Hall Sensor Device/Toroid Hysteresis Tests executed at ±6 A, which induce a substantial output voltage signal
swing, reveal that any error involving hysteresis is rather minor
(≈ 1% for the combination of linear HED (A3516) and gapped
toroid). Inherently, linear Hall sensor ICs exhibit no hysteresis.
However, different slotted toroids (and varied magnetic materials) may possess differing hysteretic properties.
The actual measured voltage differentials ranged from ≈ 16 mV
to ≈ 22 mV with >2.1 V changes. Hysteresis is a minor concern
when using ferrite cores, but other ferrous cores (such as powdered iron) may exhibit different characteristics.
Thus, a complete, thorough evaluation of specific toroids and the
associated linear sensor IC would be a very prudent (and recommended) suggestion.
Core (Toroid) Saturation Normally, the saturation of a core
should not be an issue. A current-sensor application design that
employs sufficient turns to drive the output voltage of the HED
to nearly full scale (at the maximum design current) first induces
saturation of the sensor IC. For optimum accuracy, the number of
turns used should induce output voltage transitions that (just) fall
short of saturating the sensor IC (more on this).
Zero Crossover With a linear Hall-effect sensor IC, zero crossover corresponds to a zero magnetic field (no induced flux field
as B = 0 with 0 A). The HED output voltage with a zero magnetic
field equates to 1⁄2 supply (i.e., the quiescent output voltage).
Wide-Band Output Noise of Linear HEDs The wide-band
noise of these linear Hall ICs is inconsequential, and its value
linked to the device chosen. The testing specifications for the
recent, stable linear Hall IC series are:
•B=0
• BW = 10 Hz to 10 kHz
• IOUT ≤1 mA
Typical equivalent input noise voltage (Vn) values for the two
series of linears are:
• A3506, A3507, A3508: 125 mV
• A3515, A3516: 400 mV
Given that the lowest sensitivity of these HEDs is 2.5 mV/G,
plus that accurate measurement is not feasible at very low flux
strengths (more on this later), the consequences of wide-band
noise is (typically) a very minor consideration. Other factors
STP98-1-AN, Rev. 2
(particularly quiescent output voltage drift with temperature) are
much more significant.
The System Temperature A crucial constituent to consider,
the temperature range must be well understood, properly specified (without inordinate margins), and controlling this very vital
design element greatly aids the ability to realize reasonable
accuracy. Note: Open-loop designs cannot (easily) resolve small
variations in current. A core hysteresis of ≈ 1% precludes this
without contemplating the other (and more acute) effects of temperature upon a linear HED output parameters and their relationship to performance.
Quiescent Output Voltage (DC Offset) Essentially, the DC
offset of a ratiometric, linear Hall IC relates to its deviation from
the nominal quiescent output voltage (i.e., 1⁄2 supply). Lacking a
system calibration or individual ‘look-up’ table, this DC parameter very tangibly affects accuracy of any current-sensing system
utilizing linear Hall ICs. By referring back to both figures 3 and
4, and tables 2 and 3, the significance of DC offset (VOQ , or
quiescent output voltage) is very plain.
The latest ratiometric Hall-effect sensor ICs specify the DC quiescent output voltage limits as 1⁄2 supply ±0.2 V [Ed. Note: Refer
to Addendum.]. The quiescent output voltage drift over the HED
operating-temperature range corresponds to ±10 gauss with the
newest linear Hall ICs.
A significant facet of the static quiescent voltage is its tolerance limits. Present specifications list ±0.2 V [Ed. Note: Refer
to Addendum.] from the nominal, and this translates into a
±8% maximum error without any temperature-induced effects
(A3515/3516). Obviously, this latent error factor poses a formidable constraint, and must be given serious deliberation if accurate
voltages are prerequisite to system performance.
DC compensation for the quiescent output voltage is feasible by
regulating the supply to achieve the 2.5 V nominal, but this also
influences sensitivity and any interrelated offsets are likely to
prove intolerable in production. Per figures 3 and 4, boosting the
supply offsets a low quiescent output voltage, and reducing the
supply compensates for a high quiescent voltage. However, such
offsets adversely influence sensitivity and counteract the positive
aspects of ‘nulling’ the quiescent voltage.
Because the sensitivity specifications for the newest linears
encompass a ±10% tolerance without any temperature effects,
‘nulling’ the quiescent output voltage (to 2.5 V) to escape a ±8%
error in the quiescent output voltage seems rather absurd.
The DC drift of the earlier linears equated to ±20 gauss for the
‘premium’ type, and ranged to ±50 gauss for a ‘limited’ temperature unit. Also, the ranges of tolerances for quiescent output
voltage of prior ICs was broader (or very much broader) than the
newest ICs with offset cancellation.
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This impedes the capacity to design an accurate, precise linear
sensing system that operates over a broad temperature range.
Designs necessitating tight current-sensing tolerances must
confront and reconcile any concerns linked to quiescent output
voltage (value and drift), and these are discussed in greater detail
in the section Accuracy of Open-Loop Linear Hall Sensor ICs.
Applying the drift relationships mentioned above, the maximum
quiescent output voltage drift error can be closely approximated.
These calculations are based upon the (nominal) linear sensitivities:
• A3515: ±10 G × 5.0 mV/G ≈ ±50 mV
• A3516: ±10 G × 2.5 mV/G ≈ ±25 mV
• A3506: ±20 G × 2.5 mV/G ≈ ±50 mV
• A3507: ±35 G × 2.5 mV/G ≈ ±87 mV
• A3508: ±50 G × 2.5 mV/G ≈ ±125 mV
Essentially, the list establishes the A3516 as the favored linear
when the quiescent voltage drift is an important criteria, and
maximum sensitivity is not the primary consideration. In currentsensing applications this entails twice the number of turns (vs.
A3515) to attain the same voltage swing.
Over a full-scale voltage swing (≥ ±2.0 V) the maximum error
with the A3516 is ≤ ±1.3% but, consistently, quiescent voltage
drift is < ±3 G (≈ ±7.5 mV with the A3516). This error factor is
dependent upon temperature; hence, sufficient turns should be
employed to drive the output near full-scale. This minimizes the
overall effect of temperature-related quiescent output voltage
drift. Therefore, operation near full-range is absolutely advised as
the ΔVOQ error percentage is lower.
Temperature Influence upon Sensor IC Sensitivity The
nominal sensitivities (and ranges) of both of the new linears
was mentioned previously. However, the circuit tolerances were
unspecified. The ICs have different nominal sensitivities; however, the temperature-related maximum shifts are identical. Reiterating sensitivity and range, plus adding the tolerances, produces
the following Hall-effect IC parameters and device temperature
shifts:
• A3515: Sensitivity, 5.0 mV/G ±10%
▫ ΔSensitivity (ΔT) at TA= Max, –2.5% (min), +2.5% (typ),
+7.5% (max)
▫ ΔSensitivity(ΔT) at TA = Max, –2.5% (min), +2.5% (typ),
+7.5% (max)
▫ ΔSensitivity(ΔT) at TA = Min, –9.0% (min), –1.3% (typ),
+1.0% (max)
▫ Magnetic Range, ≥ ±800 G (≥ ±2.0 V)
• Temperature Ranges:
▫ TA(min), –40°C
▫ TA(max), 85°C or 125°C
Essentially, the attainable accuracy of open-loop linear HEDs
involves DC offset and sensitivity.
Accuracy of Open-Loop Linear Hall Sensor ICs In any
classic mystery, at this juncture the ‘plot’ thickens. Because
precise, exacting measurement demands are increasing, a concise
explanation of the interrelated elements associated with attaining ‘accuracy’ and dependability is next. Accuracy, repeatability,
cost, etc. are very interrelated.
Though parametric maximums can be defined, the cumulative
impact on accuracy is quite nebulous. Also, it is improbable that
all worst-case errors occur coincidentally. Increasingly, cost-sensitive designs are based upon typical specifications, and this may
precipitate a small (although tolerable) failure rate that cannot
(easily) be decreased.
Pinpointing the absolute accuracy of ‘open-loop’ current sensing
is beyond this treatise. However, reviewing the essential factors
supports analysis.
• Hysteresis, hys, ≈ ±1%
• Output Quiescent Voltage, VOQ, ±8% [Ed. Note: Refer to
Addendum.]
▫ A3515 or A3516: 2.5 V ±0.2 V
• Output Quiescent Voltage Drift, ΔVOQ, ±10 G
▫ A3515: ≤±50 mV
▫ A3516: ≤±25 mV
• Sensitivity at TA = Max, ±10%
▫ A3515: 5.0 mV/G
▫ A3516: 2.5 mV/G
• ΔSensitivity at
▫ TA = Max, –2.5% to +7.5%
▫ TA = Min, –9.0% to +1.0%
▫ ΔSensitivity (ΔT) at TA= Min, –9.0% (min), –1.3% (typ),
+1.0% (max)
• Positive/Negative Linearity, ≈ 99.7%
▫ Magnetic Range, ≥ ±400 G (≥ ±2.0 V)
• Symmetry, ≈ 99.7%
• A3516: Sensitivity, 2.5 mV/G ±10%
STP98-1-AN, Rev. 2
• Wide-Band Noise, en, 400 µV
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Clearly, some of these elements are very crucial to attaining
accurate current sensing, while others are rather inconsequential.
Fundamentally, errors correlated to hysteresis, linearity, symmetry, and wide-band noise become quite insignificant. The
factors linked to quiescent voltage and sensitivity are (absolutely)
essential to any implementation of an accurate and precise current
sensing design.
Errors linked to quiescent output voltage drift are range dependent and device related. The ±10 G (typically < ±5 G) shift correlates to a potential error of 50% with a 10 gauss applied magnetic
field. However, the ±10 G drift represents less than 1.5% with a
field strength >667 G. Thus, the quiescent voltage error factor is
‘non-linear’ and is (substantially) diminished with large outputvoltage swings of the A3516 linear HED.
The quiescent output-voltage tolerance is listed as a percentage (≤ ±8% [Ed. Note: refer to Addendum.]). This is predicated
upon a nominal ratiometric (1⁄2 supply = 2.5 V), and the specified
limits of ≤ ±0.2 V [Ed. Note: Refer to Addendum.]. Because the
majority of linear Hall sensor ICs are much closer to nominal
(≤ ±0.1 V), the ±8% tolerance represents a very ‘worst-case’
quiescent output-voltage scenario.
The sensitivity parameters also pose considerable error potential.
However, these listings equate to a worst-case analysis. Further,
the relationships between sensitivity and the effects of temperature are not (as yet) completely specified. Whether a consistent
correlation between devices near either limit of sensitivity and
temperature-induced shifts exists is not specified. The temperature-related effects might be nil, or miniscule (temperature
cancels any cumulative deviations), or cumulative (temperature
further exacerbates the tolerances).
Based upon the published parameters and limits, open-loop
current-sensing designs cannot readily expect to attain results
below ≈ ±10% to ±15%. However, after reviewing recent plots
based upon test data (A3515/16), the prospect for boosting the
measurement accuracy (absolutely) improves.
Two plots (figures 13 and 15) delineate VOQ vs. temperature. The
+25°C data register an A3515 minimum of 2.468 V; a maximum
of 2.512 V; the A3516 spans from a minimum of 2.464 V to a
maximum of 2.501 V. This is much tighter than specified. The
–3 sigma limits for the ICs are: 2.457 V (A3515), and 2.462 V
(A3516). The +3 sigma data limits are 2.520 V (A3515) and
2.509 V (A3516), and these voltages convert to well within the
published ±8% tolerances [Ed. Note: Refer to Addendum.] for the
quiescent output voltage of these linears.
Data for the A3515 provides table 5 and 6, and data for the
A3516 provides table 7 and 8.
STP98-1-AN, Rev. 2
Table 5. VOQ in volts with VCC = 5 V
Ambient
Temperature
−40°C
25°C
85°C
150°C
−3 σ
2.448
2.457
2.463
2.472
Min.
2.461
2.468
2.473
2.481
Mean
2.487
2.489
2.493
2.501
Max.
2.517
2.512
2.520
2.530
+3 σ
2.525
2.520
2.523
2.531
Table 6. VOQ in volts as a percentage drift from value at 25°C
Ambient
Temperature
−40°C
25°C
85°C
150°C
−3 σ
−4.04
0.00
−1.15
−1.54
Min.
−2.90
0.00
−0.60
−0.60
Mean
−0.59
0.00
0.74
2.38
Max.
2.60
0.00
2.40
5.50
+3 σ
2.86
0.00
2.63
6.31
85°C
150°C
Table 7. VOQ in volts with VCC = 5 V
Ambient
Temperature
−40°C
25°C
−3 σ
2.454
2.462
2.462
2.466
Min.
2.458
2.464
2.467
2.472
Mean
2.484
2.485
2.483
2.485
Max.
2.503
2.501
2.498
2.499
+3 σ
2.514
2.509
2.504
2.504
Table 8. VOQ in volts as a percentage drift from value at 25°C
Ambient
Temperature
−40°C
25°C
85°C
150°C
−3 σ
−3.97
0.00
−3.36
−5.13
Min.
−3.60
0.00
−1.60
−2.90
Mean
0.12
0.00
−0.14
0.56
Max.
3.20
0.00
3.08
5.70
+3 σ
4.22
0.00
3.60
6.25
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The data and plots of ΔVOQ vs. temperature also record better
performance than the specified limit of ±10% (earlier listed in
millivolts). Figures 14 and 16 show the VOQ drift is well within
range, and the drift is very small in any narrow temperature band
about +25°C. Clearly, temperature range affects the output voltage shift tolerances.
Because these plots and data entail characteristics that fall within
certain HED specifications, some earnest deliberation on the
achievable accuracy is absolutely advised (particularly if the
temperature range is limited). Fundamentally, the effects of temperature are the foremost consideration in any endeavor to attain
single-digit (<10%) precision without calibration and/or compensation methods.
Figure 13. VOQ vs. Temperature (A3515)
Figure 14. ΔVOQ vs. Temperature (A3515)
Figure 15. VOQ vs. Temperature (A3516)
Figure 16. ΔVOQ vs. Temperature (A3516)
STP98-1-AN, Rev. 2
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Effects of Sensitivity upon Accuracy − The plots and data for
sensitivity confirm that the new linear HEDs are within published
limits, and delineate another (albeit secondary) constituent in
the resolution of accuracy. The device sensitivity and its interrelated variation over temperature are conservative, albeit without
extreme test margins. Figures 17 through 20 depict the sensitivity
data.
Data for the A3515 provides table 9 and 10, and data for the
A3516 provides table 11 and 12.
Table 9. Sensitivity in mV/G
Ambient
Temperature
−40°C
25°C
85°C
150°C
−3 σ
4.408
4.683
4.795
4.842
Min.
4.454
4.793
4.930
4.927
Mean
4.761
4.988
5.109
5.121
Max.
5.181
5.316
5.392
5.359
+3 σ
5.113
5.293
5.423
5.400
Table 10. Sensitivity as a percentage drift from value at 25°C
Ambient
Temperature
−40°C
25°C
85°C
150°C
−3 σ
−7.6
0.0
−0.1
−0.7
Min.
−7.1
0.0
−0.9
−1.0
Mean
−4.7
0.0
2.3
2.5
Max.
−2.5
0.0
3.7
4.4
+3 σ
−1.9
0.0
4.6
5.8
25°C
85°C
150°C
Table 11. Sensitivity in mV/G
Ambient
Temperature
−40°C
−3 σ
2.174
2.313
2.393
2.410
Min.
2.263
2.401
2.465
2.476
Mean
2.340
2.457
2.530
2.528
Max.
2.586
2.700
2.758
2.728
+3 σ
2.506
2.600
2.667
2.646
Table 12. Sensitivity as a percentage drift from value at 25°C
Ambient
Temperature
STP98-1-AN, Rev. 2
−40°C
25°C
85°C
150°C
−3 σ
−7.1
0.0
1.1
−0.1
Min.
−6.8
0.0
2.0
0.9
Mean
−5.0
0.0
2.7
2.6
Max.
−4.0
0.0
3.7
4.3
+3 σ
−2.9
0.0
4.2
5.3
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Figure 17. Sensitivity vs. Temperature (A3515)
Figure 18. ΔSensitivity vs. Temperature (A3515)
Figure 19. Sensitivity vs. Temperature (A3516)
Figure 20. ΔSensitivity vs. Temperature (A3516)
STP98-1-AN, Rev. 2
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Clearly, neither data nor plots reflect the overall distribution of
the ratiometric linear Hall sensor ICs. This insight into accuracy
is intended to advise of a basic necessity to reconcile the attainable limits of precise current sensing with HEDs, but it does
not imply any definite constraint. Ultimately, the application of
innovative, thoughtful circuit-design techniques determines the
essential limits of open-loop Hall-effect current sensing.
Calibration and Compensation Current-sensing designs
endeavoring to realize an open-loop accuracy below ±10% should
consider alternatives. Implementing ‘hardware’ calibration and/or
compensation represents a costly, complex option, and (for most
designs) should be ignored.
Whereas it is very feasible to establish trip points by using a
comparator (or multiple comparators) calibrating, or compensating, for temperature and quiescent voltage to realize a full range
of linear operation is a formidable task. The comparators can
provide discrete current signals (overcurrent, normal operation,
etc.) with useful accuracy, but cannot (easily) distinguish small
current changes.
Increasingly, software is the solution to extending the accuracy of
HED current sensing. Typically, this involves microcontrollers,
µPs, or computers, and a software calibration/ compensation
scheme.
Because the linearity, symmetry, and ratiometry of linear HEDs
is ≈ 100%, these error factors can (largely) be disregarded. The
temperature range is a definite factor if the system requires a
wide operating range. However, a benign environment with a
narrow temperature span alleviates design difficulties. The use
of software (and a µC/µP) to exploit a look-up table necessitates
measuring and storing sufficient data points to implement an
acceptable (and individual) calibration technique for each current
sensor IC. This (usually) involves the following calibration/compensation steps:
tion (including equipment), and the associated costs and time of
increased accuracy.
Obviously, the data-storage demands non-volatile memory for the
parametric measurements, and an individual, initial calibration
program. A look-up table compensates for the variations in quiescent voltage, sensitivity, and temperature effects. The latent errors
associated with these constituents to system accuracy can be
minimized by a software calibration and compensation technique.
Although this may appear to be complicated and costly, the other
solutions are liable to be more complex and more expensive than
using a low-cost 8-bit µC.
Sorting of Hall-Effect Sensor ICs Although this approach
could tighten device output parameters; presently, only linears
with published datasheet limits are available for sale. Some
‘value-added’ sorting is provided by others, but this procedure
and service is neither common nor inexpensive. Despite this,
specific customers have elected to solve formidable design issues
by outside testing, sorting, and selecting linear HEDs to specific,
tightened device limits. Clearly, any improvement in availability
of presorted HED ICs is a definite advantage to current-sensing
designs, and the availability of ‘sorted’ HEDs may change.
Size and Form of Sensor Assembly Because various sizes
of toroids with slots expressly cut to fit a HED package are available (Eastern Components, Inc.), a typical size cannot be identified. Figure 21 illustrates one basic configuration that is provided
in six different current ranges (peak current ratings sensed are:
1 A, 3 A, 5 A, 8 A, 10 A, and 100 A). The length, height, and
width vary somewhat, and the largest version measures 0.950”
long, 1.025” high, and 0.500” wide; all versions are PCB
through-hole form.
• Measuring and storing VOQ (the null current),
• Measuring and storing (specific) current points,
• Computing sensitivity from VOQ and data, and
• Measuring/storing temperature drift (if needed).
Determining the current level involves employing the ‘look-up’
data to calculate the current value via using the stored VOQ and
sensitivity data.
• Measure VOUT and calculate current value, and
• Measure system temperature and compensate for its drift effects
(if a system requirement).
In essence, the ‘look-up’ table corresponds to the ‘calibrated’
linear HEDs already mentioned. This software/look-up table
method can easily achieve <±10% accuracy, and its ultimate limit
(perhaps ≈ ±1%) is probably constrained by factors linked to
software development, the requisite calibration and compensa-
STP98-1-AN, Rev. 2
Figure 21. Hall IC Current-Sensing Assembly
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Cost of a Current-Sensing ‘Sub-System’ Identifying the
costs associated with a linear Hall IC-based current sensor is
virtually as difficult as the various issues involved with system
accuracy. The costs of the indispensable components (linear HED
and slotted toroid) can readily be determined, and the prices
of the complete assembly depicted in figure 21 start at ≈ $8.00
(1000 quantity). [Ed. Note: Valuations in 1997 USD.]
Slotted ferrite cores (usually) cost <$1.00 (even in modest quantities), and the linear Hall-effect sensor costs range from <$2.50 to
<$3.25 (1k pieces). This price span reflects the various Hallsensor types and the different temperature ranges. Obviously, unit
costs diminish in higher volumes, and the combined sensor/toroid
cost could easily fall (well) below $3.00 for volume production.
A conversion from ferrite cores to powdered iron toroids with
a ‘cast’ gap can meaningfully reduce overall cost. Rather than
ferrites with an $0.80 to $0.85 cost, powdered-iron cores are
estimated to be ≈ $0.20 to $0.25 in similar quantities.
However, other factors such as engineering time, software programming, assembly labor, etc. vary (considerably) based upon
each individual design requirement. Clearly, every system temperature, resolution, and accuracy are prerequisites that affect the
system cost. The outlays of developing and implementing a highresolution, very precise design with a wide temperature range are
greatly different than sensing only excessive current. An overcur-
rent fault detection application may allow a very broad tolerance
(perhaps ±20%), and this would not warrant any of the software
‘look-up’, stringent device and temperature evaluation that a
precise, full temperature design mandates.
Therefore, only the essential components (and the assembly of
figure 21) can be identified. Costs associated with software creation, system design engineering, etc. are (well) outside the realm
of utilizing linear Hall ICs for current sensing.
Protecting High-Power Electronics
A classic example of current-sensing detection and protection
for high-power IGBTs is shown in figure 22. This diagram can
relate to a single-phase of an adjustable speed drive (ASD) for
an AC induction motor or other power circuitry that requires a
full-bridge or triple half-bridge drive (for example, a 3-phase PM
brushless DC motor). Such a configuration can detect excessive
current in the supply rail (upper current sensor). This can result
from shorting the power rail to ground, or a shorted output combined with a corrresponding IGBT that is activated. Any combination of either a shorted lower or upper output with an on output
in the opposite portion of the same ‘leg’ can result in an (unsafe)
overcurrent fault in the system.
Alternatively, the linear sensor IC in series with the winding (center sensor) provides detection from shorted loads, and also monitors the actual coil current. Current sensor ICs in both locations
should preclude fire and safety hazards (and protect any personnel); and high-speed ‘shut-down’ circuitry can prevent damage
to the power outputs (if the overcurrent results from an external
fault such as improper equipment servicing). Clearly, overall
circuit response speed (shutdown time) is critical to protecting the
system and providing safety.
Summary and Perspective
Figure 22. ‘Full-Bridge’ with Current Sensor ICs
STP98-1-AN, Rev. 2
The applications for linear Hall-effect sensor ICs in open-loop
current sensing continue to evolve and expand. Presently, the
devices available are far superior to any earlier linears, and
advancements in design, processing, packaging, testing, etc. are
incessant and relentless. As mentioned, present-day HEDs have
tolerances and temperature drifts that pose formidable challenges
to those intending to design, develop, and implement systems that
demand dependable, single-digit accuracy over a wide range of
system operating temperatures.
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Expect further progress in HED performance and temperature
stability, more functional integration, and other developments
that make linear HEDs more viable for higher resolution current
sensing.
Future linears may allow programming the sensor IC after HED
packaging. This would permit users to tune the gain (sensitivity),
calibrate the output quiescent voltage (VOQ), and compensate
for the issues of temperature variations. Clearly, this involves
an innovative, more complex technique in the circuit design and
testing. However, the opportunities for applying such Hall sensor
ICs expand exponentially.
Hall-effect sensor ICs have undergone revolutionary changes
since their integration in the late 1960s. With further advancements and improvements, the applications for new linear HEDs
are expected to expand and multiply to satisfy the many emerging
needs of future power electronics systems.
sor ICs was made. In April 1998, the new, and tightened, limits
for quiescent output voltage were changed from the original
2.5 V ±0.2 V to 2.5 V ±0.075 V. In addition to this upgrade in the
quiescent output voltage limits, the effective linear current range
can be extended by widening the toroid gap (i.e., slot) to ‘desensitize’ the magnetic coupling.
Per the section titled Quiescent Output Voltage (DC Offset),
originally, the specifications listed the ratiometric output as
(nominally) 2.5 V. The limits were 2.3 V (min) and 2.7 V (max)
with VCC = 5 V over the device operating temperature range. This
improvement affects the achievable accuracy of systems applying
the ratiometric, linear Hall-effect sensor ICs (refer to the section
that is titled Accuracy of Open- Loop Linear Hall Sensor ICs.
As mentioned, this paper shows the following output quiescent
voltage limits:
VOQ .................................. 2.48 V to 2.52 V (±8%)
The upgraded specification now shows this as:
Acknowledgement
The symbol for a linear Hall-effect current sensor [Ed. Note: As
used in figure 22. The Χ mark is standard for indicating magnetic
actuation.] was created by Raymond Dewey [formerly] of Allegro
MicroSystems. Presently, no standard or accepted schematic symbol exists for current sensors utilizing Hall-effect technology.
References
Course: P. Emerald, “Open-Loop Current Sensing for Power
Conversion and Motion System Applications” in Principles of
Current Sensing, PCIM Power Electronics Institute, Chapter six,
PowerSystems World ‘97; Baltimore, MD; plus various contributors of the chapters comprising this one-day professional
advancement course.
Workshop: P. Emerald and Joe Gilbert “Integrated Hall-Effect
Sensors for Motion Control and Positioning Applications,” PowerSystems World ‘95, Long Beach, CA.
VOQ .............................. 2.425 V to 2.575 V (±3%)
This tightened specification significantly enhances the ability to
realize more accurate measurements via utilizing these linear,
ratiometric Hall-effect sensor ICs. This means that single-digit
accuracy is a reality for some designs (especially those with limited temperature fluctuations).
Linear Current Range(s) − Per the original material on Linear
Current Range, with ‘tight’ magnetic coupling (»60 mil gap to
match the sensor package) the ranges are unchanged:
A3515: ≥±400 G ÷ 6.9 G/A » ±58 A
A3516: ≥±800 G ÷ 6.9 G/A » ±116 A
‘Desensitizing’ the magnetic coupling can readily be realized via
expanding (widening) the slot in the toroid. The first endeavor
to desensitize the magnetic coupling involved increasing the slot
to 3 mm (≈ twice the package body), and this reduced the flux
coupling and increased the upper current limit as follows:
A3516: ≥ ±800 G ÷ 3.85 G/A » ±210 A
Addendum
Since this paper was written (December 1997), and presented, the
A3515 and A3516 ratiometric, linear Hall-effect sensor ICs have
been superseded by the A132x series. Information on the newer
series is available on the Allegro website, at http://www.allegromicro.com/en/Products/Part_Numbers/1321/.
Also, after the original publication, a change to the specifications
for the A3515 and A3516 ratiometric, linear Hall-effect sen-
STP98-1-AN, Rev. 2
Testing revealed that placement of the sensor IC case had no
effect upon the magnetic coupling. Centering the ‘calibrated’ linear Hall-effect sensor IC case resulted in the same output signal
as positioning the case against either face of the slot. Because
many users endeavor to attain higher current ranges, another
evaluation ensued (after new ferrite toroids were obtained from
Eastern Components, Inc.).
The next extension of the current range limit was undertaken with
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toroids gapped at 6 mm (e.g., more than 4× the package thickness dimension). This (very) ‘desensitized’ magnetic coupling
increased the maximum current limit per the following calculation:
A3516: ≥ ±800 G ÷ 1.7 G/A » ±470 A
Further evaluations are intended as toroids gapped with differing
dimensions become available. This should offer a more complete,
albeit overlapping, set of current ranges with an upper limit (as
yet) unknown. Also, other toroid materials (powdered iron in
particular) are to be evaluated.
Summary
The tightened quiescent output voltage tolerance offers better accuracy for the ratiometric, linear HEDs, and widening the
toroid slot increases the maximum current limitation of these
devices.
This paper was presented at the International Appliance Technical
Conference, Ohio State University, May 6, 1998. Reprinted by
permission.
Portions not copyrighted by Ohio State University Copyright ©2013 Allegro MicroSystems, LLC
The information contained in this document does not constitute any representation, warranty, assurance, guaranty, or inducement by Allegro to the
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STP98-1-AN, Rev. 2
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