DATASHEET

ISL6228
®
Data Sheet
May 7, 2008
High-Performance Dual-Output Buck
Controller for Notebook Applications
Features
The ISL6228 IC is a dual channel synchronous-buck PWM
controller featuring Intersil's Robust Ripple Regulator (R3)
technology that delivers truly superior dynamic response to
input voltage and output load transients. Integrated
MOSFET drivers and bootstrap diodes result in fewer
components and smaller implementation area.
• Fast transient response
• High performance R3 technology
Intersil’s R3 technology combines the best features of fixedfrequency and hysteretic PWMs while eliminating many of
their shortcomings. R3 technology employs an innovative
modulator that synthesizes an AC ripple voltage signal VR,
analogous to the output inductor ripple current. The AC signal
VR enters a window comparator where the lower threshold is
the error amplifier output VCOMP, and the upper threshold is a
programmable voltage reference VW, resulting in generation
of the PWM signal. The voltage reference VW sets the
steady-state PWM frequency. Both edges of the PWM can be
modulated in response to input voltage transients and output
load transients, much faster than conventional fixedfrequency PWM controllers. Unlike a conventional hysteretic
converter, each channel of the ISL6228 has an error amplifier
that provides ±1% voltage regulation at the FB pin.
The ISL6228 has a 1.5ms digital soft-start and can be
started into a pre-biased output voltage. A resistor divider is
used to program the output voltage setpoint. The ISL6228
operates in continuous-conduction-mode (CCM) in heavy
load, and in diode-emulation-mode (DEM) in light load to
improve light-load efficiency. In CCM, the controller always
operates as a synchronous rectifier. In DEM, the low-side
MOSFET is permitted to stay off, blocking negative current
flow into the low-side MOSFET from the output inductor.
Pinout
OCSET2
UGATE2
VO2
EN2
FB2
26
PHASE2
PGOOD2
27
25 24
23
22
VIN2 2
20 PVCC2
VCC2 3
19 LGATE2
VIN1 5
FSET1 6
PGOOD1 7
14
BOOT1
OCSET1
13
UGATE1
FB1
VO1
11 12
1
PHASE1
10
EN1
9
• Output voltage range: +0.6V to +5V
• Diode emulation mode for increased light load efficiency
• Programmable PWM frequency: 200kHz to 600kHz
• Pre-biased output start-up capability
• Integrated MOSFET drivers and bootstrap diode
• Internal digital soft-start
• Power good monitor
• Fault protection
- Undervoltage protection
- Soft crowbar overvoltage protection
- Inductor DCR overcurrent protection
- Over-temperature protection
- Fault identification by PGOOD pull-down resistance
• Pb-free (RoHS compliant)
Applications
• General purpose switching buck regulators
• PCI express graphical processing unit
• Auxiliary power rail
• VRM
• Network adaptor
TEMP
(°C)
PACKAGE
(Pb-Free)
PKG.
DWG. #
6228HRTZ -10 to +100 28 Ld 4x4 TQFN L28.4x4A
ISL6228HRTZ-T* 6228HRTZ -10 to +100 28 Ld 4x4 TQFN L28.4x4A
Tape and Reel
ISL6228IRTZ
6228IRTZ -40 to +100 28 Ld 4x4 TQFN L28.4x4A
18 PGND2
ISL6228IRTZ-T* 6228IRTZ -40 to +100 28 Ld 4x4 TQFN L28.4x4A
Tape and Reel
17 PGND1
*Please refer to TB347 for details on reel specifications.
16 LGATE1
NOTE: These Intersil Pb-free plastic packaged products employ special Pbfree material sets; molding compounds/die attach materials and 100% matte
tin plate PLUS ANNEAL - e3 termination finish, which is RoHS compliant and
compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that meet
or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
15 PVCC1
8
• Wide input voltage range: +3.3V to +25V
ISL6228HRTZ
21 BOOT2
GND
• Individual power stage input rail for each channel
PART NUMBER
PART
(Note)
MARKING
FSET2 1
VCC1 4
• ±1% regulation accuracy: -40°C to +100°C
Ordering Information
ISL6228 (28 LD 4x4 TQFN)
28
FN9095.2
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2007, 2008. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
Block Diagram
VIN2
FB2
VCC1
VCC2
FB1
VIN1
VO1
VO2
PWM FREQUENCY
CONTROL
PWM FREQUENCY
CONTROL
−
+
VW
10µA
gm
−
−
PWM
−
−
−
EA
−
−
−
10µA
10µA
−
+150°
OT
60Ω
90Ω
30Ω
30Ω
PHASE1
SHOOT THROUGH
PROTECTION
+
OCP
−
DRIVER
UGATE1
DRIVER
DIGITAL
SOFTSTART
+
UVP
+
UVP
+
OCP
POR
−
SHOOT THROUGH
PROTECTION
BOOT1
+
OVP
+
OVP
PWM CONTROL
PHASE2
EN1
+
+
DIGITAL
SOFTSTART
+
LGATE1
DRIVER
90Ω
60Ω
+150°
OT
PACKAGE
BOTTOM
FN9095.2
May 7, 2008
PVCC2
PGND2
PGOOD2
OCSET2
GND
OCSET1
FIGURE 1. SCHEMATIC BLOCK DIAGRAM
PGOOD1
PGND1
PVCC1
ISL6228
DRIVER
VCOMP
−
−
VREF
VREF
EA
POR
CR
R
Q
S
100pF
100pF
BOOT2
VR
+
PWM CONTROL
CR
−
−
+
VCOMP
+
gm
+
gm
VR
EN2
LGATE2
+
−
PWM
+
UGATE2
+
VW
+
gm
−
2
R
Q
S
10µA
−
+
+
FSET1
−
FSET2
ISL6228
Typical Application
5V
RPGOOD2
RPVCC2
RVCC2
RVCC1
VCC2
VCC1
PVCC2
CPVCC2
RPGOOD1
RPVCC1
PVCC1
CVCC2
CVCC1
CPVCC1
PGOOD2
PGOOD1
PGOOD2
PGOOD1
VIN1
VIN2
VIN2
VIN1
3.3V TO 25V
3.3V TO 25V
CIN2
UGATE1
UGATE2
BOOT2
BOOT1
CBOOT2
LO2
VO2
CIN1
QHIGH_SIDE1
QHIGH_SIDE2
CBOOT1
PHASE2
LO1
VO1
PHASE1
0.6V TO 5V
CO2
0.6V TO 5V
CSEN2
ISL6228
QLOW_SIDE2
QLOW_SIDE1
ROCSET2
CFB2
PGND2
LGATE1
PGND1
OCSET2
OCSET1
VO2
RTOP2
VO1
RFB1
CFB1
FB1
FB2
FSET1
FSET2
RBOTTOM2
RTOP1
RO1
RO2
RFSET2
CO1
ROCSET1
LGATE2
RFB2
CSEN1
RBOTTOM1
CFSET1
CFSET2
RFSET1
EN1
EN2
GND
3
FN9095.2
May 7, 2008
ISL6228
Absolute Voltage Ratings
Thermal Information
VIN1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V
VCC, PGOOD1,2 to GND . . . . . . . . . . . . . . . . . . . . . -0.3V to +7.0V
PVCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7.0V
EN1,2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to GND, VCC +3.3V
VO1,2, FB1,2, FSET1,2 . . . . . . . . . . . . . . -0.3V to GND, VCC +0.3V
PHASE1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . (DC) -0.3V to +28V
(<100ns Pulse Width, 10µJ) . . . . . . . . . . . . . . . . . . . . . . . . . -5.0V
BOOT1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
BOOT1,2 to PHASE1,2 . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V
UGATE1,2 . . . . . . . . . . . . (DC) -0.3V to PHASE1,2, BOOT1,2 +0.3V
(<200ns Pulse Width, 20µJ) . . . . . . . . . . . . . . . . . . . . . . . . -4.0V
LGATE1,2 . . . . . . . . . . . . . . . . . . . (DC) -0.3V to GND, PVCC +0.3V
(<100ns Pulse Width, 4µJ) . . . . . . . . . . . . . . . . . . . . . . . . . . -2.0V
Thermal Resistance (Typical, Notes 1, 2) θJA (°C/W)
θJC (°C/W)
TQFN Package . . . . . . . . . . . . . . . . . .
40
3
Junction Temperature Range. . . . . . . . . . . . . . . . . .-55°C to +150°C
Operating Temperature Range . . . . . . . . . . . . . . . .-40°C to +100°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Ambient Temperature Range. . . . . . . . . . . . . . . . . .-10°C to +100°C
Supply Voltage (VIN to GND) . . . . . . . . . . . . . . . . . . . . 3.3V to 25V
VCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5V ±5%
PVCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5V ±5%
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
3. Limits established by characterization and are not production tested.
Electrical Specifications
These specifications apply for TA = -40°C to +100°C; All typical specifications TA = +25°C, VCC = 5V,
PVCC = 5V; Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified.
Temperature limits established by characterization and are not production tested.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNIT
-
16
-
µA
-
0.1
1.0
µA
VIN
VIN Input Bias Current
IVIN
EN = 5V, VIN = 15V
IVIN_SHDN EN = GND, VIN = 25V
VIN Shutdown Current
VCC and PVCC
VCC Input Bias Current in Single-Channel
IVCC_S
EN1 = 5V, FB1 = 0.65V, VIN1 = 3.3V to 25V,
EN2 = GND, FB2 = GND, VIN2 = GND
-
1
-
VCC Input Bias Current in Dual Channel
IVCC_D
EN1 = 5V, FB1 = 0.65V, VIN1 = 3.3V to 25V,
EN2 = 5V, FB2 = 0.65V, VIN2 = 3.3V to 25V
-
2
-
mA
mA
VCC Shutdown Current
IVCC_SHDN EN1 = GND, EN2 = GND, VCC = 5V
-
0.1
1.0
µA
PVCC Shutdown Current
IPVCC_SHDN EN1 = GND, EN2 = GND, PVCC = 5V
-
0.1
1.0
µA
4.33
4.45
4.55
V
VCC POR THRESHOLD
Rising VCC POR Threshold Voltage
VVCC_THR
Falling VCC POR Threshold Voltage
V
TA = -10°C to +100°C
VCC_THF
TA = -10°C to +100°C
4.35
4.45
4.55
V
4.08
4.20
4.30
V
4.10
4.20
4.30
V
REGULATION
Reference Voltage
VREF
Regulation Accuracy
Close loop
-
0.6
-
V
-1
-
+1
%
200
-
600
kHz
-12
-
+12
%
0.60
-
5
V
PWM
Frequency Range
fSW
fSW = 300kHz
Frequency-Set Accuracy
VO Range
VVO
VO Input Leakage
IVO
4
EN = 5V, VO = 0.60V
-
1
-
µA
EN = 5V, VO = 5V
-
7.0
-
µA
EN = 0V, VO = 5V
-
0.1
-
µA
FN9095.2
May 7, 2008
ISL6228
Electrical Specifications
These specifications apply for TA = -40°C to +100°C; All typical specifications TA = +25°C, VCC = 5V,
PVCC = 5V; Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified.
Temperature limits established by characterization and are not production tested. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ERROR AMPLIFIER
FB Input Bias Current
FB = 0.60V
-33
-
15
nA
FB = 0.60V TA = -10°C to +100°C
-35
-
15
nA
RPG_SS
PGOOD = 5mA Sink
70
95
125
Ω
IFB
POWER GOOD
PGOOD Pull-Down Impedance
PGOOD Leakage Current
RPG_SS
PGOOD = 5mA Sink TA = -10°C to +100°C
75
95
125
Ω
RPG_UV
PGOOD = 5mA Sink
70
95
125
Ω
RPG_UV
PGOOD = 5mA Sink TA = -10°C to +100°C
75
95
125
Ω
RPG_OV
PGOOD = 5mA Sink
45
63
85
Ω
RPG_OV
PGOOD = 5mA Sink TA = -10°C to +100°C
50
63
85
Ω
RPG_OC
PGOOD = 5mA Sink
22
32
45
Ω
RPG_OC
PGOOD = 5mA Sink TA = -10°C to +100°C
25
32
45
Ω
IPGOOD
PGOOD = 5V
-
0.1
1.0
µA
PGOOD Maximum Sink Current (Note 3)
PGOOD Soft-Start Delay
tSS
EN High to PGOOD High
-
5.0
-
mA
2.20
2.75
3.50
ms
1.5
Ω
GATE DRIVER
UGATE Pull-Up Resistance (Note 3)
RUGPU
200mA Source Current
-
1.0
UGATE Source Current (Note 3)
IUGSRC
UGATE - PHASE = 2.5V
-
2.0
-
A
UGATE Sink Resistance (Note 3)
RUGPD
250mA Sink Current
-
1.0
1.5
Ω
UGATE Sink Current (Note 3)
IUGSNK
UGATE - PHASE = 2.5V
-
2.0
-
A
LGATE Pull-Up Resistance (Note 3)
RLGPU
250mA Source Current
-
1.0
1.5
Ω
LGATE Source Current (Note 3)
ILGSRC
LGATE - PGND = 2.5V
-
2.0
-
A
LGATE Sink Resistance (Note 3)
RLGPD
250mA Sink Current
-
0.5
0.9
Ω
LGATE Sink Current (Note 3)
ILGSNK
LGATE - PGND = 2.5V
-
4.0
-
A
UGATE to LGATE Deadtime
tUGFLGR
UGATE falling to LGATE rising, no load
-
21
-
ns
LGATE to UGATE Deadtime
tLGFUGR
LGATE falling to UGATE rising, no load
-
21
-
ns
BOOTSTRAP DIODE
Forward Voltage
VF
PVCC = 5V, IF = 2mA
-
0.58
-
V
Reverse Leakage
IR
VR = 25V
-
0.2
-
µA
2.0
-
-
V
-
-
1.0
V
-
0.1
1.0
µA
CONTROL INPUTS
EN High Threshold
VENTHR
EN Low Threshold
VENTHF
EN Leakage
IENL
EN = 0V
IENH
EN = 5.0V
1.4
2
2.5
µA
EN = 5.0V TA = -10°C to +100°C
1.5
2
2.5
µA
-1.75
0
1.75
mV
PROTECTION
OCSET-VO Threshold
VOCSETTHR
OCSET 10µA Current Source
IOCSET
EN = 5V
8.8
10
10.5
µA
EN = 5V TA = -10°C to +100°C
9
10
10.5
µA
EN = 0V
-
0
-
µA
-
600
-
kΩ
OCSET 10µA Current Source Impedance ROCSETIMP EN = 5V, OCSET = 1.2V
VUV
81
86
87
%
OVP Rising Threshold
VOVR
113
116
120
%
OVP Falling Threshold
VOVF
100
102
106
%
UVP Threshold
5
FN9095.2
May 7, 2008
ISL6228
Electrical Specifications
These specifications apply for TA = -40°C to +100°C; All typical specifications TA = +25°C, VCC = 5V,
PVCC = 5V; Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified.
Temperature limits established by characterization and are not production tested. (Continued)
PARAMETER
SYMBOL
OTP Rising Threshold (Note 3)
OTP Hysteresis (Note 3)
MIN
TYP
MAX
UNIT
TOTR
-
150
-
°C
TOTHYS
-
25
-
°C
Functional Pin Descriptions
GND (Bottom Pad)
Signal common of the IC. Unless otherwise stated, signals
are referenced to the GND pin.
FSET2 (Pin 1)
TEST CONDITIONS
connected across the VO1, FB1, and GND pins. Select the
resistor values such that FB1 to GND is 600mV when the
converter output voltage is at the programmed regulation
value.
VO1 (Pin 9)
The FSET2 pin programs the PWM switching frequency of
Channel 2. Program the desired PWM frequency with a
resistor and a capacitor connected across the FSET2 and
GND pins.
The VO1 pin measures the Channel 1 converter output
voltage and is used as an input to the Channel 1 R3 PWM
modulator. It also serves as part of Channel 1 inductor
current sensing and the OCP overcurrent fault protection
circuit.
VIN2 (Pin 2)
OCSET1 (Pin 10)
The VIN2 pin measures the input voltage of the Channel 2
converter. It is a required input to the Channel 2 R3 PWM
modulator. Connect the VIN2 pin to the drain of the
Channel 2 high-side MOSFET.
The OCSET1 pin measures the Channel 1 inductor current
and programs the threshold of the OCP overcurrent fault
protection.
VCC2 (Pin 3)
The EN1 pin is the on/off switch of Channel 1. The soft-start
sequence begins when the EN1 pin is pulled above the
rising threshold voltage VENTHR and VCC1 is above the
power-on reset (POR) rising threshold voltage VVCC_THR .
When the EN1 pin is pulled below the falling threshold
voltage VENTHF PWM1 immediately stops.
The VCC2 pin is the input bias voltage for Channel 2.
Connect +5V to the VCC2 pin. Decouple with at least 1µF of
a MLCC capacitor from the VCC2 pin to the GND pin.
VCC1 (Pin 4)
The VCC1 pin is the input bias voltage for Channel 1.
Connect +5V to the VCC1 pin. Decouple with at least 1µF of
a MLCC capacitor from the VCC1 pin to the GND pin.
VIN1 (Pin 5)
The VIN1 pin measures the input voltage of the Channel 1
converter. It is a required input to the Channel 1 R3 PWM
modulator. Connect the VIN1 pin to the drain of the
Channel 1 high-side MOSFET.
FSET1 (Pin 6)
The FSET1 pin programs the PWM switching frequency of
Channel 1. Program the desired PWM frequency with a
resistor and a capacitor connected across the FSET1 and
GND pins.
EN1 (Pin 11)
PHASE1 (Pin 12)
The PHASE1 pin is the current return path for the Channel 1
high-side MOSFET gate driver. Connect the PHASE1 pin to
the node consisting of the high-side MOSFET source, the
low-side MOSFET drain, and the output inductor of the
Channel 1 converter.
UGATE1 (Pin 13)
The UGATE1 pin is the output of the Channel 1 high-side
MOSFET gate driver. Connect the UGATE1 pin to the gate
of the Channel 1 converter high-side MOSFET.
BOOT1 (Pin 14)
The PGOOD1 pin is an open-drain output that indicates
when the Channel 1 converter is able to supply regulated
voltage. Connect the PGOOD1 pin to +5V through a pull-up
resistor.
The BOOT1 pin stores the input voltage for the Channel 1
high-side MOSFET gate driver. Connect an MLCC capacitor
across the BOOT1 and PHASE1 pins. The boot capacitor is
charged through an internal boot diode connected from the
PVCC1 pin to the BOOT1 pin, each time the PHASE1 pin
drops below PVCC1 minus the voltage dropped across the
internal boot diode.
FB1 (Pin 8)
PVCC1 (Pin 15)
The FB1 pin is the inverting input of the control-loop error
amplifier for Channel 1. The Channel 1 converter output
voltage regulates to 600mV from the FB1 pin to the GND pin.
Program the desired output voltage with a resistor network
The PVCC1 pin is the input voltage bias for the Channel 1
low-side MOSFET gate drivers. Connect +5V to the PVCC1
PGOOD1 (Pin 7)
6
FN9095.2
May 7, 2008
ISL6228
pin. Decouple with at least 1µF of an MLCC capacitor across
the PVCC1 and PGND1 pin.
LGATE1 (Pin 16)
The LGATE1 pin is the output of the Channel 1 converter
low-side MOSFET gate driver. Connect the LGATE1 pin to
the gate of the Channel 1 converter low-side MOSFET.
PGND1 (Pin 17)
The PGND1 pin is the current return path for the Channel 1
converter low-side MOSFET gate driver. Connect the
PGND1 pin to the source of the Channel 1 converter lowside MOSFET through a low impedance path, preferably in
parallel with the trace connecting the LGATE1 pin to the gate
of the Channel 1 converter low-side MOSFET.
PGND2 (Pin 18)
The PGND2 pin is the current return path for the Channel 2
converter low-side MOSFET gate driver. Connect the
PGND2 pin to the source of the Channel 2 converter lowside MOSFET through a low impedance path, preferably in
parallel with the trace connecting the LGATE2 pin to the gate
of the Channel 2 converter low-side MOSFET.
LGATE2 (Pin 19)
The LGATE2 pin is the output of the Channel 2 converter
low-side MOSFET gate driver. Connect to the gate of the
Channel 2 converter low-side MOSFET.
PVCC2 (Pin 20)
The PVCC2 pin is the input voltage bias for the Channel 2
low-side MOSFET gate drivers. Connect +5V to the PVCC2
pin. Decouple with at least 1µF of an MLCC capacitor across
the PVCC2 and PGND2 pin.
BOOT2 (Pin 21)
The BOOT2 pin stores the input voltage for the Channel 2
high-side MOSFET gate driver. Connect an MLCC capacitor
across the BOOT2 and PHASE2 pins. The boot capacitor is
charged through an internal boot diode connected from the
PVCC2 pin to the BOOT2 pin, each time the PHASE2 pin
drops below PVCC2 minus the voltage dropped across the
internal boot diode.
UGATE2 (Pin 22)
The UGATE2 pin is the output of the Channel 2 high-side
MOSFET gate driver. Connect to the gate of the Channel 2
converter high-side MOSFET.
PHASE2 (Pin 23)
The PHASE2 pin is the current return path for the Channel 2
high-side MOSFET gate driver. Connect the PHASE2 pin to
the node consisting of the high-side MOSFET source, the
low-side MOSFET drain, and the output inductor of the
Channel 2 converter.
7
EN2 (Pin 24)
The EN2 pin is the on/off switch of Channel 2. The soft-start
sequence begins when the EN2 pin is pulled above the
rising threshold voltage VENTHR and VCC2 is above the
power-on reset (POR) rising threshold voltage VVCC_THR .
When the EN2 pin is pulled below the falling threshold
voltage VENTHF, PWM2 immediately stops.
OCSET2 (Pin 25)
The OCSET2 pin measures the Channel 2 inductor current
and programs the threshold of the OCP overcurrent fault
protection.
VO2 (Pin 26)
The VO2 pin measures the Channel 2 converter output
voltage and is used as an input to the Channel 2 R3 PWM
modulator. It also serves as part of Channel 2 inductor
current sensing and the OCP overcurrent fault protection
circuit.
FB2 (Pin 27)
The FB2 pin is the inverting input of the control-loop error
amplifier for Channel 2. The Channel 2 converter output
voltage regulates to 600mV from the FB2 pin to the GND pin.
Program the desired output voltage with a resistor network
connected across the VO2, FB2, and GND pins. Select the
resistor values such that FB2 to GND is 600mV when the
converter output voltage is at the programmed regulation
value.
PGOOD2 (Pin 28)
The PGOOD2 pin is an open-drain output that indicates
when the Channel 2 converter is able to supply regulated
voltage. Connect the PGOOD2 pin to +5V through a pull-up
resistor.
Theory of Operation
Two Separate Channels
The ISL6228 is a dual channel controller. Pins 4~17 are
dedicated to Channel 1, and pins 1~3 and pins 18~28 are
dedicated to Channel 2. The two channels are identical and
almost entirely independent, with the exception of sharing
the GND pin. Unless otherwise stated, only an individual
channel is discussed, and the conclusion applies to both
channels.
Modulator
The ISL6228 modulator features Intersil’s R3 technology, a
hybrid of fixed frequency PWM control and variable
frequency hysteretic control. Intersil’s R3 technology can
simultaneously affect the PWM switching frequency and
PWM duty cycle in response to input voltage and output load
transients. The R3 modulator synthesizes an AC signal VR,
which is an analog representation of the output inductor
ripple current. The duty-cycle of VR is the result of charge
and discharge current through a ripple capacitor CR. The
FN9095.2
May 7, 2008
ISL6228
current through CR is provided by a transconductance
amplifier gm that measures the VIN and VO pin voltages.
The positive slope of VR can be written as Equation 1:
V RPOS = ( g m ) ⋅ ( V IN – V OUT ) ⁄ C R
(EQ. 1)
The negative slope of VR can be written as Equation 2:
V RNEG = g m ⋅ V OUT ⁄ C R
Soft-Start Delay tSS begins and the output voltage begins to
rise. The FB pin ramps to 0.6V in approximately 1.5ms and
the PGOOD pin goes to high impedance approximately
1.25ms after the FB pin voltage reaches 0.6V.
1.5ms
Vo
(EQ. 2)
VCC and PVCC
Where gm is the gain of the transconductance amplifier.
A window voltage VW is referenced with respect to the error
amplifier output voltage VCOMP, creating an envelope into
which the ripple voltage VR is compared. The amplitude of
VW is set by a resistor connected across the FSET and GND
pins. The VR, VCOMP, and VW signals feed into a window
comparator in which VCOMP is the lower threshold voltage
and VW is the higher threshold voltage. Figure 2 shows
PWM pulses being generated as VR traverses the VW and
VCOMP thresholds. The PWM switching frequency is
proportional to the slew rates of the positive and negative
slopes of VR; it is inversely proportional to the voltage
between VW and VCOMP.
Ripple Capacitor Voltage CR
Window Voltage VW
EN
FB
PGOOD
1.25ms
FIGURE 3. SOFT-START SEQUENCE
The PGOOD pin indicates when the converter is capable of
supplying regulated voltage. The PGOOD pin is an
undefined impedance if VCC has not reached the rising POR
threshold VCCR, or if VCC is below the falling POR threshold
VCCF. The ISL6228 features a unique fault-identification
capability that can drastically reduce trouble-shooting time
and effort. The pull-down resistance of the PGOOD pin
corresponds to the fault status of the controller. The PGOOD
pull-down resistance is 95Ω during soft-start or if an UVP
occurs, 30Ω for an OCP, or 60Ω for OVP.
TABLE 1. PGOOD PULL-DOWN RESISTANCE
Error Amplifier Voltage VCOMP
PWM
CONDITION
PGOOD RESISTANCE
VCC Below POR
Undefined
Soft-start or Undervoltage
90Ω
Overvoltage
60Ω
Overcurrent
30Ω
MOSFET Gate-Drive Outputs LGATE and UGATE
FIGURE 2. MODULATOR WAVEFORMS DURING LOAD
TRANSIENT
Power-On Reset
The ISL6228 is disabled until the voltage at the VCC pin has
increased above the rising power-on reset (POR) VCCR
threshold voltage. The controller will be disabled when the
voltage at the VCC pin decreases below the falling POR
VCCF threshold voltage.
EN, Soft-Start and PGOOD
The ISL6228 uses a digital soft-start circuit to ramp the
output voltage of the converter to the programmed regulation
setpoint at a predictable slew rate. The slew rate of the
soft-start sequence has been selected to limit the in-rush
current through the output capacitors as they charge to the
desired regulation voltage. When the EN pin is pulled above
the rising EN threshold voltage VENTHR, the PGOOD
8
The ISL6228 has internal gate-drivers for the high-side and
low-side N-Channel MOSFETs. The low-side gate-drivers
are optimized for low duty-cycle applications where the lowside MOSFET conduction losses are dominant, requiring a
low r DS(ON) MOSFET. The LGATE pull-down resistance is
small in order to clamp the gate of the MOSFET below the
VGS(th) at turnoff. The current transient through the gate at
turn-off can be considerable because the gate charge of a
low r DS(ON) MOSFET can be large. Adaptive shoot-through
protection prevents a gate-driver output from turning on until
the opposite gate-driver output has fallen below
approximately 1V. The dead-time shown in Figure 4 is
extended by the additional period that the falling gate voltage
stays above the 1V threshold. The typical dead-time is 21ns.
The high-side gate-driver output voltage is measured across
the UGATE and PHASE pins while the low-side gate-driver
output voltage is measured across the LGATE and PGND
FN9095.2
May 7, 2008
ISL6228
pins. The power for the LGATE gate-driver is sourced
directly from the PVCC pin. The power for the UGATE gatedriver is sourced from a “boot” capacitor connected across
the BOOT and PHASE pins. The boot capacitor is charged
from a 5V bias supply through a “boot diode” each time the
low-side MOSFET turns on, pulling the PHASE pin low. The
ISL6228 has an integrated boot diode connected from the
PVCC pin to the BOOT pin.
tLGFUGR
tUGFLGR
50%
UGATE
LGATE
detected positive voltage and LGATE was allowed to go high
for eight consecutive PWM switching cycles. The ISL6228
will turn off the low-side MOSFET once the phase voltage
turns positive, indicating negative inductor current. The
ISL6228 will return to CCM on the following cycle after the
PHASE pin detects negative voltage, indicating that the body
diode of the low-side MOSFET is conducting positive
inductor current.
Efficiency can be further improved with a reduction of
unnecessary switching losses by reducing the PWM
frequency. It is characteristic of the R3 architecture for the
PWM frequency to decrease while in diode emulation. The
extent of the frequency reduction is proportional to the
reduction of load current. Upon entering DEM, the PWM
frequency makes an initial step-reduction because of a 33%
step-increase of the window voltage V W.
Overcurrent Protection
The overcurrent protection (OCP) setpoint is programmed
with resistor ROCSET that is connected across the OCSET
and PHASE pins.
50%
L
DCR
FIGURE 4. LGATE AND UGATE DEAD-TIME
Diode Emulation
+
ISL6228
The ISL6228 implements forced continuous-conductionmode (CCM) at heavy load and diode-emulation-mode
(DEM) at light load, to optimize efficiency in the entire load
range. The transition is automatically achieved by detecting
the output load current.
Positive-going inductor current flows from either the source
of the high-side MOSFET, or the drain of the low-side
MOSFET. Negative-going inductor current flows into the
drain of the low-side MOSFET. When the low-side MOSFET
conducts positive inductor current, the phase voltage will be
negative with respect to the GND and PGND pins.
Conversely, when the low-side MOSFET conducts negative
inductor current, the phase voltage will be positive with
respect to the GND and PGND pins. The ISL6228 monitors
the phase voltage, when the low-side MOSFET is
conducting inductor current, to determine the direction of the
inductor current.
When the output load current is greater than or equal to ½
the inductor ripple current, the inductor current is always
positive, and the converter is always in CCM. The ISL6228
minimizes the conduction loss in this condition by forcing the
low-side MOSFET to operate as a synchronous rectifier.
When the output load current is less than ½ the inductor
ripple current, negative inductor current occurs. Sinking
negative inductor through the low-side MOSFET lowers
efficiency through unnecessary conduction losses. The
ISL6228 automatically enters DEM after the PHASE pin has
9
IL
PHASE
10µA
OCSET
ROCSET
+ VROCSET
VDCR
CSEN
VO
_
CO
_
RO
VO
FIGURE 5. OVERCURRENT-SET CIRCUIT
Figure 5 shows the overcurrent-set circuit. The inductor
consists of inductance L and the DC resistance DCR. The
inductor DC current IL creates a voltage drop across DCR,
given by Equation 3:
V DCR = I L • DCR
(EQ. 3)
The ISL6228 sinks 10µA current into the OCSET pin,
creating a DC voltage drop across the resistor ROCSET,
given by Equation 4:
V ROCSET = 10μA • R OCSET
(EQ. 4)
Resistor RO is connected between the VO pin and the actual
output voltage of the converter. During normal operation, the
VO pin is a high impedance path, therefore there is no
voltage drop across RO. The DC voltage difference between
the OCSET pin and the VO pin can be established using
Equation 5:
V OCSET – V VO = V DCR – V ROCSET = I L • DCR – 10μA • R OCSET
(EQ. 5)
FN9095.2
May 7, 2008
ISL6228
The ISL6228 monitors the OCSET pin and the VO pin
voltages. Once the OCSET pin voltage is higher than the VO
pin voltage for more than 10µs, the ISL6228 declares an OCP
fault. The value of ROCSET is then written as Equation 6:
I OC • DCR
R OCSET = --------------------------10μA
(EQ. 6)
Where:
- ROCSET (Ω) is the resistor used to program the
overcurrent setpoint
- IOC is the output current threshold that will activate the
OCP circuit
- DCR is the inductor DC resistance
For example, if IOC is 20A and DCR is 4.5mΩ, the choice of
ROCSET is ROCSET = 20A x 4.5mΩ/10µA = 9kΩ.
Resistor ROCSET and capacitor CSEN form an R-C network
to sense the inductor current. To sense the inductor current
correctly not only in DC operation, but also during dynamic
operation, the R-C network time constant ROCSETCSEN
needs to match the inductor time constant L/DCR. The value
of CSEN is then written as Equation 7:
L
C SEN = ----------------------------------------R OCSET • DCR
(EQ. 7)
For example, if L is 1.5µH, DCR is 4.5mΩ, and ROCSET is
9kΩ, the choice of CSEN = 1.5µH/(9kΩ x 4.5mΩ) = 0.037µF.
Upon converter startup, capacitor CSEN initial voltage is 0V.
To prevent false OCP, a 10µA current source flows out of the
VO pin during start up, generating a voltage drop on resistor
RO, which has the same resistance as ROCSET. When
PGOOD pin goes high, the VO pin current source will
terminate.
When an OCP fault is declared, the PGOOD pin will pull
down to 30Ω and latch off the converter. The fault will remain
latched until the EN pin has been pulled below the falling EN
threshold voltage VENTHF or if VCC has decayed below the
falling POR threshold voltage VVCC_THF.
Overvoltage Protection
The OVP fault detection circuit triggers after the FB pin voltage
is above the rising overvoltage threshold VOVR for more than
2µs. The FB pin voltage is 0.6V in normal operation. The rising
overvoltage threshold VOVR is typically 116%. That means if
the FB pin voltage is above 116% x 0.6V = 0.696V, for more
than 2µs, an OVP fault is declared.
When an OVP fault is declared, the PGOOD pin will pull
down to 60Ω and latch-off the converter. The OVP fault will
remain latched until the EN pin has been pulled below the
falling EN threshold voltage VENTHF or if VCC has decayed
below the falling POR threshold voltage VVCC_THF.
Although the converter has latched-off in response to an
OVP fault, the LGATE gate-driver output will retain the ability
to toggle the low-side MOSFET on and off, in response to
10
the output voltage transversing the VOVR and VOVF
thresholds. The LGATE gate-driver will turn on the low-side
MOSFET to discharge the output voltage, protecting the
load. The LGATE gate-driver will turn off the low-side
MOSFET once the FB pin voltage is lower than the falling
overvoltage threshold VOVF for more than 2µs. The falling
overvoltage threshold VOVF is typically 106%. That means if
the FB pin voltage falls below 106% x 0.6V = 0.636V, for
more than 2µs, the LGATE gate-driver will turn off the lowside MOSFET. If the output voltage rises again, the LGATE
driver will again turn on the low-side MOSFET when the FB
pin voltage is above the rising overvoltage threshold VOVR
for more than 2µs. By doing so, the ISL6228 protects the
load when there is a consistent overvoltage condition.
Undervoltage Protection
The UVP fault detection circuit triggers after the FB pin
voltage is below the undervoltage threshold VUV for more
than 2µs. The FB pin voltage is 0.6V in normal operation.
The undervoltage threshold VUV is typically 86%. That
means if the FB pin voltage is below 86% x 0.6V = 0.516V,
for more than 2µs, an UVP fault is declared, and the
PGOOD pin will pull down to 95Ω and latch-off the converter.
The fault will remain latched until the EN pin has been pulled
below the falling EN threshold voltage VENTHF or if VCC has
decayed below the falling POR threshold voltage
V
VCC_THF.
Programming the Output Voltage
When the converter is in regulation there will be 0.6V from
the FB pin to the GND pin. Connect a two-resistor voltage
divider across the VO pin and the GND pin with the output
node connected to the FB pin. Scale the voltage-divider
network such that the FB pin is 0.6V with respect to the GND
pin when the converter is regulating at the desired output
voltage. The output voltage can be programmed from 0.6V
to 5V.
Programming the output voltage is written as Equation 8:
R BOTTOM
V REF = V O • -------------------------------------------------R TOP + R BOTTOM
(EQ. 8)
Where:
- VO is the desired output voltage of the converter
- The voltage to which the converter regulates the FB pin
is the VREF
- RTOP is the voltage-programming resistor that connects
from the FB pin to the converter output. In addition to
setting the output voltage, this resistor is part of the loop
compensation network
- RBOTTOM is the voltage-programming resistor that
connects from the FB pin to the GND pin
Choose RTOP value first, and calculate RBOTTOM according
to Equation 9:
V REF • R
TOP
R BOTTOM = ----------------------------------V O – V REF
(EQ. 9)
FN9095.2
May 7, 2008
ISL6228
Programming the PWM Switching Frequency
General Application Design Guide
The ISL6228 does not use a clock signal to produce PWMs.
The PWM switching frequency fSW is programmed by the
resistor RFSET that is connected from the FSET pin to the
GND pin. The approximate PWM switching frequency is
written as Equation 10:
This design guide is intended to provide a high-level
explanation of the steps necessary to design a single-phase
power converter. It is assumed that the reader is familiar with
many of the basic skills and techniques referenced in the
following section. In addition to this guide, Intersil provides
complete reference designs that include schematics, bills of
materials, and example board layouts.
1
f SW = --------------------------K ⋅ R FSET
(EQ. 10)
Estimating the value of RFSET is written as Equation 11:
1
R FSET = -----------------K • f SW
(EQ. 11)
Where:
- fSW is the PWM switching frequency
- RFSET is the fSW programming resistor
- K = 1.5 x 10-10
Selecting the LC Output Filter
The duty cycle of an ideal buck converter is a function of the
input and the output voltage. This relationship is written as
Equation 13:
VO
D = --------V IN
It is recommended that whenever the control loop
compensation network is modified, fSW should be checked
for the correct frequency and if necessary, adjust RFSET.
(EQ. 13)
The output inductor peak-to-peak ripple current is written as
Equation 14:
VO • ( 1 – D )
I PP = -----------------------------f SW • L
(EQ. 14)
Compensation Design
Figure 6 shows the recommended Type-II compensation
circuit. The FB pin is the inverting input of the error amplifier.
The COMP signal, the output of the error amplifier, is inside the
chip and unavailable to users. CINT is a 100pF capacitor
integrated inside the IC, connecting across the FB pin and the
COMP signal. RTOP, RFB, CFB and CINT form the Type-II
compensator. The frequency domain transfer function is given
by Equation 12:
1 + s • ( R TOP + R FB ) • C
FB
G COMP ( s ) = ------------------------------------------------------------------------------------------s • R TOP • C INT • ( 1 + s • R FB • C )
(EQ. 12)
FB
CINT = 100pF
CFB
RFB
RTOP
-
VO
FB
EA
RBOTTOM
COMP
+
A typical step-down DC/DC converter will have an IP-P of
20% to 40% of the maximum DC output load current. The
value of IPP is selected based upon several criteria such as
MOSFET switching loss, inductor core loss, and the resistive
loss of the inductor winding. The DC copper loss of the
inductor can be estimated by Equation 15:
P COPPER = I LOAD
2
•
DCR
(EQ. 15)
Where ILOAD is the converter output DC current.
The copper loss can be significant so attention has to be
given to the DCR selection. Another factor to consider when
choosing the inductor is its saturation characteristics at
elevated temperature. A saturated inductor could cause
destruction of circuit components, as well as nuisance OCP
faults.
A DC/DC buck regulator must have output capacitance CO
into which ripple current IP-P can flow. Current IPP develops
a corresponding ripple voltage VP-P across CO, which is the
sum of the voltage drop across the capacitor ESR and of the
voltage change stemming from charge moved in and out of
the capacitor. These two voltages are written as
Equation 16:
REF
ΔV ESR = I P-P • E SR
ISL6228
FIGURE 6. COMPENSATION REFERENCE CIRCUIT
The LC output filter has a double pole at its resonant frequency
that causes rapid phase change. The R3 modulator used in the
ISL6228 makes the LC output filter resemble a first order
system in which the closed loop stability can be achieved with
the recommended Type-II compensation network. Intersil
provides a PC-based tool (example page is shown later) that
can be used to calculate compensation network component
values and help simulate the loop frequency response.
11
(EQ. 16)
and Equation 17:
I P-P
ΔV C = ----------------------------8 • CO • f
(EQ. 17)
SW
If the output of the converter has to support a load with high
pulsating current, several capacitors will need to be paralleled
to reduce the total ESR until the required VP-P is achieved.
The inductance of the capacitor can cause a brief voltage dip
if the load transient has an extremely high slew rate. Low
inductance capacitors should be considered. A capacitor
FN9095.2
May 7, 2008
ISL6228
dissipates heat as a function of RMS current and frequency.
Be sure that IP-P is shared by a sufficient quantity of paralleled
capacitors so that they operate below the maximum rated
RMS current at fSW. Take into account that the rated value of
a capacitor can fade as much as 50% as the DC voltage
across it increases.
Selection of the Input Capacitor
MOSFET Selection and Considerations
Typically, a MOSFET cannot tolerate even brief excursions
beyond their maximum drain to source voltage rating. The
MOSFETs used in the power stage of the converter should
have a maximum VDS rating that exceeds the sum of the
upper voltage tolerance of the input power source and the
voltage spike that occurs when the MOSFET switches off.
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and capable of
supplying the RMS current required by the switching circuit.
Their voltage rating should be at least 1.25 times greater
than the maximum input voltage, while a voltage rating of 1.5
times is a preferred rating. Figure 7 is a graph of the input
RMS ripple current, normalized relative to output load current,
as a function of duty cycle that is adjusted for converter
efficiency. The ripple current calculation is written as
Equation 18:
There are several power MOSFETs readily available that are
optimized for DC/DC converter applications. The preferred
high-side MOSFET emphasizes low gate charge so that the
device spends the least amount of time dissipating power in
the linear region. Unlike the low-side MOSFET which has the
drain-source voltage clamped by its body diode during turn
off, the high-side MOSFET turns off with VIN - VOUT, plus the
spike, across it. The preferred low-side MOSFET
emphasizes low r DS(ON) when fully saturated to minimize
conduction loss.
2
2 D
2
( I MAX ⋅ ( D – D ) ) + ⎛ x ⋅ I MAX ⋅ ------ ⎞
⎝
12 ⎠
I IN_RMS, NORMALIZED = ----------------------------------------------------------------------------------------------------I MAX
P CON_LS ≈ I LOAD ⋅ r DS ( ON )_LS • ( 1 – D )
(EQ. 18)
Where:
- IMAX is the maximum continuous ILOAD of the converter
- x is a multiplier (0 to 1) corresponding to the inductor
peak-to-peak ripple amplitude expressed as a
percentage of IMAX (0% to 100%)
- D is the duty cycle that is adjusted to take into account
the efficiency of the converter which is written as:
VO
D = -------------------------V IN ⋅ EFF
(EQ. 19)
NORMALIZED INPUT RMS RIPPLE CURRENT
In addition to the bulk capacitance, some low ESL ceramic
capacitance is recommended to decouple between the drain
of the high-side MOSFET and the source of the low-side
MOSFET.
0.60
0.55
0.50
2
(EQ. 20)
For the high-side (HS) MOSFET, the its conduction loss is
written as Equation 21:
P CON_HS = I LOAD
2
•
r DS ( ON )_HS • D
(EQ. 21)
For the high-side MOSFET, the switching loss is written as
Equation 22:
V IN • I VALLEY • t ON • f
V IN • I PEAK • t OFF • f
SW
SW
P SW_HS = ----------------------------------------------------------------- + ------------------------------------------------------------2
2
(EQ. 22)
Where:
- IVALLEY is the difference of the DC component of the
inductor current minus 1/2 of the inductor ripple current
- IPEAK is the sum of the DC component of the inductor
current plus 1/2 of the inductor ripple current
- tON is the time required to drive the device into
saturation
- tOFF is the time required to drive the device into cut-off
Selecting The Bootstrap Capacitor
0.45
The selection of the bootstrap capacitor is written as
Equation 23:
0.40
0.35
0.30
Qg
C BOOT = -----------------------ΔV BOOT
x=1
x = 0.75
x = 0.50
x = 0.25
x=0
0.25
0.20
0.15
0.05
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
DUTY CYCLE
FIGURE 7. NORMALIZED RMS INPUT CURRENT
12
(EQ. 23)
Where:
0.10
0
0
For the low-side (LS) MOSFET, the power loss can be
assumed to be conductive only and is written as Equation 20:
1.0
- Qg is the total gate charge required to turn on the
high-side MOSFET
- ΔVBOOT, is the maximum allowed voltage decay across
the boot capacitor each time the high-side MOSFET is
switched on
FN9095.2
May 7, 2008
ISL6228
As an example, suppose the high-side MOSFET has a total
gate charge Qg, of 25nC at VGS = 5V, and a ΔVBOOT of
200mV. The calculated bootstrap capacitance is 0.125µF; for
a comfortable margin, select a capacitor that is double the
calculated capacitance. In this example, 0.22µF will suffice.
Use an X7R or X5R ceramic capacitor.
Layout Considerations
As a general rule, power should be on the bottom layer of
the PCB and weak analog or logic signals are on the top
layer of the PCB. The ground-plane layer should be adjacent
to the top layer to provide shielding. The ground plane layer
should have an island located under the IC, the
compensation components, and the FSET components. The
island should be connected to the rest of the ground plane
layer at one point.
VIAS TO
GROUND
PLANE
GND
VOUT
INDUCTOR
PHASE
NODE
HIGH-SIDE
MOSFETS
VIN
OUTPUT
CAPACITORS
SCHOTTKY
DIODE
EN (Pins 11 and 24), and PGOOD (Pins 7 and 28)
These are logic signals that are referenced to the GND pin.
Treat as a typical logic signal.
OCSET (Pins 10 and 25)
The current-sensing network consisting of ROCSET and
CSEN needs to be connected to the inductor pads for
accurate measurement. Connect ROCSET to the phasenode side pad of the inductor, and connect CSEN to the
output side pad of the inductor. Connect the OCSET pin to
the common node of node of ROCSET and CSEN.
FB (Pins 8 and 27), and VO (Pins 9 and 26)
The VO pin is used to sense the inductor current for OCP.
Connect the VO pin to the output-side of CSEN through
resistor RO. The input impedance of the FB pin is high, so
place the voltage programming and loop compensation
components close to the VO, FB, and GND pins keeping the
high impedance trace short.
FSET (Pins 1 and 6)
LOW-SIDE
MOSFETS
This pin requires a quiet environment. The resistor RFSET
and capacitor CFSET should be placed directly adjacent to
this pin. Keep fast moving nodes away from this pin.
INPUT
CAPACITORS
LGATE (Pins 16 and 19)
FIGURE 8. TYPICAL POWER COMPONENT PLACEMENT
Signal Ground and Power Ground
The bottom of the ISL6228 TQFN package is the signal
ground (GND) terminal for analog and logic signals of the IC.
Connect the GND pad of the ISL6228 to the island of ground
plane under the top layer using several vias, for a robust
thermal and electrical conduction path. Connect the input
capacitors, the output capacitors, and the source of the
lower MOSFETs to the power ground plane.
PGND (Pins 17 and 18)
This is the return path for the pull-down of the LGATE
low-side MOSFET gate driver. Ideally, PGND should be
connected to the source of the low-side MOSFET with a
low-resistance, low-inductance path.
VIN (Pins 2 and 5)
The VIN pin should be connected close to the drain of the
high-side MOSFET, using a low resistance and low
inductance path.
VCC (Pins 3 and 4)
The signal going through this trace is both high dv/dt and
high di/dt, with high peak charging and discharging current.
Route this trace in parallel with the trace from the PGND pin.
These two traces should be short, wide, and away from
other traces. There should be no other weak signal traces in
proximity with these traces on any layer.
BOOT (Pins 14 and 21), UGATE (Pins 13 and 22), and
PHASE (Pins 12 and 23)
The signals going through these traces are both high dv/dt
and high di/dt, with high peak charging and discharging
current. Route the UGATE and PHASE pins in parallel with
short and wide traces. There should be no other weak signal
traces in proximity with these traces on any layer.
Copper Size for the Phase Node
The parasitic capacitance and parasitic inductance of the
phase node should be kept very low to minimize ringing. It is
best to limit the size of the PHASE node copper in strict
accordance with the current and thermal management of the
application. An MLCC should be connected directly across
the drain of the upper MOSFET and the source of the lower
MOSFET to suppress the turn-off voltage spike.
For best performance, place the decoupling capacitor very
close to the VCC and GND pins.
PVCC (Pins 15 and 20)
For best performance, place the decoupling capacitor very
close to the PVCC and respective PGND pins, preferably on
the same side of the PCB as the ISL6228 IC.
13
FN9095.2
May 7, 2008
ISL6228
Typical Performance
100
100
VIN = 8V
95
95
90
90
85
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 8V
VIN = 12V
VIN = 19V
80
75
85
80
75
70
70
65
65
60
VIN = 12V
VIN = 19V
60
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
6
7
8
IOUT (A)
IOUT (A)
FIGURE 9. CHANNEL 1 EFFICIENCY AT VO = 1.5V
VO1
FB1
FIGURE 10. CHANNEL 2 EFFICIENCY AT VO = 1.8V
VO1
FB1
PGOOD1
PGOOD1
PHASE1
PHASE1
FIGURE 11. START-UP, VIN = 12V, LOAD = 0.25Ω, VO = 1.05V
VO1
FIGURE 12. SHUT-DOWN, VIN = 12V, IO = 10A, VO = 1.05V
VO1
PHASE1
PHASE1
VO2
VO2
PHASE2
PHASE2
FIGURE 13. CCM STEADY-STATE OPERATION,VIN = 12V,
VO1 = 1.5V, IO1 = 3A, VO2 = 1.8A, IO2 = 4A
14
FIGURE 14. DCM STEADY-STATE OPERATION,VIN = 12V,
VO1 = 1.5V, IO1 = 1A, VO2 = 1.8V, IO2 = 1A
FN9095.2
May 7, 2008
ISL6228
Typical Performance (Continued)
IO1
IO2
VO1
VO2
PHASE2
PHASE1
FIGURE 15. TRANSIENT RESPONSE, VIN = 12V, VO = 1.5V,
IO = 0.1A/8.1A @ 2.55A/µs
FIGURE 16. TRANSIENT RESPONSE, VIN = 12V, VO = 1.8V,
IO = 0.1A/8.1A @ 2.55A/µs
IO1
IO2
VO1
VO2
PHASE1
FIGURE 17. LOAD INSERTION RESPONSE, VIN = 12V,
VO = 1.5V, IO = 0.1A/8.1A @ 2.55A/µs
IO1
VO1
PHASE1
FIGURE 19. LOAD RELEASE RESPONSE, VIN = 12V,
VO = 1.5V, IO = 0.1A/8.1A @ 2.55A/µs
PHASE2
FIGURE 18. LOAD INSERTION RESPONSE, VIN = 12V,
VO = 1.8V, IO = 0.1A/8.1A @ 2.55A/µs
IO2
VO2
PHASE2
FIGURE 20. LOAD RELEASE RESPONSE, VIN = 12V,
VO = 1.8V, IO = 0.1A/8.1A @ 2.55A/µs
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
15
FN9095.2
May 7, 2008
ISL6228
Package Outline Drawing
L28.4x4A
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 0, 3/07
4X 2.4
4.00
24X 0.40
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
28
22
1
4.00
21
2 .40 ± 0 . 15
15
(4X)
0.15
8
14
0.10 M C A B
4 28X 0.20
TOP VIEW
28X 0.45 ± 0.10
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 75
( 3. 75 TYP )
( 24X 0 . 4 )
(
C
BASE PLANE
SEATING PLANE
0.08 C
2. 40 )
SIDE VIEW
( 28X 0 . 20 )
C
0 . 2 REF
5
0 . 00 MIN.
0 . 05 MAX.
( 28X 0 . 65)
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
16
FN9095.2
May 7, 2008
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