DATASHEET

ISL6256, ISL6256A
®
Data Sheet
September 14, 2010
Highly Integrated Battery Charger with
Automatic Power Source Selector for
Notebook Computers
Features
• ±0.5% Charge Voltage Accuracy (-10°C to +100°C)
The ISL6256, ISL6256A is a highly integrated battery charger
controller for Li-ion/Li-ion polymer batteries. High Efficiency is
achieved by a synchronous buck topology and the use of a
MOSFET, instead of a diode, for selecting power from the
adapter or battery. The low side MOSFET emulates a diode at
light loads to improve the light load efficiency and prevent
system bus boosting.
The constant output voltage can be selected for 2, 3 and 4
series Li-ion cells with 0.5% accuracy over-temperature. It can
also be programmed between 4.2V + 5%/cell and
4.2V - 5%/cell to optimize battery capacity. When supplying
the load and battery charger simultaneously, the input current
limit for the AC adapter is programmable to within 3%
accuracy to avoid overloading the AC adapter, and to allow
the system to make efficient use of available adapter power
for charging. It also has a wide range of programmable
charging current. The ISL6256, ISL6256A provides outputs
that are used to monitor the current drawn from the AC
adapter, and monitor for the presence of an AC adapter. The
ISL6256, ISL6256A automatically transitions from regulating
current mode to regulating voltage mode.
ISL6256, ISL6256A has a feature for automatic power source
selection by switching to the battery when the AC adapter is
removed or switching to the AC adapter when the AC adapter is
available. It also provides a DC adapter monitor to support
aircraft power applications with the option of no battery charging.
Ordering Information
PART
NUMBER
(Notes 1, 2)
FN6499.3
• ±3% Accurate Input Current Limit
• ±3% Accurate Battery Charge Current Limit
• ±25% Accurate Battery Trickle Charge Current Limit
• Programmable Charge Current Limit, Adapter Current
Limit and Charge Voltage
• Fixed 300kHz PWM Synchronous Buck Controller with
Diode Emulation at Light Load
• Overvoltage Protection
• Output for Current Drawn from AC Adapter
• AC Adapter Present Indicator
• Fast Input Current Limit Response
• Input Voltage Range 7V to 25V
• Support 2-, 3- and 4-Cells Battery Pack
• Up to 17.64V Battery-Voltage Set Point
• Control Adapter Power Source Select MOSFET
• Thermal Shutdown
• Aircraft Power Capable
• DC Adapter Present Indicator
• Battery Discharge MOSFET Control
• Less than 10µA Battery Leakage Current
• Supports Pulse Charging
• Pb-free (RoHS Compliant)
TEMP
RANGE
(°C)
PART
MARKING
PACKAGE
(Pb-free)
PKG.
DWG. #
Applications
ISL6256HRZ
ISL 6256HRZ
-10 to +100 28 Ld 5x5 QFN L28.5x5
• Notebook, Desknote and Sub-notebook Computers
ISL6256HAZ
ISL 6256HAZ
-10 to +100 28 Ld QSOP
• Personal Digital Assistant
M28.15
ISL6256AHRZ ISL6256 AHRZ -10 to +100 28 Ld 5x5 QFN L28.5x5
ISL6256AHAZ
ISL6256 AHAZ -10 to +100 28 Ld QSOP
M28.15
NOTES:
1. Add “-T” suffix for tape and reel. Please refer to TB347 for details on
reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pbfree material sets, molding compounds/die attach materials, and
100% matte tin plate plus anneal (e3 termination finish, which is
RoHS compliant and compatible with both SnPb and Pb-free
soldering operations). Intersil Pb-free products are MSL classified at
Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information
page for ISL6256, ISL6256A. For more information on MSL please
see techbrief TB363.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2007, 2008, 2010. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6256, ISL6256A
Pinouts
ISL6256, ISL6256A
(28 LD QSOP)
TOP VIEW
28
27
26
25
24
23
CSON
ACPRN
DCPRN
DCIN
VDD
ACSET
DCSET
ISL6256, ISL6256A
(28 LD QFN)
TOP VIEW
22
EN
1
21
CSOP
CELLS
2
20
CSIN
ICOMP
3
19
CSIP
VCOMP
4
18
SGATE
ICM
5
17
BGATE
15
2
DCPRN
VDD
2
27
ACPRN
ACSET
3
26
CSON
DCSET
4
25
CSOP
EN
5
24
CSIN
CELLS
6
23
CSIP
ICOMP
7
22
SGATE
VCOMP
8
21
BGATE
ICM
9
20
PHASE
VREF
10
19
UGATE
CHLIM
11
18
BOOT
ACLIM
12
17
VDDP
16
LGATE
15
PGND
UGATE
12
13
14
VADJ
13
BOOT
11
28
VDDP
10
PGND
9
GND
8
PHASE
1
LGATE
7
VADJ
CHLIM
16
6
ACLIM
VREF
DCIN
GND
14
FN6499.3
September 14, 2010
ISL6256, ISL6256A
SGATE
ICM
CSIP CSIN
+
ACSET
ACPRN
DCSET
X19.9
CA1
DCPRN
+
+
-
1.26V
1.26V
VREF
152kΩ
ADAPTER
CURRENT
LIMIT SET
ACLIM
-
CSON
gm3
BGATE
+
+
152kΩ
DCIN
LDO
REGULATOR
MIN
CURRENT
BUFFER
ICOMP
BOOT
300kHz
RAMP
MIN
VOLTAGE
BUFFER
VDD
-
UGATE
PWM
Σ
PHASE
+
VCOMP
VDDP
VREF
-0.25
LGATE
514kΩ
gm1
+
VADJ
-
288kΩ
PGND
gm2
+
-
514kΩ
32kΩ
2.1V
CELLS
VOLTAGE
SELECTOR
48kΩ
VDD
VREF
1.065V
16kΩ
CA2
X19.9
+
+
EN
REFERENCE
GND
FB
CSON
CSOP CHLIM
FIGURE 1. FUNCTIONAL BLOCK DIAGRAM
3
FN6499.3
September 14, 2010
ISL6256, ISL6256A
AC ADAPTER
Q3
VDD
R8
130k
1%
Q5
C8
C8
DCIN
0.1µF DCIN
R9
10.2k
1%
SGATE
ACSET
ACSET
VDDP
VDDP
R10
4.7
4.7Ω
3.3V
TO HOST
CONTROLLER
C6
R6
R22 22Ω
BGATE
BGATE
VDDP
BOOT
BOOT
C9
1µF
1
10nF
4.7k FLOATING
4.2V/CELL
C1:10
C1:10µF
Q4
UGATE
UGATE
ICOMP
ICOMP
PHASE
PHASE
VCOMP
VCOMP
LGATE
LGATE
Q1
C4
0.1µF
0.1
D1
OPTIONAL
Q2
VADJ
VADJ
PGND
PGND
EN
EN
CSOP
CSOP
R23
10k
C3
0.047µF
ACLIM
ACLIM
VREF
VREF
VREF
CELLS
CELLS
BAT+
TRICKLE
CHARGE
R11
130k
1%
VDD
4 CELLS
R7: 100Ω
GND
GND
C10
22µF
BATTERY
PACK
BAT-
ICM
ICM
CHLIM
CHLIM
R13
1.87k
1%
CSON
R1
20mΩ
CSON
CSON
R12 22Ω
R12
2.6A CHARGE LIMIT
20k 1% 253mA TRICKLE CHARGE
D3
1N914
L
4.7µH
R11 22Ω
CHARGE
ENABLE
VDD
D2
ACPRN
ACPRN
6.8nF
C5
SYSTEM LOAD
CSIN
CSIN
VDD
VDD
R5
100k
R2
20m
20mΩ
C2
0.1
0.1µF
ISL6256
ISL
ISL6256
ISL6256A
IS
ISL
C7
1µF
R21 2.2Ω
CSIP
CSIP
C11
3300pF
Q6
FIGURE 2. ISL6256, ISL6256A TYPICAL APPLICATION CIRCUIT WITH FIXED CHARGING PARAMETERS
4
FN6499.3
September 14, 2010
ISL6256, ISL6256A
ADAPTER
Q5
R14
100k
1%
C8
0.1µ
0.1
R9
10.2k
1%
DCIN
DCIN
ACSET
ACSET
SGATE
SGATE
C2
0.1
0.1µF
CSIN
ISLISLISL6256 CSIN
ISL6256A
ISL
ISL
BGATE
BGATE
VDDP
VDDP
C7
1µF
R10
4.7
4.7Ω
C9
1µF
1
VCC
R16
100k
R21 2.2Ω
CSIP
CSIP
DCSET
DCSET
R15
11.5k
1%
DIGITAL
INPUT
Q3
VDD
R8
130k
1%
Q4
ACPRN
ACPRN
DCPRN
DCPRN
VDD
D2
Q1
C4
0.1µF
0.1
PHASE
PHASE
R23
10k
D1
OPTIONAL
LGATE
LGATE
Q2
D/A OUTPUT
C1:10µF
C1:10
BOOT
BOOT
UGATE
UGATE
DIGITAL
INPUT
SYSTEM LOAD
R22 22Ω
VDDP
VDD
VDD
R5
100k
R2
20mΩ
20m
CHLIM
CHLIM
PGND
PGND
EN
EN
CSOP
CSOP
R11 22Ω
OUTPUT
R7: 100Ω
A/D INPUT
C3
0.047µF
ICM
ICM
C11
3300pF
VREF
5.15A INPUT
CURRENT LIMIT
HOST
R11, R12
R13: 10k
C6
6.8nF
CSON
R1
20mΩ
CSON
CSON
ACLIM
ACLIM
CELLS
CELLS
VADJ
VADJ
ICOMP
ICOMP
AVDD/VREF
BAT+
R12 22Ω
VREF
VREF
GND
GND
D3
1N914
L
4.7µH
GND
3 CELLS
FLOATING
4.2V/CELL
C10
22µF
BATTERY
PACK
VCOMP
VCOMP
R6
4.7k
SCL
SDL
A/D INPUT
GND
C5
10nF
SCL
SDL
TEMP
BAT-
FIGURE 3. ISL6256, ISL6256A TYPICAL APPLICATION CIRCUIT WITH µP CONTROL AND AIRCRAFT POWER SUPPORT
5
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Absolute Maximum Ratings
Thermal Information
DCIN, CSIP, CSON to GND. . . . . . . . . . . . . . . . . . . . . -0.3V to +28V
CSIP-CSIN, CSOP-CSON . . . . . . . . . . . . . . . . . . . . . -0.3V to +0.3V
CSIP-SGATE, CSIP-BGATE . . . . . . . . . . . . . . . . . . . . . -0.3V to 16V
PHASE to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -7V to 30V
BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +35V
BOOT to VDDP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -2V to 28V
ACLIM, ACPRN, CHLIM, DCPRN, VDD to GND. . . . . . . -0.3V to 7V
BOOT-PHASE, VDDP-PGND . . . . . . . . . . . . . . . . . . . . . -0.3V to 7V
ACSET and DCSET to GND (Note 4) . . . . . . . -0.3V to VDD +0.3V
ICM, ICOMP, VCOMP to GND. . . . . . . . . . . . . . -0.3V to VDD +0.3V
VREF, CELLS to GND . . . . . . . . . . . . . . . . . . . . -0.3V to VDD +0.3V
EN, VADJ, PGND to GND . . . . . . . . . . . . . . . . . -0.3V to VDD +0.3V
UGATE. . . . . . . . . . . . . . . . . . . . . . . . PHASE -0.3V to BOOT +0.3V
LGATE . . . . . . . . . . . . . . . . . . . . . . . . . PGND -0.3V to VDDP +0.3V
Thermal Resistance
θJA (°C/W)
θJC (°C/W)
QFN Package (Notes 5, 6). . . . . . . . . .
39
9.5
QSOP Package (Note 5) . . . . . . . . . . .
80
N/A
Junction Temperature Range. . . . . . . . . . . . . . . . . .-10°C to +150°C
Operating Temperature Range . . . . . . . . . . . . . . . .-10°C to +100°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
4. ACSET may be operated 1V below GND if the current through ACSET is limited to less than 1mA.
5. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
6. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
DCIN = CSIP = CSIN = 18V, CSOP = CSON = 12V, ACSET = DCSET = 1.5V, ACLIM = VREF,
VADJ = Floating, EN = VDD = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDD = 1µF, IVDD = 0mA,
TA = -10°C to +100°C, TJ ≤ +125°C, Unless Otherwise Noted. Boldface limits apply over the
operating temperature range, -10°C to +100°C.
PARAMETER
TEST CONDITIONS
MIN
(Note 8)
MAX
(Note 8)
UNITS
25
V
1.4
3
mA
3
10
µA
4.925
5.075
5.225
V
TYP
SUPPLY AND BIAS REGULATOR
DCIN Input Voltage Range
7
DCIN Quiescent Current
EN = VDD or GND, 7V ≤ DCIN ≤ 25V
Battery Leakage Current (Note 7)
DCIN = 0, no load
VDD Output Voltage/Regulation
7V ≤ DCIN ≤ 25V, 0 ≤ IVDD ≤ 30mA
VDD Undervoltage Lockout Trip Point
VDD Rising
4.0
4.4
4.6
V
Hysteresis
200
250
400
mV
2.365
2.39
2.415
V
Reference Output Voltage VREF
0 ≤ IVREF ≤ 300µA
Battery Charge Voltage Accuracy
CSON = 16.8V, CELLS = VDD, VADJ = Float
-0.5
0.5
%
CSON = 12.6V, CELLS = GND, VADJ = Float
-0.5
0.5
%
CSON = 8.4V, CELLS = Float, VADJ = Float
-0.5
0.5
%
CSON = 17.64V, CELLS = VDD,
VADJ = VREF
-0.5
0.5
%
CSON = 13.23V, CELLS = GND,
VADJ = VREF
-0.5
0.5
%
CSON = 8.82V, CELLS = Float, VADJ = VREF
-0.5
0.5
%
CSON = 15.96V, CELLS = VDD, VADJ = GND
-0.5
0.5
%
CSON = 11.97V, CELLS = GND, VADJ = GND
-0.5
0.5
%
CSON = 7.98V, CELLS = Float, VADJ = GND
-0.5
0.5
%
TRIP POINTS
ACSET Threshold
1.24
1.26
1.28
V
ACSET Input Bias Current Hysteresis
2.4
3.4
4.4
µA
6
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Electrical Specifications
DCIN = CSIP = CSIN = 18V, CSOP = CSON = 12V, ACSET = DCSET = 1.5V, ACLIM = VREF,
VADJ = Floating, EN = VDD = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDD = 1µF, IVDD = 0mA,
TA = -10°C to +100°C, TJ ≤ +125°C, Unless Otherwise Noted. Boldface limits apply over the
operating temperature range, -10°C to +100°C. (Continued)
PARAMETER
TEST CONDITIONS
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
ACSET Input Bias Current
ACSET ≥ 1.26V
2.4
3.4
4.4
µA
ACSET Input Bias Current
ACSET < 1.26V
-1
0
1
µA
DCSET Threshold
1.24
1.26
1.28
V
DCSET Input Bias Current Hysteresis
2.4
3.4
4.4
µA
DCSET Input Bias Current
DCSET ≥ 1.26V
2.4
3.4
4.4
µA
DCSET Input Bias Current
DCSET < 1.26V
-1
0
1
µA
245
300
355
kHz
OSCILLATOR
Frequency
PWM Ramp Voltage (peak-to-peak)
CSIP = 18V
1.6
V
CSIP = 11V
1
V
SYNCHRONOUS BUCK REGULATOR
Maximum Duty Cycle
97
99
99.6
%
3.0
Ω
UGATE Pull-up Resistance
BOOT-PHASE = 5V, 500mA source current
1.8
UGATE Source Current
BOOT-PHASE = 5V, BOOT-UGATE = 2.5V
1.0
UGATE Pull-down Resistance
BOOT-PHASE = 5V, 500mA sink current
1.0
UGATE Sink Current
BOOT-PHASE = 5V, UGATE-PHASE = 2.5V
1.8
LGATE Pull-up Resistance
VDDP-PGND = 5V, 500mA source current
1.8
LGATE Source Current
VDDP-PGND = 5V, VDDP-LGATE = 2.5V
1.0
LGATE Pull-down Resistance
VDDP-PGND = 5V, 500mA sink current
1.0
LGATE Sink Current
VDDP-PGND = 5V, LGATE = 2.5V
1.8
Dead Time
Falling UGATE to rising LGATE or
falling LGATE to rising UGATE
A
Ω
1.8
A
Ω
3.0
A
Ω
1.8
A
10
30
ns
0
18
V
CHARGING CURRENT SENSING AMPLIFIER
Input Common-Mode Range
Input Bias Current at CSOP
5 < CSOP < 18V
0.25
2
µA
Input Bias Current at CSON
5 < CSON < 18V
75
100
µA
3.6
V
CHLIM Input Voltage Range
0
ISL6256: CHLIM = 3.3V
ISL6256
CSOP to CSON Full-Scale Current Sense
ISL6256: CHLIM = 2.0V
Voltage
ISL6256: CHLIM = 0.2V
ISL6256A: CHLIM = 3.3V
ISL6256A
CSOP to CSON Full-Scale Current Sense
ISL6256A: CHLIM = 2.0V
Voltage
ISL6256A: CHLIM = 0.2V
160
165
170
mV
95
100
105
mV
5.0
10
15.0
mV
161.7
165
168.3
mV
97
100
103
mV
7.5
10
12.5
mV
ISL6256 CSOP to CSON Full-Scale
Current Sense Voltage Formula
Charge current limit mode
0.2V < CHLIM < 3.3V
CHLIM*50
-5
CHLIM*50
+5
mV
ISL6256A CSOP to CSON Full-Scale
Current Sense Voltage Formula
Charge current limit mode
0.2V < CHLIM < 3.3V
CHLIM*49.72
-2.4
CHLIM*50.28
+2.4
mV
CHLIM Input Bias Current
CHLIM = GND or 3.3V, DCIN = 0V
-1
1
µA
7
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Electrical Specifications
DCIN = CSIP = CSIN = 18V, CSOP = CSON = 12V, ACSET = DCSET = 1.5V, ACLIM = VREF,
VADJ = Floating, EN = VDD = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDD = 1µF, IVDD = 0mA,
TA = -10°C to +100°C, TJ ≤ +125°C, Unless Otherwise Noted. Boldface limits apply over the
operating temperature range, -10°C to +100°C. (Continued)
PARAMETER
TEST CONDITIONS
CHLIM Power-down Mode Threshold
Voltage
CHLIM rising
CHLIM Power-down Mode Hysteresis
Voltage
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
80
88
95
mV
15
25
40
mV
25
V
130
µA
ADAPTER CURRENT SENSING AMPLIFIER
Input Common-Mode Range
7
Input Bias Current at CSIP and CSIN
Combined
CSIP = CSIN = 25V
100
Input Bias Current at CSIN
0 < CSIN < DCIN
0.10
µA
ADAPTER CURRENT LIMIT THRESHOLD
CSIP to CSIN Full-Scale Current Sense
Voltage
ACLIM Input Bias Current
ACLIM = VREF
97
100
103
mV
ACLIM = Float
72
75
78
mV
ACLIM = GND
47
50
53
mV
ACLIM = VREF
10
16
20
µA
ACLIM = GND
-20
-16
-10
µA
VOLTAGE REGULATION ERROR AMPLIFIER
Error Amplifier Transconductance from
CSON to VCOMP
CELLS = VDD
30
µA/V
Charging Current Error Amplifier
Transconductance
50
µA/V
Adapter Current Error Amplifier
Transconductance
50
µA/V
CURRENT REGULATION ERROR AMPLIFIER
BATTERY CELL SELECTOR
CELLS Input Voltage for 4-Cell Select
4.3
V
CELLS Input Voltage for 3-Cell Select
CELLS Input Voltage for 2-Cell Select
2.1
2
V
4.2
V
MOSFET DRIVER
BGATE Pull-up Current
CSIP-BGATE = 3V
10
30
45
mA
BGATE Pull-down Current
CSIP-BGATE = 5V
2.7
4.0
5.0
mA
CSIP-BGATE Voltage High
8
9.6
11
V
CSIP-BGATE Voltage Low
-50
0
50
mV
-100
0
100
mV
250
300
400
mV
DCIN-CSON Threshold for CSIP-BGATE
Going High
DCIN = 12V, CSON Rising
DCIN-CSON Threshold Hysteresis
SGATE Pull-up Current
CSIP-SGATE = 3V
7
12
15
mA
SGATE Pull-down Current
CSIP-SGATE = 5V
50
160
370
µA
CSIP-SGATE Voltage High
8
9
11
V
CSIP-SGATE Voltage Low
-50
0
50
mV
CSIP-CSIN Threshold for CSIP-SGATE
Going High
2.5
8
13
mV
8
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Electrical Specifications
DCIN = CSIP = CSIN = 18V, CSOP = CSON = 12V, ACSET = DCSET = 1.5V, ACLIM = VREF,
VADJ = Floating, EN = VDD = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDD = 1µF, IVDD = 0mA,
TA = -10°C to +100°C, TJ ≤ +125°C, Unless Otherwise Noted. Boldface limits apply over the
operating temperature range, -10°C to +100°C. (Continued)
PARAMETER
TEST CONDITIONS
CSIP-CSIN Threshold Hysteresis
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
2
5
8
mV
VDD
V
LOGIC INTERFACE
EN Input Voltage Range
0
EN Threshold Voltage
Rising
1.030
1.06
1.100
V
Falling
0.985
1.000
1.025
V
Hysteresis
30
60
90
mV
EN Input Bias Current
EN = 2.5V
1.8
2.0
2.2
µA
ACPRN Sink Current
ACPRN = 0.4V
3
8
11
mA
ACPRN Leakage Current
ACPRN = 5V
0.5
µA
DCPRN Sink Current
DCPRN = 0.4V
11
mA
DCPRN Leakage Current
DCPRN = 5V
0.5
µA
ICM Output Accuracy
(VICM = 19.9 x (VCSIP-VCSIN))
CSIP-CSIN = 100mV
-3
0
+3
%
CSIP-CSIN = 75mV
-4
0
+4
%
CSIP-CSIN = 50mV
-5
0
+5
%
-0.5
3
8
-0.5
Thermal Shutdown Temperature
150
°C
Thermal Shutdown Temperature
Hysteresis
25
°C
NOTE:
7. This is the sum of currents in these pins (CSIP, CSIN, BOOT, UGATE, PHASE, CSOP, CSON) all tied to 16.8V. No current in pins EN, ACSET,
VADJ, CELLS, ACLIM, CHLIM.
8. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits are
established by characterization and are not production tested.
9
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Typical Operating Performance
DCIN = 20V, 4S2P Li-Battery, TA = +25°C, Unless Otherwise Noted.
0.3
0.0
-0.3
-0.6
0.10
VREF LOAD REGULATION ACCURACY (%)
VDD LOAD REGULATION ACCURACY (%)
0.6
0
5
10
15
20
0.08
0.06
0.04
0.02
0.00
40
0
100
LOAD CURRENT (mA)
200
300
400
LOAD CURRENT (µA)
FIGURE 4. VDD LOAD REGULATION
FIGURE 5. VREF LOAD REGULATION
10
100
9
96
7
EFFICIENCY (%)
| ACCURACY | (%)
8
6
5
4
3
2
92
VCSON = 8.4V
2 CELLS
88
VCSON = 12.6V
3 CELLS
84
VCSON = 16.8V
4 CELLS
80
1
0
0
10
20
30
40
50
60
70
80
90
100
76
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
LOAD CURRENT (A)
CSIP-CSIN (mV)
FIGURE 6. ACCURACY vs AC ADAPTER CURRENT
FIGURE 7. SYSTEM EFFICIENCY vs CHARGE CURRENT
LOAD
CURRENT
5A/div
DCIN
10V/div
ADAPTER
CURRENT
5A/div
ACSET
1V/div
CHARGE
CURRENT
2A/div
DCSET
1V/div
DCPRN
5V/div
LOAD STEP:
STEP: 0-4A
0A TO 4A
CHARGE CURRENT: 3A
AC
ADAPTER CURRENT
CURRENT LIMIT:
LIMIT: 5.15A
5.15A
AC ADAPTER
BATTERY
VOLTAGE
2V/div
ACPRN
5V/div
FIGURE 8. AC AND DC ADAPTER DETECTION
10
FIGURE 9. LOAD TRANSIENT RESPONSE
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Typical Operating Performance
DCIN = 20V, 4S2P Li-Battery, TA = +25°C, Unless Otherwise Noted. (Continued)
CSON
5V/div
INDUCTOR
CURRENT
2A/div
EN
5V/div
BATTERY
BATTERY
INSERTION
INSERTION
BATTERY
BATTERY
REMOVAL
REMOVAL
CSON
10V/div
INDUCTOR
CURRENT
2A/div
CHARGE
CURRENT
2A/div
FIGURE 10. CHARGE ENABLE AND SHUTDOWN
VCOMP
VCOMP
VCOMP
2V/div
ICOMP
ICOMP
ICOMP
2V/div
FIGURE 11. BATTERY INSERTION AND REMOVAL
CHLIM
= 0.2V
CHLIM=0.2V
CSON = 8V
CSON=8V
PHASE
10V/div
PHASE
10V/div
INDUCTOR
CURRENT
1A/div
UGATE
2V/div
UGATE
5V/div
FIGURE 12. SWITCHING WAVEFORMS AT DIODE EMULATION
LGATE
2V/div
FIGURE 13. SWITCHING WAVEFORMS IN CC MODE
BGATE-CSIP
2V/div
SGATE-CSIP
2V/div
ADAPTER REMOVAL
REMOVAL
ADAPTER
SYSTEM BUS
VOLTAGE
10V/div
SYSTEM BUS
VOLTAGE
10V/div
SGATE-CSIP
2V/div
BGATE-CSIP
2V/div
INDUCTOR
CURRENT
2A/div
FIGURE 14. AC ADAPTER REMOVAL
11
ADAPTER
INSERTION
ADAPTER
INSERTION
INDUCTOR
CURRENT
2A/div
FIGURE 15. AC ADAPTER INSERTION
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Typical Operating Performance
DCIN = 20V, 4S2P Li-Battery, TA = +25°C, Unless Otherwise Noted. (Continued)
CHARGE
CURRENT
1A/div
CHLIM
1V/div
FIGURE 16. TRICKLE TO FULL-SCALE CHARGING
Functional Pin Descriptions
BOOT
Connect BOOT to a 0.1µF ceramic capacitor to PHASE pin
and connect to the cathode of the bootstrap Schottky diode.
compared with the AC adapter current limit to regulate the
AC adapter current.
GND
GND is an analog ground.
UGATE
DCIN
UGATE is the high side MOSFET gate drive output.
The DCIN pin is the input of the internal 5V LDO. Connect it
to the AC adapter output. Connect a 0.1µF ceramic
capacitor from DCIN to CSON.
SGATE
SGATE is the AC adapter power source select output. The
SGATE pin drives an external P-MOSFET used to switch to
AC adapter as the system power source.
BGATE
Battery power source select output. This pin drives an
external P-Channel MOSFET used to switch the battery as
the system power source. When the voltage at CSON pin is
higher than the AC adapter output voltage at DCIN, BGATE
is driven to low and selects the battery as the power source.
LGATE
ACSET
ACSET is an AC adapter detection input. Connect to a
resistor divider from the AC adapter output.
ACPRN
Open-drain output signals AC adapter is present. ACPRN
pulls low when ACSET is higher than 1.26V; and pulled high
when ACSET is lower than 1.26V.
DCSET
LGATE is the low side MOSFET gate drive output; swing
between 0V and VDDP.
DCSET is a lower voltage adapter detection input (like
aircraft power 15V). Allows the adapter to power the system
where battery charging has been disabled.
PHASE
DCPRN
The Phase connection pin connects to the high side
MOSFET source, output inductor, and low side MOSFET
drain.
Open-drain output signals DC adapter is present. DCPRN
pulls low when DCSET is higher than 1.26V; and pulled high
when DCSET is lower than 1.26V.
CSOP/CSON
EN
CSOP/CSON is the battery charging current sensing
positive/negative input. The differential voltage across CSOP
and CSON is used to sense the battery charging current,
and is compared with the charging current limit threshold to
regulate the charging current. The CSON pin is also used as
the battery feedback voltage to perform voltage regulation.
EN is the Charge Enable input. Connecting EN to high
enables the charge control function, connecting EN to low
disables charging functions. Use with a thermistor to detect
a hot battery and suspend charging.
CSIP/CSIN
ICM
ICM is the adapter current output. The output of this pin
produces a voltage proportional to the adapter current.
CSIP/CSIN is the AC adapter current sensing
positive/negative input. The differential voltage across CSIP
and CSIN is used to sense the AC adapter current, and is
12
FN6499.3
September 14, 2010
ISL6256, ISL6256A
PGND
PGND is the power ground. Connect PGND to the source of
the low side MOSFET.
VDD
VDD is an internal LDO output to supply IC analog circuit.
Connect a 1µF ceramic capacitor to ground.
VDDP
VDDP is the supply voltage for the low-side MOSFET gate
driver. Connect a 4.7Ω resistor to VDD and a 1µF ceramic
capacitor to power ground.
ICOMP
ICOMP is a current loop error amplifier output.
VCOMP
VCOMP is a voltage loop amplifier output.
CELLS
This pin is used to select the battery voltage. CELLS = VDD
for a 4S battery pack, CELLS = GND for a 3S battery pack,
CELLS = Float for a 2S battery pack.
VADJ
VADJ adjusts battery regulation voltage. VADJ = VREF for
4.2V + 5%/cell; VADJ = Floating for 4.2V/cell; VADJ = GND
for 4.2V - 5%/cell. Connect to a resistor divider to program
the desired battery cell voltage between 4.2V - 5% and
4.2V + 5%.
CHLIM
CHLIM is the battery charge current limit set pin. CHLIM
input voltage range is 0.1V to 3.6V. When CHLIM = 3.3V, the
set point for CSOP-CSON is 165mV. The charger shuts
down if CHLIM is forced below 88mV.
ACLIM
ACLIM is the adapter current limit set pin. ACLIM = VREF for
100mV, ACLIM = Floating for 75mV, and ACLIM = GND for
50mV. Connect a resistor divider to program the adapter
current limit threshold between 50mV and 100mV.
VREF
VREF is a 2.39V reference output pin. It is internally
compensated. Do not connect a decoupling capacitor.
Theory of Operation
Introduction
Note: Unless otherwise noted, all descriptions that refer to
the ISL6256 also refer to the ISL6256A.
The ISL6256 includes all of the functions necessary to
charge 2 to 4 cell Li-ion and Li-polymer batteries. A high
efficiency synchronous buck converter is used to control the
charging voltage and charging current up to 10A. The
ISL6256 has input current limiting and analog inputs for
setting the charge current and charge voltage; CHLIM inputs
13
are used to control charge current and VADJ inputs are used
to control charge voltage.
The ISL6256 charges the battery with constant charge
current, set by CHLIM input, until the battery voltage rises up
to a programmed charge voltage set by VADJ input; then the
charger begins to operate at a constant voltage charge mode.
The charger also drives an adapter isolation P-Channel
MOSFET to efficiently switch in the adapter supply.
ISL6256 is a complete power source selection controller for
single battery systems and also aircraft power applications.
It drives a battery selector P-Channel MOSFET to efficiently
select between a single battery and the adapter. It controls
the battery discharging MOSFET and switches to the battery
when the AC adapter is removed, or, switches to the AC
adapter when the AC adapter is inserted for single battery
system.
The EN input allows shutdown of the charger through a
command from a micro-controller. It also uses EN to safely
shutdown the charger when the battery is in extremely hot
conditions. The amount of adapter current is reported on the
ICM output. Figure 1 shows the IC functional block diagram.
The synchronous buck converter uses external N-Channel
MOSFETs to convert the input voltage to the required
charging current and charging voltage. Figure 2 shows the
ISL6256 typical application circuit with charging current and
charging voltage fixed at specific values. The typical
application circuit shown in Figure 3 shows the ISL6256
typical application circuit which uses a micro-controller to
adjust the charging current set by CHLIM input for aircraft
power applications. The voltage at CHLIM and the value of
R1 sets the charging current. The DC/DC converter
generates the control signals to drive two external
N-Channel MOSFETs to regulate the voltage and current set
by the ACLIM, CHLIM, VADJ and CELLS inputs.
The ISL6256 features a voltage regulation loop (VCOMP)
and two current regulation loops (ICOMP). The VCOMP
voltage regulation loop monitors CSON to ensure that its
voltage never exceeds the voltage and regulates the battery
charge voltage set by VADJ. The ICOMP current regulation
loops regulate the battery charging current delivered to the
battery to ensure that it never exceeds the charging current
limit set by CHLIM; and the ICOMP current regulation loops
also regulate the input current drawn from the AC adapter to
ensure that it never exceeds the input current limit set by
ACLIM, and to prevent a system crash and AC adapter
overload.
PWM Control
The ISL6256 employs a fixed frequency PWM current mode
control architecture with a feed-forward function. The
feed-forward function maintains a constant modulator gain of
11 to achieve fast line regulation as the buck input voltage
changes. When the battery charge voltage approaches the
input voltage, the DC/DC converter operates in dropout
FN6499.3
September 14, 2010
ISL6256, ISL6256A
mode, where there is a timer to prevent the frequency from
dropping into the audible frequency range. It can achieve
duty cycle of up to 99.6%.
To minimize accuracy loss due to interaction with VADJ's
internal resistor divider, ensure the AC resistance looking
back into the external resistor divider is less than 25k.
To prevent boosting of the system bus voltage, the battery
charger operates in standard-buck mode when CSOP-CSON
drops below 4.25mV. Once in standard-buck mode, hysteresis
does not allow synchronous operation of the DC/DC converter
until CSOP-CSON rises above 12.5mV.
Connect CELLS as shown in Table 1 to charge 2, 3 or 4 Li+
cells. When charging other cell chemistries, use CELLS to
select an output voltage range for the charger. The internal
error amplifier gm1 maintains voltage regulation. The voltage
error amplifier is compensated at VCOMP. The component
values shown in Figure 3 provide suitable performance for most
applications. Individual compensation of the voltage regulation
and current-regulation loops allows for optimal compensation.
An adaptive gate drive scheme is used to control the dead
time between two switches. The dead time control circuit
monitors the LGATE output and prevents the upper side
MOSFET from turning on until LGATE is fully off, preventing
cross-conduction and shoot-through. In order for the dead
time circuit to work properly, there must be a low resistance,
low inductance path from the LGATE driver to MOSFET
gate, and from the source of MOSFET to PGND. The
external Schottky diode is between the VDDP pin and BOOT
pin to keep the bootstrap capacitor charged.
TABLE 1. CELL NUMBER PROGRAMMING
CELLS
CELL NUMBER
VDD
4
GND
3
Float
2
Setting the Battery Regulation Voltage
Setting the Battery Charge Current Limit
The ISL6256 uses a high-accuracy trimmed band-gap
voltage reference to regulate the battery charging voltage.
The VADJ input adjusts the charger output voltage, and the
VADJ control voltage can vary from 0 to VREF, providing a
10% adjustment range (from 4.2V - 5% to 4.2V + 5%) on
CSON regulation voltage. An overall voltage accuracy of
better than 0.5% is achieved.
The CHLIM input sets the maximum charging current. The
current set by the current sense-resistor connects between
CSOP and CSON. The full-scale differential voltage between
CSOP and CSON is 165mV for CHLIM = 3.3V, so the
maximum charging current is 4.125A for a 40mΩ sensing
resistor. Other battery charge current-sense threshold
values can be set by connecting a resistor divider from
VREF or 3.3V to ground, or by connecting a low impedance
voltage source like a D/A converter in the micro-controller.
Unlike VADJ and ACLIM, CHLIM does not have an internal
resistor divider network. The charge current limit threshold is
given by Equation 3:
The per-cell battery termination voltage is a function of the
battery chemistry (consult the battery manufacturers to
determine this voltage)
• Float VADJ to set the battery voltage VCSON = 4.2V ×
number of the cells
• Connect VADJ to VREF to set 4.41V × number of cells
• Connect VADJ to ground to set 3.99V × number of the
cells
so, the maximum battery voltage of 17.6V can be achieved.
Note that other battery charge voltages can be set by
connecting a resistor divider from VREF to ground. The resistor
divider should be sized to draw no more than 100µA from
VREF; or connect a low impedance voltage source like the D/A
converter in the micro-controller. The programmed battery
voltage per cell can be determined by Equation 1:
V CELL = 0.175 ⋅ V VADJ + 3.99V
(EQ. 1)
An external resistor divider from VREF sets the voltage at
VADJ according to Equation 2:
R bot_VADJ || 514kΩ
V VADJ = VREF × ----------------------------------------------------------------------------------------------------------------R top_VADJ || 514kΩ + R bot_VADJ || 514kΩ
(EQ. 2)
14
165mV V CHLIM
I CHG = ⎛ -------------------⎞ ⎛ ----------------------⎞
⎝ R
⎠ ⎝ 3.3V ⎠
(EQ. 3)
1
To set the trickle charge current for the dumb charger, an
A/D output controlled by the micro-controller is connected to
CHLIM pin. The trickle charge current is determined by
Equation 4:
165mV V CHLIM ,trickle
I CHG = ⎛ -------------------⎞ ⎛ ----------------------------------------⎞
⎝ R
⎠⎝
⎠
3.3V
(EQ. 4)
1
When the CHLIM voltage is below 88mV (typical), it will
disable the battery charge. When choosing the current
sensing resistor, note that the voltage drop across the
sensing resistor causes further power dissipation, reducing
efficiency. However, adjusting CHLIM voltage to reduce the
voltage across the current sense resistor R1 will degrade
accuracy due to the smaller signal to the input of the current
sense amplifier. There is a trade-off between accuracy and
power dissipation. A low pass filter is recommended to
eliminate switching noise. Connect the resistor to the CSOP
pin instead of the CSON pin, as the CSOP pin has lower
bias current and less influence on current-sense accuracy
and voltage regulation accuracy.
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Charge Current Limit Accuracy
The “Electrical Specifications” table on page 7 gives
minimum and maximum values for the CSOP-CSON voltage
resulting from IC variations at 3 different CHLIM voltages
(see “CSOP to CSON Full-Scale Current Sense Voltage” on
page 7). It also gives formulae for calculating the minimum
and maximum CSOP-CSON voltage at any CHLIM voltage.
Equation 5 shows the formula for the max full scale
CSOP-CSON voltage (in mV) for the ISL6256A:
ISL6256A
( CSOP – CSON ) MAX = CHLIM • 50.28 + 2.4
(EQ. 5)
( CSOP – CSON ) MIN = CHLIM • 49.72 – 2.4
Equation 5 shows the formula for the max full scale CSOPCSON voltage (in mV) for the ISL6256:
ISL6256
MAX ( CSOP – CSON ) = CHLIM • 50 + 5
(EQ. 6)
MIN ( CSOP – CSON ) = CHLIM • 50 – 5
With CHLIM = 1.5V, the maximum CSOP-CSON voltage is
78mV and the minimum CSOP-CSON voltage is 72mV.
When ISL6256A is in charge current limiting mode, the
maximum charge current is the maximum CSOP-CSON
voltage divided by the minimum sense resistor. This can be
calculated for ISL6256A with Equation 7:
ISL6256A
I CHG, MAX = ( CHLIM • 50.28 + 2.4 ) ⁄ R 1min
I CHG, MIN = ( CHLIM • 49.72 – 2.4 ) ⁄ R 1max
(EQ. 7)
Maximum charge current can be calculated for ISL6256 with
Equation 8:
ISL6256
I CHG, MAX = ( CHLIM • 50 + 5 ) ⁄ R 1min
(EQ. 8)
I CHG, MIN = ( CHLIM • 50 – 5 ) ⁄ R 1max
With CHLIM = 0.7V and R1 = 0.02Ω, 1%:
ISL6256A
I CHG, MAX = ( 1.5V • 50.28 + 2.4 ) ⁄ 0.0198 = 3930mA
I CHG, MIN = ( 1.5V • 49.72 – 2.4 ) ⁄ 0.0202 = 3573mA
(EQ. 9)
Setting the Input Current Limit
The total input current from an AC adapter, or other DC
source, is a function of the system supply current and the
battery-charging current. The input current regulator limits
the input current by reducing the charging current, when the
input current exceeds the input current limit set point.
System current normally fluctuates as portions of the system
are powered up or down. Without input current regulation,
the source must be able to supply the maximum system
current and the maximum charger input current
simultaneously. By using the input current limiter, the current
15
capability of the AC adapter can be lowered, reducing
system cost.
The ISL6256 limits the battery charge current when the input
current-limit threshold is exceeded, ensuring the battery
charger does not load down the AC adapter voltage. This
constant input current regulation allows the adapter to fully
power the system and prevent the AC adapter from
overloading and crashing the system bus.
An internal amplifier gm3 compares the voltage between
CSIP and CSIN to the input current limit threshold voltage
set by ACLIM. Connect ACLIM to REF, Float and GND for
the full-scale input current limit threshold voltage of 100mV,
75mV and 50mV, respectively, or use a resistor divider from
VREF to ground to set the input current limit as Equation 10:
0.05
1
I INPUT = ------- ⋅ ⎛ ----------------- ⋅ V ACLIM + 0.05⎞
⎠
R 2 ⎝ VREF
(EQ. 10)
An external resistor divider from VREF sets the voltage at
ACLIM according to Equation 11:
R bot, ACLIM || 152kΩ
⎛
⎞
V ACLIM = VREF ⋅ ⎜ ------------------------------------------------------------------------------------------------------------------------⎟
||
||
R
152kΩ
+
R
152kΩ
⎝ top, ACLIM
⎠
bot, ACLIM
(EQ. 11)
where Rbot_ACLIM and Rtop_ACLIM are external resistors at
ACLIM.
To minimize accuracy loss due to interaction with ACLIM's
internal resistor divider, ensure the AC resistance looking
back into the resistor divider is less than 25k.
When choosing the current sense resistor, note that the
voltage drop across this resistor causes further power
dissipation, reducing efficiency. The AC adapter current
sense accuracy is very important. Use a 1% tolerance
current-sense resistor. The highest accuracy of ±3% is
achieved with 100mV current-sense threshold voltage for
ACLIM = VREF, but it has the highest power dissipation. For
example, it has 400mW power dissipation for rated 4A AC
adapter and 1Ω sensing resistor may have to be used. ±4%
and ±6% accuracy can be achieved with 75mV and 50mV
current-sense threshold voltage for ACLIM = Floating and
ACLIM = GND, respectively.
A low pass filter is suggested to eliminate the switching
noise. Connect the resistor to CSIN pin instead of CSIP pin
because CSIN pin has lower bias current and less influence
on the current-sense accuracy.
AC Adapter Detection
Connect the AC adapter voltage through a resistor divider to
ACSET to detect when AC power is available, as shown in
Figure 2. ACPRN is an open-drain output and is high when
ACSET is less than Vth,rise, and active low when ACSET is
above Vth,fall. Vth,rise and Vth,fall are given by Equation 12
and Equation 13:
FN6499.3
September 14, 2010
ISL6256, ISL6256A
⎛ R8
⎞
V th, rise = ⎜ ------- + 1⎟ ⋅ V ACSET
R
⎝ 9
⎠
(EQ. 12)
⎛ R8
⎞
V th, fall = ⎜ ------- + 1⎟ ⋅ V ACSET – I hys ⋅ R 8
⎝ R9
⎠
(EQ. 13)
where:
• Ihys is the ACSET input bias current hysteresis, and
• VACSET = 1.24V (min), 1.26V (typ) and 1.28V (max).
The hysteresis is IhysR8, where Ihys = 2.2µA (min),
3.4µA (typ) and 4.4µA (max).
DC Adapter Detection
Connect the DC adapter voltage like aircraft power through a
resistor divider to DCSET to detect when DC power is
available, as shown in Figure 3. DCPRN is an open-drain
output and is high when DCSET is less than Vth,rise, and
active low when DCSET is above Vth,fall. Vth,rise and Vth,fall
are given by Equations 14 and 15:
⎛ R 14
⎞
V th, rise = ⎜ ---------- + 1⎟ • V DCSET
R
⎝ 15
⎠
(EQ. 14)
⎛ R 14
⎞
V th, fall = ⎜ ---------- + 1⎟ • V DCSET – I hys R 14
⎝ R 15
⎠
(EQ. 15)
Where Ihys is the DCSET input bias current hysteresis and
VDCSET = 1.24V (min), 1.26V (typ) and 1.28V (max). The
hysteresis is Ihys R14, where Ihys = 2.2µA (min), 3.4µA (typ)
and 4.4µA (max).
Current Measurement
Use ICM to monitor the input current being sensed across
CSIP and CSIN. The output voltage range is 0V to 2.5V. The
voltage of ICM is proportional to the voltage drop across
CSIP and CSIN, and is given by Equation 16:
ICM = 19.9 • I INPUT • R 2
(EQ. 16)
where IINPUT is the DC current drawn from the AC adapter.
ICM has ±3% accuracy. It is recommended to have an RC
filter at the ICM output for minimizing the switching noise.
LDO Regulator
VDD provides a 5.0V supply voltage from the internal LDO
regulator from DCIN and can deliver up to 30mA of current.
The MOSFET drivers are powered by VDDP, which must be
connected to VDDP as shown in Figure 2. VDDP connects
to VDD through an external low pass filter. Bypass VDDP
and VDD with a 1µF capacitor.
ground. The ICM, ACPRN and DCPRN outputs continue to
function.
EN can be driven by a thermistor to allow automatic
shutdown of the ISL6256 when the battery pack is hot. Often
a NTC thermistor is included inside the battery pack to
measure its temperature. When connected to the charger,
the thermistor forms a voltage divider with a resistive pull-up
to the VREF. The threshold voltage of EN is 1.0V with 60mV
hysteresis. The thermistor can be selected to have a
resistance vs temperature characteristic that abruptly
decreases above a critical temperature. This arrangement
automatically shuts down the ISL6256 when the battery pack
is above a critical temperature.
Another method for inhibiting charging is to force CHLIM
below 85mV (typ).
Supply Isolation
If the voltage across the adapter sense resistor R2 is
typically greater than 8mV, the P-Channel MOSFET
controlled by SGATE is turned on reducing the power
dissipation. If the voltage across the adapter sense resistor
R2 is less than 3mV, SGATE turns off the P-Channel
MOSFET isolating the adapter from the system bus.
Battery Power Source Selection and Aircraft
Power Application
The battery voltage is monitored by CSON. If the battery
voltage measured on CSON is less than the adapter voltage
measured on DCIN, then the P-Channel MOSFET controlled
by BGATE turns off and the P-Channel MOSFET controlled by
SGATE is allowed to turn on when the adapter current is high
enough. If it is greater, then the P-Channel MOSFET controlled
by SGATE turns off and BGATE turns on the battery discharge
P-Channel MOSFET to minimize the power loss. Also, the
charging function is disabled. If designing for airplane power,
DCSET is tied to a resistor divider sensing the adapter voltage.
When a user is plugged into the 15V airplane supply and the
battery voltage is lower than 15V, the MOSFET driven by
BGATE (see Figure 3) is turned off which keeps the battery
from supplying the system bus. The comparator looking at
CSON and DCIN has 300mV of hysteresis to avoid chattering.
Only 2S and 3S are supported for DC aircraft power
applications. For 4S battery packs, set DCSET = 0.
Short Circuit Protection and 0V Battery Charging
Since the battery charger will regulate the charge current to
the limit set by CHLIM, it automatically has short circuit
protection and is able to provide the charge current to wake
up an extremely discharged battery.
Over-Temperature Protection
Shutdown
The ISL6256 features a low-power shutdown mode. Driving
EN low shuts down the ISL6256. In shutdown, the DC/DC
converter is disabled, and VCOMP and ICOMP are pulled to
16
If the die temp exceeds +150°C, it stops charging. Once the
die temp drops below +125°C, charging will start-up again.
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Overvoltage Protection
Application Information
ISL6256 has an Overvoltage Protection circuit that limits the
output voltage when the battery is removed or disconnected
by a pulse charging circuit. If CSON exceeds the output
voltage set point by more than VOVP an internal comparator
pulls VCOMP down and turns off both upper and lower FETs
of the buck as in Figure 17. The trip point for Overvoltage
Protection is always above the nominal output voltage and
can be calculated from Equation 17:
The following battery charger design refers to the typical
application circuit in Figure 2, where typical battery
configuration of 4S2P is used. This section describes how to
select the external components including the inductor, input
and output capacitors, switching MOSFETs, and current
sensing resistors.
V ADJ
V OVP = V OUT, NOM + N CELLS × ⎛ 42.2mV – 22.2mV × ----------------⎞
⎝
2.39V⎠
The inductor selection has trade-offs between cost, size,
crossover frequency and efficiency. For example, the lower
the inductance, the smaller the size, but ripple current is
higher. This also results in higher AC losses in the magnetic
core and the windings, which decrease the system
efficiency. On the other hand, the higher inductance results
in lower ripple current and smaller output filter capacitors,
but it has higher DCR (DC resistance of the inductor) loss,
lower saturation current and has slower transient response.
So, the practical inductor design is based on the inductor
ripple current being ±15% to ±20% of the maximum
operating DC current at maximum input voltage. Maximum
ripple is at 50% duty cycle or VBAT = VIN,MAX/2. The
required inductance can be calculated from Equation 18:
(EQ. 17)
For example, if the CELLS pin is connected to ground
(NCELLS = 3) and VADJ is floating (VADJ = 1.195V) then
VOUT,NOM = 12.6V and VOVP = 12.693V or
VOUT,NOM + 93mV.
There is a delay of approximately 400ns between VOUT
exceeding the OVP trip point and pulling VCOMP, LGATE
and UGATE low.
VOUT
VCOMP
WHEN VOUT EXCEEDS
THE OVP THRESHOLD
VCOMP IS PULLED LOW
ICOMP
AND FETS TURN OFF
Inductor Selection
V IN, MAX
L = --------------------------------------------4 ⋅ f SW ⋅ I RIPPLE
(EQ. 18)
Where VIN,MAX and fSW are the maximum input voltage,
and switching frequency, respectively.
The inductor ripple current ΔI is found from Equation 19:
BATTERY
REMOVAL
CURRENT FLOWS IN THE
LOWER FET BODY DIODE
UNTIL INDUCTOR CURRENT
REACHES ZERO
PHASE
FIGURE 17. OVERVOLTAGE PROTECTION IN ISL6256
During normal operation with cells installed, the CSON pin
voltage will be the cell stack voltage. When EN is low and the
cells are removed, this voltage may drop below 100mV. Due
to non-linearities in the OVP comparator at this low input
level, the VCOMP pin may be held low even after EN is
commanded high. If regulation is required in the absence of
cells then a series resistor and diode need to be installed
which inject current into the CSON pin from the VDD pin.
See R23 and D3 in Figure 3. This will maintain the CSON pin
voltage well within its linear range in the absence of cells,
and will be effectively out of the circuit when the diode is
reversed biased by the cell stack. Resistor values from 10k
to 100k have been found to be effective.
17
I RIPPLE = 0.3 ⋅ I L, MAX
(EQ. 19)
where the maximum peak-to-peak ripple current is 30% of
the maximum charge current is used.
For VIN,MAX = 19V, VBAT = 16.8V, IBAT,MAX = 2.6A, and
fs = 300kHz, the calculated inductance is 8.3µH. Choosing
the closest standard value gives L = 10µH. Ferrite cores are
often the best choice since they are optimized at 300kHz to
600kHz operation with low core loss. The core must be large
enough not to saturate at the peak inductor current IPeak in
Equation 20:
1
I PEAK = I L, MAX + --- ⋅ I RIPPLE
2
(EQ. 20)
Inductor saturation can lead to cascade failures due to very
high currents. Conservative design limits the peak and RMS
current in the inductor to less than 90% of the rated
saturation current.
Crossover frequency is heavily dependent on the inductor
value. fCO should be less than 20% of the switching
frequency and a conservative design has fCO less than 10%
of the switching frequency. The highest fCO is in voltage
FN6499.3
September 14, 2010
ISL6256, ISL6256A
control mode with the battery removed and may be
calculated (approximately) from Equation 21:
5 ⋅ 11 ⋅ R SENSE
f CO = ------------------------------------------2π ⋅ L
(EQ. 21)
Output Capacitor Selection
The output capacitor in parallel with the battery is used to
absorb the high frequency switching ripple current and
smooth the output voltage. The RMS value of the output
ripple current IRMS is given by Equation 22:
V IN, MAX
I RMS = --------------------------------- ⋅ D ⋅ ( 1 – D )
12 ⋅ L ⋅ f SW
(EQ. 22)
where the duty cycle D is the ratio of the output voltage
(battery voltage) over the input voltage for continuous
conduction mode, which is typical operation for the battery
charger. During the battery charge period, the output voltage
varies from its initial battery voltage to the rated battery
voltage. So, the duty cycle change can be in the range of
between 0.53 and 0.88 for the minimum battery voltage of
10V (2.5V/Cell) and the maximum battery voltage of 16.8V.
The maximum RMS value of the output ripple current occurs
at the duty cycle of 0.5 and is expressed as Equation 23:
V IN, MAX
I RMS = ------------------------------------------4 ⋅ 12 ⋅ L ⋅ F SW
(EQ. 23)
For VIN,MAX = 19V, VBAT = 16.8V, L = 10µH, and
fs = 300kHz, the maximum RMS current is 0.19A. A typical
10F ceramic capacitor is a good choice to absorb this
current and also has very small size. Organic polymer
capacitors have high capacitance with small size and have a
significant equivalent series resistance (ESR). Although
ESR adds to ripple voltage, it also creates a high frequency
zero that helps the closed loop operation of the buck
regulator.
EMI considerations usually make it desirable to minimize
ripple current in the battery leads. Beads may be added in
series with the battery pack to increase the battery
impedance at 300kHz switching frequency. Switching ripple
current splits between the battery and the output capacitor
depending on the ESR of the output capacitor and battery
impedance. If the ESR of the output capacitor is 10mΩ and
battery impedance is raised to 2Ω with a bead, then only
0.5% of the ripple current will flow in the battery.
achieved by selecting a high side MOSFET that has the
conduction losses equal to the switching losses. Switching
losses in the low-side FET are very small. The choice of
low-side FET is a trade-off between conduction losses
(rDS(ON)) and cost. A good rule of thumb for the rDS(ON) of
the low-side FET is 2x the rDS(ON) of the high-side FET.
The LGATE gate driver can drive sufficient gate current to
switch most MOSFETs efficiently. However, some FETs may
exhibit cross conduction (or shoot through) due to current
injected into the drain-to-source parasitic capacitor (Cgd) by
the high dV/dt rising edge at the phase node when the
high-side MOSFET turns on. Although LGATE sink current
(1.8A typical) is more than enough to switch the FET off
quickly, voltage drops across parasitic impedances between
LGATE and the MOSFET can allow the gate to rise during
the fast rising edge of voltage on the drain. MOSFETs with
low threshold voltage (<1.5V) and low ratio of Cgs/Cgd (<5)
and high gate resistance (>4Ω) may be turned on for a few
ns by the high dV/dt (rising edge) on their drain. This can be
avoided with higher threshold voltage and Cgs/Cgd ratio.
Another way to avoid cross conduction is slowing the turn-on
speed of the high-side MOSFET by connecting a resistor
between the BOOT pin and the boot strap capacitor.
For the high-side MOSFET, the worst-case conduction
losses occur at the minimum input voltage as shown in
Equation 24:
V OUT
2
P Q1, conduction = ---------------- ⋅ I BAT ⋅ r DS ( ON )
V IN
(EQ. 24)
The optimum efficiency occurs when the switching losses
equal the conduction losses. However, it is difficult to
calculate the switching losses in the high-side MOSFET
since it must allow for difficult-to-quantify factors that
influence the turn-on and turn-off times. These factors
include the MOSFET internal gate resistance, gate charge,
threshold voltage, stray inductance, pull-up and pull-down
resistance of the gate driver.
The following switching loss calculation (Equation 25)
provides a rough estimate.
P Q1, Switching =
(EQ. 25)
⎛ Q gd ⎞ 1
⎛ Q gd ⎞
1
-⎟ + --- V IN I LP f sw ⎜ ----------------⎟ + Q rr V IN f sw
--- V IN I LV f sw ⎜ -----------------------2
⎝ I g, source⎠ 2
⎝ I g, sin k⎠
where the following are the peak gate-drive source/sink
current of Q1, respectively:
MOSFET Selection
The Notebook battery charger synchronous buck converter
has the input voltage from the AC adapter output. The
maximum AC adapter output voltage does not exceed 25V.
Therefore, 30V logic MOSFET should be used.
The high side MOSFET must be able to dissipate the
conduction losses plus the switching losses. For the battery
charger application, the input voltage of the synchronous
buck converter is equal to the AC adapter output voltage,
which is relatively constant. The maximum efficiency is
18
• Qgd: drain-to-gate charge,
• Qrr: total reverse recovery charge of the body-diode in
low-side MOSFET,
• ILV: inductor valley current,
• ILP: Inductor peak current,
• Ig,sink
FN6499.3
September 14, 2010
ISL6256, ISL6256A
• Ig,source
Low switching loss requires low drain-to-gate charge Qgd.
Generally, the lower the drain-to-gate charge, the higher the
ON-resistance. Therefore, there is a trade-off between the
ON-resistance and drain-to-gate charge. Good MOSFET
selection is based on the figure of Merit (FOM), which is a
product of the total gate charge and ON-resistance. Usually,
the smaller the value of FOM, the higher the efficiency for
the same application.
For the low-side MOSFET, the worst-case power dissipation
occurs at minimum battery voltage and maximum input
voltage (Equation 26):
V OUT⎞
⎛
2
P Q2 = ⎜ 1 – ----------------⎟ ⋅ I BAT ⋅ r DS ( ON )
V IN ⎠
⎝
(EQ. 26)
Choose a low-side MOSFET that has the lowest possible
ON-resistance with a moderate-sized package like the SO-8
and is reasonably priced. The switching losses are not an
issue for the low-side MOSFET because it operates at
zero-voltage-switching.
Choose a Schottky diode in parallel with low-side MOSFET
Q2 with a forward voltage drop low enough to prevent the
low-side MOSFET Q2 body-diode from turning on during the
dead time. This also reduces the power loss in the high-side
MOSFET associated with the reverse recovery of the
low-side MOSFET Q2 body diode.
As a general rule, select a diode with DC current rating equal
to one-third of the load current. One option is to choose a
combined MOSFET with the Schottky diode in a single
package. The integrated packages may work better in
practice because there is less stray inductance due to a
short connection. This Schottky diode is optional and may be
removed if efficiency loss can be tolerated. In addition,
ensure that the required total gate drive current for the
selected MOSFETs should be less than 24mA. So, the total
gate charge for the high-side and low-side MOSFETs is
limited by Equation 27:
1 GATE
Q GATE ≤ ------------------f sw
(EQ. 27)
Where IGATE is the total gate drive current and should be
less than 24mA. Substituting IGATE = 24mA and fs = 300kHz
into Equation 27 yields that the total gate charge should be
less than 80nC. Therefore, the ISL6256 easily drives the
battery charge current up to 10A.
Snubber Design
ISL6256's buck regulator operates in discontinuous current
mode (DCM) when the load current is less than half the
peak-to-peak current in the inductor. After the low-side FET
turns off, the phase voltage rings due to the high impedance
with both FETs off. This can be seen in Figure 9. Adding a
snubber (resistor in series with a capacitor) from the phase
node to ground can greatly reduce the ringing. In some
19
situations a snubber can improve output ripple and
regulation.
The snubber capacitor should be approximately twice the
parasitic capacitance on the phase node. This can be
estimated by operating at very low load current (100mA) and
measuring the ringing frequency.
CSNUB and RSNUB can be calculated from Equations 28
and 29:
2
C SNUB = ------------------------------------2
( 2πF ring ) ⋅ L
R SNUB =
(EQ. 28)
2⋅L
-------------------C SNUB
(EQ. 29)
Input Capacitor Selection
The input capacitor absorbs the ripple current from the
synchronous buck converter, which is given by Equation 30:
V OUT ⋅ ( V IN – V OUT )
I RMS = I BAT ------------------------------------------------------------V IN
(EQ. 30)
This RMS ripple current must be smaller than the rated RMS
current in the capacitor datasheet. Non-tantalum chemistries
(ceramic, aluminum, or OSCON) are preferred due to their
resistance to power-up surge currents when the AC adapter
is plugged into the battery charger. For Notebook battery
charger applications, it is recommended that ceramic
capacitors or polymer capacitors from Sanyo be used due to
their small size and reasonable cost.
Table 2 shows the component lists for the typical application
circuit in Figure 2.
TABLE 2. COMPONENT LIST
PARTS
C1, C10
PART NUMBERS AND MANUFACTURER
10µF/25V ceramic capacitor, Taiyo Yuden
TMK325 MJ106MY X5R (3.2mmx2.5mmx1.9mm)
C2, C4, C8
0.1µF/50V ceramic capacitor
C3, C7, C9
1µF/10V ceramic capacitor, Taiyo Yuden
LMK212BJ105MG
C5
10nF ceramic capacitor
C6
6.8nF ceramic capacitor
C11
3300pF ceramic capacitor
D1
30V/3A Schottky diode, EC31QS03L (optional)
D2
100mA/30V Schottky Diode, Central Semiconductor
L
10µH/3.8A/26mΩ, Sumida, CDRH104R-100
Q1, Q2
30V/35mΩ, FDS6912A, Fairchild
Q3, Q4
-30V/30mΩ, SI4835BDY, Siliconix
Q5
Signal P-Channel MOSFET, NDS352AP
Q6
Signal N-Channel MOSFET, 2N7002
FN6499.3
September 14, 2010
ISL6256, ISL6256A
TABLE 2. COMPONENT LIST (Continued)
PARTS
PART NUMBERS AND MANUFACTURER
R1
40mΩ, ±1%, LRC-LR2512-01-R040-F, IRC
R2
20mΩ, ±1%, LRC-LR2010-01-R020-F, IRC
R3
18Ω, ±5%, (0805)
R4
2.2Ω, ±5%, (0805)
R5
100kΩ, ±5%, (0805)
R6
4.7k, ±5%, (0805)
R7
100Ω, ±5%, (0805)
R8, R11
130k, ±1%, (0805)
R9
10.2kΩ, ±1%, (0805)
R10
4.7Ω, ±5%, (0805)
R12
20kΩ, ±1%, (0805)
R13
1.87kΩ, ±1%, (0805)
resistance is the sum of the sense resistor and the internal
resistance of the MOSFETs, inductor and capacitor.
Figure 18 shows the small signal model of the pulse width
modulator (PWM), power stage, output filter and battery.
In most cases the Battery resistance is very small (<200mΩ)
resulting in a very low Q in the output filter. This results in a
frequency response from the input of the PWM to the
inductor current with a single pole at the frequency
calculated in Equation 31:
( R SENSE + r DS ( ON ) + R DCR + R BAT )
f POLE1 = ------------------------------------------------------------------------------------------------------2π ⋅ L
(EQ. 31)
The output capacitor creates a pole at a very high frequency
due to the small resistance in parallel with it. The frequency
of this pole is calculated in Equation 32:
1
f POLE2 = --------------------------------------2π ⋅ C o ⋅ R BAT
(EQ. 32)
LOOP COMPENSATION DESIGN
VDD
RAMP GEN
VRAMP = VDD/11
+
PWM GAIN FM
The Pulse Width Modulator in the ISL6256 converts voltage
at VCOMP to a duty cycle by comparing VCOMP to a
triangle wave (duty = VCOMP/VP-P RAMP). The low-pass
filter formed by L and CO convert the duty cycle to a DC
output voltage (Vo = VDCIN*duty). In ISL6256, the triangle
wave amplitude is proportional to VDCIN. Making the ramp
amplitude proportional to DCIN makes the gain from
VCOMP to the PHASE output a constant 11 and is
independent of DCIN. For small signal AC analysis, the
battery is modeled by it’s internal resistance. The total output
20
CO
PWM
INPUT
PWM
GAIN = 11
L
RSENSE
11
TRANSCONDUCTANCE AMPLIFIERS GM1, GM2 AND
GM3
The ISL6256 uses several transconductance amplifiers (also
known as gm amps). Most commercially available op amps
are voltage controlled voltage sources with gain expressed
as A = VOUT/VIN. Transconductance amps are voltage
controlled current sources with gain expressed as
gm = IOUT/VIN. Transconductance gain (gm) will appear in
some of the equations for poles and zeros in the
compensation.
L
DRIVERS
ISL6256 has three closed loop control modes. One controls
the output voltage when the battery is fully charged or
absent. A second controls the current into the battery when
charging and the third limits current drawn from the adapter.
The charge current and input current control loops are
compensated by a single capacitor on the ICOMP pin. The
voltage control loop is compensated by a network on the
VCOMP pin. Descriptions of these control loops and
guidelines for selecting compensation components will be
given in the following sections. Which loop controls the
output is determined by the minimum current buffer and the
minimum voltage buffer shown in Figure 1 on page 3. These
three loops will be described separately.
R FET_rDS(ON)
RL_DCR
CO
RBAT
PWM
INPUT
RESR
FIGURE 18. SMALL SIGNAL AC MODEL
Charge Current Control Loop
When the battery voltage is less than the fully charged
voltage, the voltage error amplifier goes to it’s maximum
output (limited to 1.2V above ICOMP) and the ICOMP
voltage controls the loop through the minimum voltage
buffer. Figure 19 shows the charge current control loop.
FN6499.3
September 14, 2010
ISL6256, ISL6256A
:
60
L
PHASE
R FET_rDS(ON)
+
0.25
-
20
C F2
CO
CHLIM
RS2
CSON
gm2
+
+
-
LOOP
20
RF2
CSOP
-
CICOMP
40
+
CA2
ICOMP
RL_DCR
GAIN (dB)
Σ
COMPENSATOR
MODULATOR
fZERO
11
RBAT
0
-20
fPOLE1
fFILTER
RESR
-40
FIGURE 19. CHARGE CURRENT LIMIT LOOP
The compensation capacitor (CICOMP) gives the error
amplifier (GMI) a pole at a very low frequency (<<1Hz) and a
a zero at fZ1. fZ1 is created by the 0.25*CA2 output added to
ICOMP. The frequency of can be calculated from Equation 33.
4 ⋅ gm2
f ZERO = --------------------------------------( 2π ⋅ C ICOMP )
gm2 = 50μA ⁄ V
(EQ. 33)
Placing this zero at a frequency equal to the pole calculated
in Equation 31 will result in maximum gain at low frequencies
and phase margin near 90°. If the zero is at a higher
frequency (smaller CICOMP), the DC gain will be higher but
the phase margin will be lower. Use a capacitor on ICOMP
that is equal to or greater than the value calculated in
Equation 34:
4 ⋅ ( 50μA ⁄ V )
C ICOMP = -----------------------------------------------------------------------------------------( R S2 + r DS ( ON ) + R DCR + R BAT )
(EQ. 34)
A filter should be added between RS2 and CSOP and CSON
to reduce switching noise. The filter roll off frequency should
be between the crossover frequency and the switching
frequency (~100kHz). RF2 should be small (<10Ω) to
minimize offsets due to leakage current into CSOP. The filter
cut off frequency is calculated using Equation 35:
1
f FILTER = ------------------------------------------( 2π ⋅ C F2 ⋅ R F2 )
(EQ. 35)
fPOLE2
-60
0.01k
0.1k
1k
10k
FREQUENCY (Hz)
Adapter Current Limit Control Loop
If the combined battery charge current and system load
current draws current that equals the adapter current limit
set by the ACLIM pin, ISL6256 will reduce the current to the
battery and/or reduce the output voltage to hold the adapter
current at the limit. Above the adapter current limit, the
minimum current buffer equals the output of gm3 and
ICOMP controls the charger output. Figure 21 shows the
adapter current limit control loop.
.
DCIN
(EQ. 36)
11 ⋅ R S2
f CO = A DC ⋅ f POLE1 = ---------------------2π ⋅ L
(EQ. 37)
The Bode plot of the loop gain, the compensator gain and
the power stage gain is shown in Figure 20.
21
L
PHASE
RS1
11
R FET_rDS(ON)
RF1
RL_DCR
CF1
+
0.25
-
Σ
20
CF2
-
RF2
CSOP
+
CA2
CSIN
+
11 ⋅ R S2
A DC = ---------------------------------------------------------------------------------------------------------( R S2 + r DS ( ON ) + R DCR + R BATTERY )
1M
FIGURE 20. CHARGE CURRENT LOOP BODE PLOTS
CSIP
The crossover frequency is determined by the DC gain of the
modulator and output filter and the pole in Equation 31. The
DC gain is calculated in Equation 36 and the crossover
frequency is calculated with Equation 37.
100k
R
CSON
20
CO
CA1
RESR
ICOMP
CICOMP
gm3
+
ACLIM
+
-
FIGURE 21. ADAPTER CURRENT LIMIT LOOP
The loop response equations, bode plots and the selection
of CICOMP are the same as the charge current control loop
with loop gain reduced by the duty cycle and the ratio of
RS1/RS2. In other words, if RS1= RS2 and the duty cycle
D = 50%, the loop gain will be 6dB lower than the loop gain
in Figure 21. This gives lower crossover frequency and
higher phase margin in this mode. If RS1/RS2 = 2 and the
duty cycle is 50% then the adapter current loop gain will be
identical to the gain in Figure 21.
FN6499.3
September 14, 2010
Voltage Control Loop
When the battery is charged to the voltage set by CELLS and
VADJ the voltage error amplifier (gm1) takes control of the
output (assuming that the adapter current is below the limit set
by ACLIM). The voltage error amplifier (gm1) discharges the
cap on VCOMP to limit the output voltage. The current to the
battery decreases as the cells charge to the fixed voltage and
the voltage across the internal battery resistance decreases.
As battery current decreases the 2 current error amplifiers
(gm2 and gm3) output their maximum current and charge the
capacitor on ICOMP to its maximum voltage (limited to 1.2V
above VCOMP). With high voltage on ICOMP, the minimum
voltage buffer output equals the voltage on VCOMP.
L
11
R FET_rDS(ON)
CA2
+
0.25
-
S
Σ
RL_DCR
RF2
CSOP
+
20
C F2
-
RS2
CSON
R3
VCOMP
gm1
+
CO
R4
C VCOMP
RVCOMP
2.1V
RBAT
RESR
+
-
FIGURE 22. VOLTAGE CONTROL LOOP
Output LC Filter Transfer Functions
The gain from the phase node to the system output and
battery depend entirely on external components. Typical
output LC filter response is shown in Figure 23. Transfer
function ALC(s) is shown in Equation 38:
s ⎞
⎛ 1 – --------------⎝
ω ESR⎠
A LC = ----------------------------------------------------------⎛ s2
⎞
s
⎜ ------------ + ------------------------- + 1⎟
⎝ ω DP ( ω LC ⋅ Q )
⎠
1
ω ESR = --------------------------------( R ESR ⋅ C o )
20
10
0
-10
-20
-30
-40
-50
-60
NO BATTERY
RBATTERY = 200mΩ
RBATTERY = 50mΩ
-20
-40
-60
-80
-100
-120
-140
-160
100 200
500 1k
2k
5k
10k 20k
50k 100k 200k 500k
FREQUENCY
FIGURE 23. FREQUENCY RESPONSE OF THE LC OUTPUT
FILTER
The resistance RO is a combination of MOSFET rDS(ON),
inductor DCR, RSENSE and the internal resistance of the
battery (normally between 50mΩ and 200mΩ). The worst case
for voltage mode control is when the battery is absent. This
results in the highest Q of the LC filter and the lowest phase
margin.
The voltage control loop is shown in Figure 22.
PHASE
PHASE (°)
A filter should be added between RS1 and CSIP and CSIN to
reduce switching noise. The filter roll off frequency should be
between the crossover frequency and the switching
frequency (~100kHz).
GAIN (dB)
ISL6256, ISL6256A
The compensation network consists of the voltage error
amplifier gm1 and the compensation network RVCOMP,
CVCOMP which give the loop very high DC gain, a very low
frequency pole and a zero at fZERO1. Inductor current
information is added to the feedback to create a second zero
fZERO2. The low pass filter RF2, CF2 between RSENSE and
ISL6256 add a pole at fFILTER. R3 and R4 are internal divider
resistors that set the DC output voltage. For a 3-cell battery,
R3 = 320kΩ and R4 = 64kΩ. Equations 39, 40, 41, 42, 43 and
44 relate the compensation network’s poles, zeros and gain to
the components in Figure 22. Figure 2424 shows an
asymptotic bode plot of the DC/DC converter’s gain vs
frequency. It is strongly recommended that fZERO1 is
approximately 30% of fLC and fZERO2 is approximately 70%
of fLC.
(EQ. 38)
1
ω LC = -----------------------( L ⋅ Co )
22
LQ = R o ⋅ -----Co
FN6499.3
September 14, 2010
ISL6256, ISL6256A
PCB Layout Considerations
.
COMPENSATOR
MODULATOR
40
fLC
Power and Signal Layers Placement on the PCB
fPOLE1
LOOP
As a general rule, power layers should be close together,
either on the top or bottom of the board, with signal layers on
the opposite side of the board. As an example, layer
arrangement on a 4-layer board is shown below:
GAIN (dB)
20
0
1. Top Layer: signal lines, or half board for signal lines and
the other half board for power lines
fFILTER
-20
2. Signal Ground
3. Power Layers: Power Ground
-40
fZERO1
4. Bottom Layer: Power MOSFET, Inductors and other
Power traces
fZERO2
fESR
-60
0.1k
1k
10k
100k
1M
FREQUENCY (Hz)
FIGURE 24. ASYMPTOTIC BODE PLOT OF THE VOLTAGE
CONTROL LOOP GAIN
COMPENSATION BREAK FREQUENCY EQUATIONS
1
f ZERO1 = ----------------------------------------------------------------------( 2π ⋅ C VCOMP ⋅ R 1COMP )
(EQ. 39)
R VCOMP
⎛
⎞ ⎛ R 4 ⎞ gm1
f ZERO2 = ⎜ --------------------------------------------------------⎟ ⋅ ⎜ ---------------------⎟ ⋅ ⎛ ------------⎞
⎝ 2π ⋅ R SENSE ⋅ C OUT⎠ ⎝ R 4 + R 3⎠ ⎝ 5 ⎠
Separate the power voltage and current flowing path from
the control and logic level signal path. The controller IC will
stay on the signal layer, which is isolated by the signal
ground to the power signal traces.
Component Placement
The power MOSFET should be close to the IC so that the
gate drive signal, the LGATE, UGATE, PHASE, and BOOT,
traces can be short.
Place the components in such a way that the area under the
IC has less noise traces with high dv/dt and di/dt, such as
gate signals and phase node signals.
(EQ. 40)
Signal Ground and Power Ground Connection
1
f LC = ------------------------------( 2π L ⋅ C o )
(EQ. 41)
1
f FILTER = ------------------------------------------( 2π ⋅ R F2 ⋅ C F2 )
(EQ. 42)
1
f POLE1 = ---------------------------------------------------( 2π ⋅ R SENSE ⋅ C o )
(EQ. 43)
GND and VDD Pin
(EQ. 44)
At least one high quality ceramic decoupling cap should be
used to cross these two pins. The decoupling cap can be put
close to the IC.
1
f ESR = -------------------------------------------( 2π ⋅ C o ⋅ R ESR )
At minimum, a reasonably large area of copper, which will
shield other noise couplings through the IC, should be used
as signal ground beneath the IC. The best tie-point between
the signal ground and the power ground is at the negative
side of the output capacitor on each side, where there is little
noise; a noisy trace beneath the IC is not recommended.
LGATE Pin
TABLE 3.
CELLS
R3
R4
2
288kΩ
48kΩ
3
320kΩ
64kΩ
4
336kΩ
96kΩ
This is the gate drive signal for the bottom MOSFET of the
buck converter. The signal going through this trace has both
high dv/dt and high di/dt, and the peak charging and
discharging current is very high. These two traces should be
short, wide, and away from other traces. There should be no
other traces in parallel with these traces on any layer.
Choose RVCOMP equal or lower than the value calculated
from Equation 45.
⎛ R 3 + R 4⎞
5
R VCOMP = ( 0.7 ⋅ F LC ) ⋅ ( 2π ⋅ C o ⋅ R SENSE ) ⋅ ⎛ ------------⎞ ⋅ ⎜ ---------------------⎟
⎝ gm1⎠ ⎝ R
4 ⎠
(EQ. 45)
PGND Pin
PGND pin should be laid out to the negative side of the
relevant output cap with separate traces.The negative side
of the output capacitor must be close to the source node of
the bottom MOSFET. This trace is the return path of LGATE.
Next, choose CVCOMP equal or higher than the value
calculated from Equation 46.
PHASE Pin
1
C VCOMP = --------------------------------------------------------------------------( 0.3 ⋅ F LC ) ⋅ ( 2π ⋅ R VCOMP )
This trace should be short, and positioned away from other
weak signal traces. This node has a very high dv/dt with a
23
(EQ. 46)
FN6499.3
September 14, 2010
ISL6256, ISL6256A
voltage swing from the input voltage to ground. No trace should
be in parallel with it. This trace is also the return path for
UGATE. Connect this pin to the high-side MOSFET source.
UGATE Pin
This pin has a square shape waveform with high dv/dt. It
provides the gate drive current to charge and discharge the
top MOSFET with high di/dt. This trace should be wide,
short, and away from other traces similar to the LGATE.
This pin stays high at enable mode and low at idle mode and
is relatively robust. Enable signals should refer to the signal
ground.
DCIN Pin
This pin connects to AC adapter output voltage, and should
be less noise sensitive.
Copper Size for the Phase Node
BOOT Pin
This pin’s di/dt is as high as the UGATE; therefore, this trace
should be as short as possible.
CSOP, CSON Pins
Accurate charge current and adapter current sensing is
critical for good performance. The current sense resistor
connects to the CSON and the CSOP pins through a low
pass filter with the filter cap very near the IC (see Figure 2).
Traces from the sense resistor should start at the pads of the
sense resistor and should be routed close together,
throughout the low pass filter and to the CSON and CSON
pins (see Figure 25). The CSON pin is also used as the
battery voltage feedback. The traces should be routed away
from the high dv/dt and di/dt pins like PHASE, BOOT pins. In
general, the current sense resistor should be close to the IC.
These guidelines should also be followed for the adapter
current sense resistor and CSIP and CSIN. Other layout
arrangements should be adjusted accordingly.
HIGH
CURRENT
TRACE
EN Pin
SENSE
RESISTOR
RESISTER
The capacitance of PHASE should be kept very low to
minimize ringing. It would be best to limit the size of the
PHASE node copper in strict accordance with the current
and thermal management of the application.
Identify the Power and Signal Ground
The input and output capacitors of the converters, the source
terminal of the bottom switching MOSFET PGND should
connect to the power ground. The other components should
connect to signal ground. Signal and power ground are tied
together at one point.
Clamping Capacitor for Switching MOSFET
It is recommended that ceramic caps be used closely
connected to the drain of the high-side MOSFET, and the
source of the low-side MOSFET. This capacitor reduces the
noise and the power loss of the MOSFET.
HIGH
CURRENT
TRACE
KELVIN CONNECTION TRACES
TO THE LOW PASS FILTER
AND
CSOP AND CSON
FIGURE 25. CURRENT SENSE RESISTOR LAYOUT
24
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Package Outline Drawing
L28.5x5
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 2, 10/07
4X 3.0
5.00
24X 0.50
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
28
22
1
5.00
21
3 .10 ± 0 . 15
15
(4X)
7
0.15
8
14
TOP VIEW
0.10 M C A B
- 0.07
4 28X 0.25 + 0.05
28X 0.55 ± 0.10
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 90 ± 0.1
C
BASE PLANE
SEATING PLANE
0.08 C
( 4. 65 TYP )
( 24X 0 . 50)
(
SIDE VIEW
3. 10)
(28X 0 . 25 )
C
0 . 2 REF
5
0 . 00 MIN.
0 . 05 MAX.
( 28X 0 . 75)
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
25
FN6499.3
September 14, 2010
ISL6256, ISL6256A
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
M28.15
N
INDEX
AREA
H
0.25(0.010) M
28 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
B M
E
1
2
INCHES
GAUGE
PLANE
-B-
SYMBOL
3
L
0.25
0.010
SEATING PLANE
-A-
A
D
h x 45°
-C-
α
e
A2
A1
B
C
0.10(0.004)
0.17(0.007) M
C A M
B S
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2
of Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
MIN
MAX
MILLIMETERS
MIN
MAX
NOTES
A
0.053
0.069
1.35
1.75
-
A1
0.004
0.010
0.10
0.25
-
A2
-
0.061
-
1.54
-
B
0.008
0.012
0.20
0.30
9
C
0.007
0.010
0.18
0.25
-
D
0.386
0.394
9.81
10.00
3
E
0.150
0.157
3.81
3.98
4
e
0.025 BSC
0.635 BSC
-
H
0.228
0.244
5.80
6.19
-
h
0.0099
0.0196
0.26
0.49
5
L
0.016
0.050
0.41
1.27
6
N
α
28
0°
28
8°
0°
7
8°
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
Rev. 1 6/04
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch)
per side.
5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of “B”
dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
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26
FN6499.3
September 14, 2010
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