Multiphase PWM Regulator for IMVP-6.5™ Mobile CPUs and GPUs ISL62882, ISL62882B Features The ISL62882 is a multiphase PWM buck regulator for miroprocessor or graphics processor core power supply. The multiphase buck converter uses interleaved phases to reduce the total output voltage ripple with each phase carrying a portion of the total load current, providing better system performance, superior thermal management, lower component cost, reduced power dissipation, and smaller implementation area. The ISL62882 uses two integrated gate drivers to provide a complete solution. The PWM modulator is based on Intersil's Robust Ripple Regulator (R3) technology™. Compared with traditional modulators, the R3™ modulator commands variable switching frequency during load transients, achieving faster transient response. With the same modulator, the switching frequency is reduced at light load, increasing the regulator efficiency. • Programmable 1- or 2-Phase CPU Mode Operation or 1-Phase GPU Mode Operation • Precision Multiphase Core Voltage Regulation - 0.5% System Accuracy Over-Temperature - Enhanced Load Line Accuracy The ISL62882 can be configured as CPU or graphics Vcore controller and is fully compliant with IMVP-6.5™ specifications. It responds to PSI# and DPRSLPVR signals by adding or dropping Phase 2, adjusting overcurrent protection threshold accordingly, and entering/exiting diode emulation mode. It reports the regulator output current through the IMON pin. It senses the current by using either discrete resistor or inductor DCR whose variation over temperature can be thermally compensated by a single NTC thermistor. It uses differential remote voltage sensing to accurately regulate the processor die voltage. The unique split LGATE function further increases light load efficiency. The adaptive body diode conduction time reduction function minimizes the body diode conduction loss in diode emulation mode. User-selectable overshoot reduction function offers an option to aggressively reduce the output capacitors as well as the option to disable it for users concerned about increased system thermal stress. The ISL62882 offers the FB2 function to optimize 1-phase performance. The ISL62882B has the same functions as the ISL62882, but comes in a different package. • Microprocessor Voltage Identification Input - 7-Bit VID Input, 0V to 1.500V in 12.5mV Steps - Supports VID Changes On-The-Fly • Supports Multiple Current Sensing Methods - Lossless Inductor DCR Current Sensing - Precision Resistor Current Sensing • Supports PSI# and DPRSLPVR modes • Superior Noise Immunity and Transient Response • Current Monitor and Thermal Monitor • Differential Remote Voltage Sensing • High Efficiency Across Entire Load Range • Programmable 1- or 2-Phase Operation • Two Integrated Gate Drivers • Excellent Dynamic Current Balance Between Phases • Split LGATE1 Drivers Increases Light Load Efficiency • FB2 Function Optimizes 1-Phase Mode Performance • Adaptive Body Diode Conduction Time Reduction • User-selectable Overshoot Reduction Function • Small Footprint 40 Ld 5x5 or 48 Ld 6x6 TQFN Packages • Pb-Free (RoHS Compliant) Applications • Notebook Core Voltage Regulator • Notebook GPU Voltage Regulator June 21, 2011 FN6890.4 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2009-2011. All Rights Reserved Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries. All other trademarks mentioned are the property of their respective owners. ISL62882, ISL62882B Ordering Information PART NUMBER (Notes 1, 2, 3) PART MARKING TEMP. RANGE (°C) PACKAGE (Pb-Free) PKG. DWG. # ISL62882IRTZ 62882 IRTZ -40 to +100 40 Ld 5x5 TQFN L40.5x5 ISL62882HRTZ 62882 HRTZ -10 to +100 40 Ld 5x5 TQFN L40.5x5 ISL62882BHRTZ 62882 BHRTZ -10 to +100 48 Ld 6x6 TQFN L48.6x6 NOTES: 1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL62882, ISL62882B. For more information on MSL please see techbrief TB363. Pin Configurations 30 BOOT2 PSI# 2 29 UGATE2 RBIAS 3 28 PHASE2 VR_TT# 4 27 VSSP2 NTC 5 VW GND PAD (BOTTOM) 6 COMP 7 FB NC VID0 VID1 VID2 VID4 VID3 VID5 VR_ON CLK_EN# PGOOD 2 NC 1 36 BOOT2 35 UGATE2 PSI# 3 34 PHASE2 RBIAS 4 NTC 6 25 VCCP GND 7 23 LGATE1a 33 VSSP2 VR_TT# 5 26 LGATE2 24 LGATE1b 8 DPRSLPVR NC VID1 VID3 VID2 VID4 VID5 VID6 VR_ON DPRSLPVR CLK_EN# VID0 48 47 46 45 44 43 42 41 40 39 38 37 40 39 38 37 36 35 34 33 32 31 PGOOD 1 VID6 ISL62882B (48 LD TQFN) TOP VIEW ISL62882 (40 LD TQFN) TOP VIEW 32 LGATE2 31 NC (BOTTOM) 30 VCCP 8 29 LGATE1b COMP 9 28 LGATE1a VW FB 10 2 NC 12 25 UGATE1 BOOT1 NC NC IMON VIN VDD ISUM+ RTN 13 14 15 16 17 18 19 20 21 22 23 24 ISUM- BOOT1 UGATE1 VIN IMON VDD ISUM+ RTN ISUM- VSEN ISEN1 11 12 13 14 15 16 17 18 19 20 26 PHASE1 VSEN 21 PHASE1 ISEN2 10 27 VSSP1 FB2 11 ISEN1 22 VSSP1 9 ISEN2 FB2 FN6890.4 June 21, 2011 ISL62882, ISL62882B Functional Pin Descriptions ISL62882 ISL62882B SYMBOL - 7 GND 1 2 PGOOD Power-Good open-drain output indicating when the regulator is able to supply regulated voltage. Pull-up externally with a 680Ω resistor to VCCP or 1.9kΩ to 3.3V. 2 3 PSI# Low load current indicator input. When asserted low, indicates a reduced load current condition. 3 4 RBIAS A resistor to GND sets internal current reference. Use 147kΩ or 47kΩ. The choice of Rbias value, together with the ISEN2 pin configuration and the external resistance from the COMP pin to GND, programs the controller to enable/disable the overshoot reduction function and to select the CPU/GPU mode. 4 5 VR_TT# Thermal overload output indicator. 5 6 NTC Thermistor input to VR_TT# circuit. 6 8 VW A resistor from this pin to COMP programs the switching frequency (8kΩ gives approximately 300kHz). 7 9 COMP 8 10 FB This pin is the inverting input of the error amplifier. 9 11 FB2 There is a switch between the FB2 pin and the FB pin. The switch is on in 2-phase mode and is off in 1-phase mode. The components connecting to FB2 are used to adjust the compensation in 1-phase mode to achieve optimum performance. 10 13 ISEN2 Individual current sensing for Phase 2. When ISEN2 is pulled to 5V VDD, the controller will disable Phase 2. 11 14 ISEN1 Individual current sensing for phase 1. 12 15 VSEN Remote core voltage sense input. Connect to microprocessor die. 13 16 RTN Remote voltage sensing return. Connect to ground at microprocessor die. 14, 15 17, 18 ISUM- and ISUM+ 16 19 VDD 5V bias power. 17 20 VIN Battery supply voltage, used for feed-forward. 18 22 IMON An analog output. IMON outputs a current proportional to the regulator output current. 19 24 BOOT1 Connect an MLCC capacitor across the BOOT1 and the PHASE1 pins. The boot capacitor is charged through an internal boot diode connected from the VCCP pin to the BOOT1 pin, each time the PHASE1 pin drops below VCCP minus the voltage dropped across the internal boot diode. 20 25 UGATE1 Output of the Phase-1 high-side MOSFET gate driver. Connect the UGATE1 pin to the gate of the Phase-1 high-side MOSFET. 21 26 PHASE1 Current return path for the Phase-1 high-side MOSFET gate driver. Connect the PHASE1 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of Phase-1. 22 27 VSSP1 Current return path for the Phase-1 low-side MOSFET gate driver. Connect the VSSP1 pin to the source of the Phase-1 low-side MOSFET through a low impedance path, preferably in parallel with the traces connecting the LGATE1a and the LGATE1b pins to the gates of the Phase-1 low-side MOSFETs. 23 28 LGATE1a Output of the Phase-1 low-side MOSFET gate driver that is always active. Connect the LGATE1a pin to the gate of the Phase-1 low-side MOSFET that is active all the time. 24 29 LGATE1b Another output of the Phase-1 low-side MOSFET gate driver. This gate driver will be pulled low when the DPRSLPVR pin logic is high. Connect the LGATE1b pin to the gate of the Phase-1 low-side MOSFET that is idle in deeper sleep mode. - - LGATE1 Output of the Phase-1 low-side MOSFET gate driver. Connect the LGATE1 pin to the gate of the Phase-1 low-side MOSFET. 3 DESCRIPTION Signal common of the IC. Unless otherwise stated, signals are referenced to the GND pin. This pin is the output of the error amplifier. Also, a resistor across this pin and GND adjusts the overcurrent threshold. Droop current sense input. FN6890.4 June 21, 2011 ISL62882, ISL62882B Functional Pin Descriptions (Continued) ISL62882 ISL62882B SYMBOL 25 30 VCCP Input voltage bias for the internal gate drivers. Connect +5V to the VCCP pin. Decouple with at least 1µF of an MLCC capacitor to VSSP1 and VSSP2 pins respectively. 26 32 LGATE2 Output of the Phase-2 low-side MOSFET gate driver. Connect the LGATE2 pin to the gate of the Phase-2 low-side MOSFET. 27 33 VSSP2 Current return path for the Phase-2 converter low-side MOSFET gate driver. Connect the VSSP2 pin to the source of the Phase-2 low-side MOSFET through a low impedance path, preferably in parallel with the trace connecting the LGATE2 pin to the gate of the Phase-2 low-side MOSFET. 28 34 PHASE2 Current return path for the Phase-2 high-side MOSFET gate driver. Connect the PHASE2 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of Phase-2. 29 35 UGATE2 Output of the Phase-2 high-side MOSFET gate driver. Connect the UGATE2 pin to the gate of the Phase-2 high-side MOSFET. 30 36 BOOT2 Connect an MLCC capacitor across the BOOT2 and the PHASE2 pins. The boot capacitor is charged through an internal boot diode connected from the VCCP pin to the BOOT2 pin, each time the PHASE2 pin drops below VCCP minus the voltage dropped across the internal boot diode. 31 thru 37 38 thru 44 VID0 thru VID6 38 45 VR_ON 39 46 DPRSLPVR 40 47 CLK_EN# - 48 NC pad pad BOTTOM 4 DESCRIPTION VID input with VID0 = LSB and VID6 = MSB. Voltage regulator enable input. A high level logic signal on this pin enables the regulator. Deeper sleep enable signal. A high level logic signal on this pin indicates that the microprocessor is in deeper sleep mode. Open drain output to enable system PLL clock. It goes low 13 switching cycles after Vcore is within 10% of Vboot. No connect. The bottom pad of ISL62882B is electrically connected to the GND pin inside the IC. FN6890.4 June 21, 2011 ISL62882, ISL62882B Block Diagram VIN VSEN ISEN2 ISEN1 PGOOD 6µA VR_ON PSI# VDD CLK_EN# MODE CONTROL 54µA 1.20V PGOOD & CLK_EN# LOGIC CURRENT BALANCE VR_TT# 1.24V NTC DPRSLPVR IBAL RBIAS PROTECTION BOOT2 FLT VID0 IBAL VID1 WOC VIN VDAC OC VIN VID2 VID3 MODULATOR DAC AND SOFTSTART CLOCK COMP VDAC COMP VID4 PWM CONTROL LOGIC DRIVER UGATE2 PHASE2 SHOOT THROUGH PROTECTION DRIVER VW LGATE2 VID5 VSSP2 VID6 BOOT1 DRIVER E/A FB IBAL COMP VIN VDAC MODULATOR VW IDROOP PWM CONTROL LOGIC Σ RTN PHASE1 SHOOT THROUGH PROTECTION VCCP DRIVER FB2 WOC IMON IMON 2.5X ISUM+ CURRENT SENSE LGATE1A COMP CURRENT COMPARATORS OC UGATE1 VSSP1 60µA NUMBER OF PHASES DRIVER LGATE1B ISUMGAIN SELECT 5 Σ ADJ. OCP THRESHOLD COMP GND FN6890.4 June 21, 2011 ISL62882, ISL62882B Absolute Maximum Ratings Thermal Information Supply Voltage, VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +28V Boot Voltage (BOOT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V Boot to Phase Voltage (BOOT-PHASE) . . . . . . . . . . . . . . . . -0.3V to +7V(DC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +9V(<10ns) Phase Voltage (PHASE) . . . . . . . . . . . . . . . . -7V (<20ns Pulse Width, 10µJ) UGATE Voltage (UGATE) . . . . . . . . . . . . . . . . . . . . PHASE-0.3V (DC) to BOOT . . . . . . . . . . . . . . . . . . . . . PHASE-5V (<20ns Pulse Width, 10µJ) to BOOT LGATE1a and 1b and LGATE2 Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V (DC) to VDD+0.3V LGATE1a and 1b . . . . . . . . . . . . . . . . . . . . . . -2.5V (<20ns Pulse Width, 2.5µJ) to VDD+0.3V LGATE1a and 1b . . . . . . . . . . . . . . . . . . . . . . . . -2.5V (<20ns Pulse Width, 5µJ) to VDD+0.3V All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VDD +0.3V) Open Drain Outputs, PGOOD, VR_TT#, CLK_EN# . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V Thermal Resistance (Typical) θJA (°C/W) θJC (°C/W) 40 Ld TQFN Package (Notes 4, 5) . . . . . . . 32 3 48 Ld TQFN Package (Notes 4, 5) . . . . . . . 29 2 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . .+150°C Storage Temperature Range. . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Recommended Operating Conditions Supply Voltage, VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5% Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +4.5V to 25V Ambient Temperature ISL62882HRTZ, ISL62882BHRTZ . . . . . . . . . . . . . . . . .-10°C to +100°C ISL62882IRTZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +100°C Junction Temperature ISL62882HRTZ, ISL62882BHRTZ . . . . . . . . . . . . . . . . .-10°C to +125°C ISL62882IRTZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTE: 4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. 5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications Operating Conditions: VDD = 5V, TA = -40°C to +100°C, fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS 4 4.6 mA INPUT POWER SUPPLY +5V Supply Current IVDD VR_ON = 3.3V VR_ON = 0V 1 µA Battery Supply Current IVIN VR_ON = 0V 1 µA VIN Input Resistance RVIN VR_ON = 3.3V 900 Power-On-Reset Threshold PORr VDD rising 4.35 PORf VDD falling 4.00 kΩ 4.5 4.15 V V SYSTEM AND REFERENCES System Accuracy HRTZ No load; closed loop, active mode range %Error (VCC_CORE) VID = 0.75V to 1.50V, -0.5 +0.5 % VID = 0.5V to 0.7375V -8 +8 mV VID = 0.3V to 0.4875V -15 +15 mV -0.8 +0.8 % -10 +10 mV +18 mV 1.1055 V IRTZ No load; closed loop, active mode range %Error (VCC_CORE) VID = 0.75V to 1.50V VID = 0.5V to 0.7375V VID = 0.3V to 0.4875V VBOOT -18 1.0945 1.100 Maximum Output Voltage VCC_CORE(max) VID = [0000000] 1.500 V Minimum Output Voltage VCC_CORE(min) VID = [1100000] 0.300 V RBIAS Voltage RBIAS = 147kΩ 6 1.45 1.47 1.49 V FN6890.4 June 21, 2011 ISL62882, ISL62882B Electrical Specifications Operating Conditions: VDD = 5V, TA = -40°C to +100°C, fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued) PARAMETER SYMBOL TEST CONDITIONS fSW(nom) Rfset = 7kΩ, 2-channel operation, VCOMP = 1V MIN (Note 6) TYP 285 300 MAX (Note 6) UNITS CHANNEL FREQUENCY Nominal Channel Frequency Adjustment Range 200 315 kHz 500 kHz +0.15 mV AMPLIFIERS Current-Sense Amplifier Input Offset IFB = 0A Error Amp DC Gain Av0 Error Amp Gain-Bandwidth Product GBW -0.15 CL = 20pF 90 dB 18 MHz ISEN Imbalance Voltage Maximum of ISENs - Minimum of ISENs 1 Input Bias Current 20 mV nA POWER-GOOD AND PROTECTION MONITORS PGOOD Low Voltage VOL IPGOOD = 4mA PGOOD Leakage Current IOH PGOOD = 3.3V PGOOD Delay tpgd CLK_ENABLE# LOW to PGOOD HIGH 0.26 0.4 1 μA 7.6 8.9 ms 1.5 Ω -1 6.3 V GATE DRIVER UGATE Pull-Up Resistance RUGPU 200mA Source Current 1.0 UGATE Source Current IUGSRC UGATE - PHASE = 2.5V 2.0 UGATE Sink Resistance RUGPD 250mA Sink Current 1.0 UGATE Sink Current IUGSNK UGATE - PHASE = 2.5V 2.0 LGATE1a and 1b Pull-Up Resistance RLGPU 250mA Source Current 2.0 LGATE1a and 1b Source Current ILGSRC LGATE1a and 1b - VSSP1 = 2.5V 1.0 LGATE1a and 1b Sink Resistance RLGPD 250mA Sink Current LGATE1a and 1b Sink Current ILGSNK LGATE1a and 1b - VSSP1 = 2.5V 2.0 UGATE1 to LGATE1a and 1b Deadtime tUGFLGR UGATE1 falling to LGATE1a and 1b rising, no load 23 LGATE1a and 1b to UGATE1 Deadtime tLGFUGR LGATE1a and 1b falling to UGATE1 rising, no load 28 1 A 1.5 Ω A 3 Ω 1.8 Ω A A ns ns Ω LGATE Pull-Up Resistance RLGPU 250mA Source Current 1.0 LGATE Source Current ILGSRC LGATE - VSSP = 2.5V 2.0 LGATE Sink Resistance RLGPD 250mA Sink Current 0.5 LGATE Sink Current ILGSNK LGATE - VSSP = 2.5V 4.0 A UGATE to LGATE Deadtime tUGFLGR UGATE falling to LGATE rising, no load 23 ns LGATE to UGATE Deadtime tLGFUGR LGATE falling to UGATE rising, no load 28 ns 1.5 A 0.9 Ω BOOTSTRAP DIODE Forward Voltage VF PVCC = 5V, IF = 2mA 0.58 V Reverse Leakage IR VR = 25V 0.2 µA PROTECTION Overvoltage Threshold OVH VSEN rising above setpoint for >1ms Severe Overvoltage Threshold OVHS OC Threshold Offset at Rcomp = Open Circuit Current Imbalance Threshold 150 195 240 VSEN rising for >2µs 1.525 1.55 1.575 V 2-phase configuration, ISUM- pin current 18.3 20.2 22.1 µA 1-phase configuration, ISUM- pin current 8.2 10.1 12.0 µA -355 -295 -235 mV One ISEN above another ISEN for >1.2ms Undervoltage Threshold UVf 7 VSEN falling below setpoint for >1.2ms 9 mV mV FN6890.4 June 21, 2011 ISL62882, ISL62882B Electrical Specifications Operating Conditions: VDD = 5V, TA = -40°C to +100°C, fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS 0.3 V LOGIC THRESHOLDS VR_ON Input Low VIL(1.0V) VR_ON Input High VIH(1.0V) ISL62882HRTZ 0.7 VIH(1.0V) ISL62882IRTZ 0.75 VID0-VID6, PSI#, and DPRSLPVR Input Low VIL(1.0V) VID0-VID6, PSI#, and DPRSLPVR Input High VIH(1.0V) V V 0.3 0.7 V V THERMAL MONITOR NTC Source Current NTC = 1.3V Over-Temperature Threshold V (NTC) falling VR_TT# Low Output Resistance RTT I = 20mA CLK_EN# Low Output Voltage VOL I = 4mA CLK_EN# Leakage Current IOH CLK_EN# = 3.3V 53 60 67 µA 1.18 1.2 1.22 V 6.5 9 Ω 0.26 0.4 V 1 µA CLK_EN# OUTPUT LEVELS -1 CURRENT MONITOR IMON Output Current IIMON IMON Clamp Voltage ISUM- pin current = 20µA 108 120 132 µA ISUM- pin current = 10µA 51 60 69 µA ISUM- pin current = 5µA 22 30 37.5 µA 1.1 1.15 VIMONCLAMP Current Sinking Capability 275 V µA INPUTS VR_ON Leakage Current IVR_ON VR_ON = 0V -1 VR_ON = 1V VIDx Leakage Current IVIDx PSI# Leakage Current IPSI# VIDx = 0V -1 VIDx = 1V PSI# = 0V IDPRSLPVR DPRSLPVR = 0V DPRSLPVR = 1V 1 µA µA 1 0 0.45 µA µA 0 0.45 -1 µA 1 0 0.45 -1 PSI# = 1V DPRSLPVR Leakage Current 0 0 µA µA 1 µA 6.5 mV/µs SLEW RATE Slew Rate (For VID Change) SR 5 NOTES: 6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. 8 FN6890.4 June 21, 2011 ISL62882, ISL62882B Gate Driver Timing Diagram PWM tLGFUGR tFU tRU 1V UGATE 1V LGATE tRL tFL tUGFLGR 9 FN6890.4 June 21, 2011 ISL62882, ISL62882B Simplified Application Circuits V+5 V+5 Vin VDD VCCP VIN Rbias RBIAS Rntc NTC o C PGOOD VR_TT# CLK_EN# VIDs PSI# DPRSLPVR VR_ON VW PGOOD VR_TT# CLK_EN# VID<0:6> PSI# DPRSLPVR VR_ON L2 Vo PHASE2 LGATE2 VSSP2 Rs2 ISEN2 Cs2 ISL62882 Rfset BOOT1 UGATE1 PHASE1 LGATE1b LGATE1a VSSP1 COMP FB2 FB Rdroop Vin BOOT2 UGATE2 L1 Rs1 ISEN1 Cs1 VSEN ISUM+ Rsum2 Rn Ris VCCSENSE VSSSENSE Cn RTN o C Rsum1 Cis Rimon Ri IMON IMON (Bottom Pad) VSS ISUM- FIGURE 1. TYPICAL CPU APPLICATION CIRCUIT USING DCR SENSING V+5 V+5 Vin VDD VCCP VIN Rbias RBIAS Rntc NTC o C PGOOD VR_TT# CLK_EN# VIDs PSI# DPRSLPVR VR_ON VW IMVP6_PWRGD VR_TT# CLK_ENABLE VID<0:6> # PSI# DPRSLPVR VR_ON UGATE2 COMP FB2 FB L2 Rsen2 L1 Rsen1 PHASE2 LGATE2 VSSP2 Vo Rs2 ISEN2 Cs2 ISL62882 Rfset Rdroop Vin BOOT2 BOOT1 UGATE1 PHASE1 LGATE1b LGATE1a VSSP1 Rs1 ISEN1 Cs1 VSEN ISUM+ Rsum2 Ris VCCSENSE VSSSENSE Cn RTN Cis Rimon Ri IMON IMON Rsum1 (Bottom Pad) VSS ISUM- FIGURE 2. TYPICAL CPU APPLICATION CIRCUIT USING RESISTOR SENSING 10 FN6890.4 June 21, 2011 ISL62882, ISL62882B V+5 V+5 Vin VDD VCCP VIN Rbias RBIAS Rntc NTC o C PGOOD VR_TT# CLK_EN# VID<0:6> PSI# DPRSLPVR VR_ON BOOT2 UGATE2 PGOOD VR_TT# CLK_EN# VIDs PSI# DPRSLPVR VR_ON VW PHASE2 LGATE2 VSSP2 ISEN2 ISL62882 Rfset COMP FB2 FB Rdroop Vin BOOT1 UGATE1 PHASE1 LGATE1b LGATE1a VSSP1 L Vo ISEN1 VSEN ISUM+ Rsum Rn Ris VCCSENSE VSSSENSE Cn RTN o C Cis Rimon Ri IMON IMON (Bottom Pad) VSS ISUM- FIGURE 3. TYPICAL GPU APPLICATION CIRCUIT USING DCR SENSING V+5 V+5 Vin VDD VCCP VIN Rbias RBIAS Rntc NTC o C IMVP6_PWRGD VR_TT# CLK_ENABLE VID<0:6> # PSI# DPRSLPVR VR_ON BOOT2 PGOOD VR_TT# CLK_EN# VIDs PSI# DPRSLPVR VR_ON VW PHASE2 LGATE2 VSSP2 ISEN2 ISL62882 Rfset COMP Rdroop UGATE2 FB2 FB Vin BOOT1 UGATE1 PHASE1 LGATE1b LGATE1a VSSP1 L Rsen Vo ISEN1 VSEN ISUM+ Rsum2 Ris VCCSENSE VSSSENSE Cn RTN Cis Rimon Ri IMON IMON (Bottom Pad) VSS ISUM- FIGURE 4. TYPICAL GPU APPLICATION CIRCUIT USING RESISTOR SENSING 11 FN6890.4 June 21, 2011 ISL62882, ISL62882B Theory of Operation VW Multiphase R3™ Modulator Master Clock Circuit Master COMP Clock Phase Vcrm Sequencer COMP VW Master Clock Vcrm Clock1 Clock2 Master Clock Crm gmVo Clock1 VW VW Slave Circuit 1 Clock1 S PWM1 Phase1 Q R L1 PWM1 Vo Clock2 IL1 Vcrs1 Co PWM2 VW gm Crs1 VW Slave Circuit 2 Phase2 S PWM2 Q R Clock2 L2 IL2 Vcrs2 Vcrs1 Vcrs2 FIGURE 7. R3™ MODULATOR OPERATION PRINCIPLES IN LOAD INSERTION RESPONSE gm Crs2 FIGURE 5. R3™ MODULATOR CIRCUIT VW Hysteretic Window Vcrm COMP The ISL62882 is a multiphase regulator implementing Intel® IMVP-6.5™ protocol. It can be programmed for 1- or 2-phase operation for microprocessor core applications. It uses Intersil patented R3™ (Robust Ripple Regulator™) modulator. The R3™ modulator combines the best features of fixed frequency PWM and hysteretic PWM while eliminating many of their shortcomings. Figure 5 conceptually shows the ISL62882 multiphase R3™ modulator circuit, and Figure 6 shows the operation principles. A current source flows from the VW pin to the COMP pin, creating a voltage window set by the resistor between the two pins. This voltage window is called VW window in the following discussion. Master Clock Clock1 PWM1 Clock2 PWM2 VW Vcrs2 Vcrs1 FIGURE 6. R3™ MODULATOR OPERATION PRINCIPLES IN STEADY STATE 12 Inside the IC, the modulator uses the master clock circuit to generate the clocks for the slave circuits. The modulator discharges the ripple capacitor Crm with a current source equal to gmVo, where gm is a gain factor. Crm voltage Vcrm is a sawtooth waveform traversing between the VW and COMP voltages. It resets to VW when it hits COMP, and generates a one-shot master clock signal. A phase sequencer distributes the master clock signal to the slave circuits. If the ISL62882 is in 2-phase mode, the master clock signal will be distributed to Phases 1 and 2, and the Clock1 and Clock2 signals will be 180° out-of-phase. If the ISL62882 is in 1-phase mode, the master clock signal will be distributed to Phases 1 only and be the Clock1 signal. Each slave circuit has its own ripple capacitor Crs, whose voltage mimics the inductor ripple current. A gm amplifier converts the inductor voltage into a current source to charge and discharge Crs. The slave circuit turns on its PWM pulse upon receiving the clock signal, and the current source charges Crs. When Crs voltage VCrs hits VW, the slave circuit turns off the PWM pulse, and the current source discharges Crs. FN6890.4 June 21, 2011 ISL62882, ISL62882B Since the ISL62882 works with Vcrs, which are large amplitude and noise-free synthesized signals, the ISL62882 achieves lower phase jitter than conventional hysteretic mode and fixed PWM mode controllers. Unlike conventional hysteretic mode converters, the ISL62882 has an error amplifier that allows the controller to maintain a 0.5% output voltage accuracy. Figure 7 shows the operation principles during load insertion response. The COMP voltage rises during load insertion, generating the master clock signal more quickly, so the PWM pulses turn on earlier, increasing the effective switching frequency, which allows for higher control loop bandwidth than conventional fixed frequency PWM controllers. The VW voltage rises as the COMP voltage rises, making the PWM pulses wider. During load release response, the COMP voltage falls. It takes the master clock circuit longer to generate the next master clock signal so the PWM pulse is held off until needed. The VW voltage falls as the VW voltage falls, reducing the current PWM pulse width. This kind of behavior gives the ISL62882 excellent response speed. The fact that both phases share the same VW window voltage also ensures excellent dynamic current balance between phases. Figure 9 shows the operation principle in diode emulation mode at light load. The load gets incrementally lighter in the three cases from top to bottom. The PWM on-time is determined by the VW window size, therefore is the same, making the inductor current triangle the same in the three cases. The ISL62882 clamps the ripple capacitor voltage Vcrs in DE mode to make it mimic the inductor current. It takes the COMP voltage longer to hit Vcrs, naturally stretching the switching period. The inductor current triangles move further apart from each other such that the inductor current average value is equal to the load current. The reduced switching frequency helps to increase light load efficiency. CCM/DCM BOUNDARY VW Vcrs iL VW LIGHT DCM Vcrs Diode Emulation and Period Stretching iL DEEP DCM PHASE VW Vcrs UGATE iL LGATE FIGURE 9. PERIOD STRETCHING IL FIGURE 8. DIODE EMULATION ISL62882 can operate in diode emulation (DE) mode to improve light load efficiency. In DE mode, the low-side MOSFET conducts when the current is flowing from source to drain and does not allow reverse current, emulating a diode. As Figure 8 shows, when LGATE is on, the low-side MOSFET carries current, creating negative voltage on the phase node due to the voltage drop across the ON-resistance. The ISL62882 monitors the current through monitoring the phase node voltage. It turns off LGATE when the phase node voltage reaches zero to prevent the inductor current from reversing the direction and creating unnecessary power loss. If the load current is light enough, as Figure 8 shows, the inductor current will reach and stay at zero before the next phase node pulse, and the regulator is in discontinuous conduction mode (DCM). If the load current is heavy enough, the inductor current will never reach 0A, and the regulator is in CCM although the controller is in DE mode. 13 Start-up Timing With the controller's VDD voltage above the POR threshold, the start-up sequence begins when VR_ON exceeds the 3.3V logic high threshold. Figure 10 shows the typical start-up timing when the ISL62882 is configured for CPU VR application. The ISL62882 uses digital soft-start to ramp-up DAC to the boot voltage of 1.1V at about 2.5mV/µs. Once the output voltage is within 10% of the boot voltage for 13 PWM cycles (43µs for frequency = 300kHz), CLK_EN# is pulled low and DAC slews at 5mV/µs to the voltage set by the VID pins. PGOOD is asserted high in approximately 7ms. Similar results occur if VR_ON is tied to VDD, with the soft-start sequence starting 120µs after VDD crosses the POR threshold. Figure 11 shows the typical start-up timing when the ISL62882 is configured for GPU VR application. The ISL62882 uses digital soft start to ramp up DAC to the voltage set by the VID pins. The slew rate is 5mV/µs when there is DPRSLPVR = 0, and is doubled when there is DPRSLPVR = 1. Once the output voltage is within 10% of the target voltage for 13 PWM cycles (43µs for frequency = 300kHz), CLK_EN# is pulled low. PGOOD is asserted high in approximately 7ms. Similar results occur if VR_ON is tied to VDD, with the soft-start sequence starting 120µs after VDD crosses the POR threshold. FN6890.4 June 21, 2011 ISL62882, ISL62882B TABLE 1. VID TABLE (Continued) VDD 5mV/µs VR_ON 2.5mV/µs 90% Vboot 800µs VID COMMAND VOLTAGE DAC 13 SWITCHING CYCLES CLK_EN# ~7ms PGOOD FIGURE 10. SOFT-START WAVEFORMS FOR CPU VR APPLICATION VDD VR_ON SLEW RATE 90% 120µs VID COMMAND VOLTAGE DAC 13 SWITCHING CYCLES CLK_EN# ~7ms PGOOD FIGURE 11. SOFT-START WAVEFORMS FOR GPU VR APPLICATION Voltage Regulation and Load Line Implementation After the start sequence, the ISL62882 regulates the output voltage to the value set by the VID inputs per Table 1. The ISL62882 will control the no-load output voltage to an accuracy of ±0.5% over the range of 0.75V to 1.5V. A differential amplifier allows voltage sensing for precise voltage regulation at the microprocessor die. TABLE 1. VID TABLE VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) 0 0 0 0 0 0 0 1.5000 0 0 0 0 0 0 1 1.4875 0 0 0 0 0 1 0 1.4750 0 0 0 0 0 1 1 1.4625 0 0 0 0 1 0 0 1.4500 0 0 0 0 1 0 1 1.4375 0 0 0 0 1 1 0 1.4250 0 0 0 0 1 1 1 1.4125 0 0 0 1 0 0 0 1.4000 0 0 0 1 0 0 1 1.3875 0 0 0 1 0 1 0 1.3750 0 0 0 1 0 1 1 1.3625 14 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) 0 0 0 1 1 0 0 1.3500 0 0 0 1 1 0 1 1.3375 0 0 0 1 1 1 0 1.3250 0 0 0 1 1 1 1 1.3125 0 0 1 0 0 0 0 1.3000 0 0 1 0 0 0 1 1.2875 0 0 1 0 0 1 0 1.2750 0 0 1 0 0 1 1 1.2625 0 0 1 0 1 0 0 1.2500 0 0 1 0 1 0 1 1.2375 0 0 1 0 1 1 0 1.2250 0 0 1 0 1 1 1 1.2125 0 0 1 1 0 0 0 1.2000 0 0 1 1 0 0 1 1.1875 0 0 1 1 0 1 0 1.1750 0 0 1 1 0 1 1 1.1625 0 0 1 1 1 0 0 1.1500 0 0 1 1 1 0 1 1.1375 0 0 1 1 1 1 0 1.1250 0 0 1 1 1 1 1 1.1125 0 1 0 0 0 0 0 1.1000 0 1 0 0 0 0 1 1.0875 0 1 0 0 0 1 0 1.0750 0 1 0 0 0 1 1 1.0625 0 1 0 0 1 0 0 1.0500 0 1 0 0 1 0 1 1.0375 0 1 0 0 1 1 0 1.0250 0 1 0 0 1 1 1 1.0125 0 1 0 1 0 0 0 1.0000 0 1 0 1 0 0 1 0.9875 0 1 0 1 0 1 0 0.9750 0 1 0 1 0 1 1 0.9625 0 1 0 1 1 0 0 0.9500 0 1 0 1 1 0 1 0.9375 0 1 0 1 1 1 0 0.9250 0 1 0 1 1 1 1 0.9125 0 1 1 0 0 0 0 0.9000 0 1 1 0 0 0 1 0.8875 0 1 1 0 0 1 0 0.8750 FN6890.4 June 21, 2011 ISL62882, ISL62882B TABLE 1. VID TABLE (Continued) TABLE 1. VID TABLE (Continued) VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) 0 1 1 0 0 1 1 0.8625 1 0 1 1 0 1 0 0.3750 0 1 1 0 1 0 0 0.8500 1 0 1 1 0 1 1 0.3625 0 1 1 0 1 0 1 0.8375 1 0 1 1 1 0 0 0.3500 0 1 1 0 1 1 0 0.8250 1 0 1 1 1 0 1 0.3375 0 1 1 0 1 1 1 0.8125 1 0 1 1 1 1 0 0.3250 0 1 1 1 0 0 0 0.8000 1 0 1 1 1 1 1 0.3125 0 1 1 1 0 0 1 0.7875 1 1 0 0 0 0 0 0.3000 0 1 1 1 0 1 0 0.7750 1 1 0 0 0 0 1 0.2875 0 1 1 1 0 1 1 0.7625 1 1 0 0 0 1 0 0.2750 0 1 1 1 1 0 0 0.7500 1 1 0 0 0 1 1 0.2625 0 1 1 1 1 0 1 0.7375 1 1 0 0 1 0 0 0.2500 0 1 1 1 1 1 0 0.7250 1 1 0 0 1 0 1 0.2375 0 1 1 1 1 1 1 0.7125 1 1 0 0 1 1 0 0.2250 1 0 0 0 0 0 0 0.7000 1 1 0 0 1 1 1 0.2125 1 0 0 0 0 0 1 0.6875 1 1 0 1 0 0 0 0.2000 1 0 0 0 0 1 0 0.6750 1 1 0 1 0 0 1 0.1875 1 0 0 0 0 1 1 0.6625 1 1 0 1 0 1 0 0.1750 1 0 0 0 1 0 0 0.6500 1 1 0 1 0 1 1 0.1625 1 0 0 0 1 0 1 0.6375 1 1 0 1 1 0 0 0.1500 1 0 0 0 1 1 0 0.6250 1 1 0 1 1 0 1 0.1375 1 0 0 0 1 1 1 0.6125 1 1 0 1 1 1 0 0.1250 1 0 0 1 0 0 0 0.6000 1 1 0 1 1 1 1 0.1125 1 0 0 1 0 0 1 0.5875 1 1 1 0 0 0 0 0.1000 1 0 0 1 0 1 0 0.5750 1 1 1 0 0 0 1 0.0875 1 0 0 1 0 1 1 0.5625 1 1 1 0 0 1 0 0.0750 1 0 0 1 1 0 0 0.5500 1 1 1 0 0 1 1 0.0625 1 0 0 1 1 0 1 0.5375 1 1 1 0 1 0 0 0.0500 1 0 0 1 1 1 0 0.5250 1 1 1 0 1 0 1 0.0375 1 0 0 1 1 1 1 0.5125 1 1 1 0 1 1 0 0.0250 1 0 1 0 0 0 0 0.5000 1 1 1 0 1 1 1 0.0125 1 0 1 0 0 0 1 0.4875 1 1 1 1 0 0 0 0.0000 1 0 1 0 0 1 0 0.4750 1 1 1 1 0 0 1 0.0000 1 0 1 0 0 1 1 0.4625 1 1 1 1 0 1 0 0.0000 1 0 1 0 1 0 0 0.4500 1 1 1 1 0 1 1 0.0000 1 0 1 0 1 0 1 0.4375 1 1 1 1 1 0 0 0.0000 1 0 1 0 1 1 0 0.4250 1 1 1 1 1 0 1 0.0000 1 0 1 0 1 1 1 0.4125 1 1 1 1 1 1 0 0.0000 1 0 1 1 0 0 0 0.4000 1 1 1 1 1 1 1 0.0000 1 0 1 1 0 0 1 0.3875 15 FN6890.4 June 21, 2011 ISL62882, ISL62882B VCC SENSE + V droop = V DAC + VSS SENSE (EQ. 3) Rdroop Rewriting Equation 3 and substitution of Equation 2 gives: VCCSENSE Vdroop FB VR LOCAL “CATCH” VO RESISTOR Idroop E/A COMP Σ VDAC DAC VIDs VID<0:6> VSSSENSE X1 VSS “CATCH” RESISTOR FIGURE 12. DIFFERENTIAL SENSING AND LOAD LINE IMPLEMENTATION Equation 4 is the exact equation required for load line implementation. Phase Current Balancing As the load current increases from zero, the output voltage will droop from the VID table value by an amount proportional to the load current to achieve the load line. The ISL62882 can sense the inductor current through the intrinsic DC Resistance (DCR) of the inductors as shown in Figure 1 or through resistors in series with the inductors as shown in Figure 2. In both methods, capacitor Cn voltage represents the inductor total currents. A droop amplifier converts Cn voltage into an internal current source with the gain set by resistor Ri. The current source is used for load line implementation, current monitor and overcurrent protection. Figure 12 shows the load line implementation. The ISL62882 drives a current source Idroop out of the FB pin, described by Equation 1. 2xV Cn I droop = ---------------Ri (EQ. 4) The VCCSENSE and VSSSENSE signals come from the processor die. The feedback will be open circuit in the absence of the processor. As Figure 12 shows, it is recommended to add a “catch” resistor to feed the VR local output voltage back to the compensator, and add another “catch” resistor to connect the VR local output ground to the RTN pin. These resistors, typically 10Ω~100Ω, will provide voltage feedback if the system is powered up without a processor installed. RTN INTERNAL TO IC VCC SENSE – VSS SENSE = V DAC – R droop × I droop (EQ. 1) When using inductor DCR current sensing, a single NTC element is used to compensate the positive temperature coefficient of the copper winding thus sustaining the load line accuracy with reduced cost. L2 RDCR2 RPCB2 PHASE2 RS ISEN2 CS INTERNAL TO IC VO IL2 L1 RDCR1 RPCB1 PHASE1 RS ISEN1 IL1 CS FIGURE 13. CURRENT BALANCING CIRCUIT The ISL62882 monitors individual phase average current by monitoring the ISEN1 and ISEN2 voltages. Figure 13 shows the current balancing circuit recommended for ISL62882. Each phase node voltage is averaged by a low-pass filter consisting of Rs and Cs, and presented to the corresponding ISEN pin. Rs should be routed to inductor phase-node pad in order to eliminate the effect of phase node parasitic PCB DCR. Equations 5 and 6 give the ISEN pin voltages: V ISEN1 = ( R dcr1 + R pcb1 ) × I L1 (EQ. 5) Idroop flows through resistor Rdroop and creates a voltage drop as shown in Equation 2. V ISEN2 = ( R dcr2 + R pcb2 ) × I L2 (EQ. 6) V droop = R droop × I droop where Rdcr1 and Rdcr2 are inductor DCR; Rpcb1 and Rpcb2 are parasitic PCB DCR between the inductor output side pad and the output voltage rail; and IL1 and IL2 are inductor average currents. (EQ. 2) Vdroop is the droop voltage required to implement load line. Changing Rdroop or scaling Idroop can both change the load line slope. Since Idroop also sets the overcurrent protection level, it is recommended to first scale Idroop based on OCP requirement, then select an appropriate Rdroop value to obtain the desired load line slope. Differential Sensing Figure 12 also shows the differential voltage sensing scheme. VCCSENSE and VSSSENSE are the remote voltage sensing signals from the processor die. A unity gain differential amplifier senses the VSSSENSE voltage and add it to the DAC output. The error amplifier regulates the inverting and the non-inverting input voltages to be equal as shown in Equation 3: 16 The ISL62882 will adjust the phase pulse-width relative to the other phase to make VISEN1 = VISEN2, thus to achieve IL1 = IL2, when there are Rdcr1 = Rdcr2 and Rpcb1 = Rpcb2. Using same components for L1 and L2 will provide a good match of Rdcr1 and Rdcr2. Board layout will determine Rpcb1 and Rpcb2. It is recommended to have symmetrical layout for the power delivery path between each inductor and the output voltage rail, such that Rpcb1 = Rpcb2 . FN6890.4 June 21, 2011 ISL62882, ISL62882B ISEN2 PHASE2 Rs Cs INTERNAL TO IC Rdcr2 L2 V2p REP RATE = 10kHz V2n Rpcb2 IL2 Rs VO Rs Rs ISEN1 Cs PHASE1 L1 Rdcr1 V1p Rpcb1 V1n IL1 FIGURE 14. DIFFERENTIAL-SENSING CURRENT BALANCING CIRCUIT REP RATE = 25kHz Sometimes, it is difficult to implement symmetrical layout. For the circuit Figure 13 shows, asymmetric layout causes different Rpcb1 and Rpcb2 thus current imbalance. Figure 14 shows a differential-sensing current balancing circuit recommended for ISL62882. The current sensing traces should be routed to the inductor pads so they only pick up the inductor DCR voltage. Each ISEN pin sees the average voltage of two sources: its own phase inductor phase-node pad, and the other phase inductor output side pad. Equations 7 and 8 give the ISEN pin voltages: V ISEN1 = V 1p + V 2n (EQ. 7) V ISEN2 = V 2p + V 1n (EQ. 8) REP RATE = 50kHz The ISL62882 will make VISEN1 = VISEN2. So there are: V 1p + V 2n = V 2p + V 1n (EQ. 9) Rewriting Equation 9 gives: V 1p – V 1n = V 2p – V 2n (EQ. 10) REP RATE = 100kHz Therefore: R dcr1 × I L1 = R dcr2 × I L2 (EQ. 11) Current balancing (IL1 = IL2) will be achieved when there is Rdcr1 = Rdcr2. Rpcb1 and Rpcb2 will not have any effect. Since the slave ripple capacitor voltages mimic the inductor currents, R3™ modulator can naturally achieve excellent current balancing during steady state and dynamic operations. Figure 15 shows current balancing performance of the ISL62882 evaluation board with load transient of 15A/50A at different rep rates. The inductor currents follow the load current dynamic change with the output capacitors supplying the difference. The inductor currents can track the load current well at a low rep rate, but cannot keep up when the rep rate gets into the hundred-kHz range, where it’s out of the control loop bandwidth. The controller achieves excellent current balancing in all cases. REP RATE = 200kHz FIGURE 15. ISL62882 EVALUATION BOARD CURRENT BALANCING DURING DYNAMIC OPERATION. Ch1: IL1, Ch2: IIoad, Ch3: IL2 17 FN6890.4 June 21, 2011 ISL62882, ISL62882B CCM Switching Frequency The Rfset resistor between the COMP and the VW pins sets the VW windows size, therefore sets the switching frequency. When the ISL62882 is in continuous conduction mode (CCM), the switching frequency is not absolutely constant due to the nature of the R3™ modulator. As explained in the “Multiphase R3™ Modulator” on page 12, the effective switching frequency will increase during load insertion and will decrease during load release to achieve fast response. On the other hand, the switching frequency is relatively constant at steady state. Variation is expected when the power stage condition, such as input voltage, output voltage, load, etc. changes. The variation is usually less than 15% and doesn’t have any significant effect on output voltage ripple magnitude. Equation 12 gives an estimate of the frequency-setting resistor Rfset value. 8kΩ Rfset gives approximately 300kHz switching frequency. Lower resistance gives higher switching frequency. R fset ( kΩ ) = ( Period ( μs ) – 0.29 ) × 2.65 (EQ. 12) Modes of Operation TABLE 2. ISL62882 CONFIGURATIONS Rbias (kΩ) ISEN2 CONFIGURATION Connected to the Power Stage 147 Tied to 5V 147 1-phase CPU VR 47 1-phase GPU VR OVERSHOOT REDUCTION FUNCTION 2-phase CPU VR Disabled 47 Enabled See Table 4 TABLE 3. ISL62882 MODES OF OPERATION CONFIG. OPERATIONAL MODE PSI# DPRSLPVR 0 0 1-phase CCM 0 1 1-phase DE 1 0 2-phase CCM 1 1 1-phase DE 1-phase CPU Configuration x 0 1-phase CCM 1 1-phase DE 1-phase GPU Configuration x 0 1-phase CCM 1 1-phase DE 2-phase CPU Configuration VOLTAGE SLEW RATE 5mV/µs Table 3 shows the ISL62882 operational modes, programmed by the logic status of the PSI# and DPRSLPVR pins. In 2-phase configuration, the ISL62882 enters 1-phase CCM for (PSI# = 0 and DPRSLPVR = 0). It drops phase 2 and reduces the overcurrent and the way-overcurrent protection levels to 1/2 of the initial values. The ISL62882 enters 1-phase DE mode when DPRSLPVR = 1 by dropping phase 2. In 1-phase configuration, the ISL62882 does not change the operational mode when the PSI# signal changes status. It enters 1-phase DE mode when DLPRSLPVR = 1. Dynamic Operation When the ISL62882 is configured for CPU VR application, it responds to VID changes by slewing to the new voltage at 5mV/µs slew rate. As the output approaches the VID command voltage, the dv/dt moderates to prevent overshoot. Geyserville-III transitions commands one LSB VID step (12.5mV) every 2.5µs, controlling the effective dv/dt at 5mv/µs. The ISL62882 is capable of 5mV/µs slew rate. When the ISL62882 is configured for GPU VR application, it responds to VID changes by slewing to the new voltage at a slew rate set by the logic status on the DPRSLPVR pin. The slew rate is 5mV/µs when DPRSLPVR = 0 and is doubled when DPRSLPVR = 1. When the ISL62882 is in DE mode, it will actively drive the output voltage up when the VID changes to a higher value. It’ll resume DE mode operation after reaching the new voltage level. If the load is light enough to warrant DCM, it will enter DCM after the inductor current has crossed zero for four consecutive cycles. The ISL62882 will remain in DE mode when the VID changes to a lower value. The output voltage will decay to the new value and the load will determine the slew rate. Over-voltage protection is blanked during VID down transition in DE mode until the output voltage is within 60mV of the VID value. During load insertion response, the Fast Clock function increases the PWM pulse response speed. The ISL62882 monitors the VSEN pin voltage and compares it to 100ns-filtered version. When the unfiltered version is 20mV below the filtered version, the controller knows there is a fast voltage dip due to load insertion, hence issues an additional master clock signal to deliver a PWM pulse immediately. 10mV/µs The ISL62882 can be configured for 2- or 1-phase operation. For 1-phase configuration, tie the ISEN2 pin to 5V. In this configuration, only phase-1 is active. Table 2 shows the ISL62882 configurations, programmed by the ISEN2 pin status and the Rbias value. If the ISEN2 pin is connected to the power stage, the ISL62882 is in 2-phase CPU VR configuration. Rbias = 147kΩ disables the overshoot reduction function and Rbias = 47kΩ enables it. 18 If ISEN2 is tied to 5V, the ISL62882 is configured for 1-phase operation. Rbias = 147kΩ sets 1-phase CPU VR configuration and Rbias = 47kΩ sets 1-phase GPU configuration. The R3™ modulator intrinsically has voltage feed-forward. The output voltage is insensitive to a fast slew rate input voltage change. Protections The ISL62882 provides overcurrent, current-balance, undervoltage, overvoltage, and over-temperature protections. The ISL62882 determines overcurrent protection (OCP) by comparing the average value of the droop current Idroop with an internal current source threshold. It declares OCP when Idroop is above the threshold for 120µs. A resistor Rcomp from the COMP pin to GND programs the OCP current source threshold, as well FN6890.4 June 21, 2011 ISL62882, ISL62882B as the overshoot reduction function in 1-phase configuration, as Table 4 shows. It is recommended to use the nominal Rcomp value. The ISL62882 detects the Rcomp value at the beginning of start-up, and sets the internal OCP threshold accordingly. It remembers the Rcomp value until the VR_ON signal drops below the POR threshold. TABLE 4. ISL62882 Rcomp PROGRAMABILITY 2-PHASE CONFIG. Rcomp MIN (kΩ) 1-PHASE CONFIG. NOMINAL (kΩ) MAX (kΩ) none none 40 20 320 400 480 45.3 22.7 210 235 260 41.3 20.7 155 165 175 36 18 104 120 136 37.33 20 78 85 92 38.7 22.7 62 66 70 42.7 20.7 45 50 55 44 18 OCP THRESHOLD (µA) OVERSHOOT REDUCTION FUNCTION The second level of overvoltage protection is different. If the output voltage exceeds 1.55V, the ISL62882 will immediately declare an OV fault, de-assert PGOOD, and turn on the low-side power MOSFETs. The low-side power MOSFETs remain on until the output voltage is pulled down below 0.85V when all power MOSFETs are turned off. If the output voltage rises above 1.55V again, the protection process is repeated. This behavior provides the maximum amount of protection against shorted high-side power MOSFETs while preventing output ringing below ground. Resetting VR_ON cannot clear the 1.55V OVP. Only resetting VDD will clear it. The 1.55V OVP is active all the time when the controller is enabled, even if one of the other faults have been declared. This ensures that the processor is protected against high-side power MOSFET leakage while the MOSFETs are commanded off. Disabled The ISL62882 has a thermal throttling feature. If the voltage on the NTC pin goes below the 1.18V OT threshold, the VR_TT# pin is pulled low indicating the need for thermal throttling to the system. No other action is taken within the ISL62882 in response to NTC pin voltage. Enabled Table 5 summarizes the fault protections. The default OCP threshold is the value when Rcomp is not populated. It is recommended to scale the droop current Idroop such that the default OCP threshold gives approximately the desired OCP level, then use Rcomp to fine tune the OCP level if necessary. For overcurrent conditions above 2.5x the OCP level, the PWM outputs will immediately shut off and PGOOD will go low to maximize protection. This protection is also referred to as wayovercurrent protection or fast-overcurrent protection, for shortcircuit protections. TABLE 5. FAULT PROTECTION SUMMARY FAULT TYPE Overcurrent 120µs Way-Overcurrent (2.5xOC) <2µs Overvoltage +200mV 1ms PROTECTION ACTION PWM tri-state, PGOOD latched low FAULT RESET VR_ON toggle or VDD toggle Undervoltage -300mV Phase Current Unbalance Overvoltage 1.55V The ISL62882 monitors the ISEN pin voltages to determine current-balance protection. If the ISEN pin voltage difference is greater than 9mV for 1ms, the controller will declare a fault and latch off. The ISL62882 will declare undervoltage (UV) fault and latch off if the output voltage is less than the VID set value by 300mV or more for 1ms. It’ll turn off the PWM outputs and de-assert PGOOD. FAULT DURATION BEFORE PROTECTION Over-Temperature Immediately Low-side MOSFET VDD toggle on until VCORE <0.85V, then PWM tri-state, PGOOD latched low. 1ms N/A The ISL62882 has two levels of overvoltage protections. The first level of overvoltage protection is referred to as PGOOD overvoltage protection. If the output voltage exceeds the VID set value by +200mV for 1ms, the ISL62882 will declare a fault and de-assert PGOOD. The ISL62882 takes the same actions for all of the above fault protections: de-assertion of PGOOD and turn-off of the high-side and low-side power MOSFETs. Any residual inductor current will decay through the MOSFET body diodes. These fault conditions can be reset by bringing VR_ON low or by bringing VDD below the POR threshold. When VR_ON and VDD return to their high operating levels, a soft-start will occur. 19 FN6890.4 June 21, 2011 ISL62882, ISL62882B Current Monitor The ISL62882 provides the current monitor function. The IMON pin outputs a high-speed analog current source that is 3 times of the droop current flowing out of the FB pin. Thus Equation 13: I IMON = 3 × I droop (EQ. 13) As Figures 1 and 2 show, a resistor Rimon is connected to the IMON pin to convert the IMON pin current to voltage. A capacitor can be paralleled with Rimon to filter the voltage information. The IMVP-6.5™ specification requires that the IMON voltage information be referenced to VSSSENSE. Overshoot Reduction Function The IMON pin voltage range is 0V to 1.1V. A clamp circuit prevents the IMON pin voltage from going above 1.1V. The ISL62882 has an optional overshoot reduction function. Tables 2 and 4 show to enable and disable it. FB2 Function The FB2 function is only available when the ISL62882 is in 2phase configuration. C1 R2 CONTROLLER IN 2-PHASE MODE C1 R2 CONTROLLER IN 1-PHASE MODE C3.1 C2 R3 VSEN FB2 C3.1 C2 R3 C3.2 R1 FB2 C3.2 R1 VSEN FB VREF E/A FB COMP E/A VREF COMP FIGURE 16. FB2 FUNCTION IN 2-PHASE MODE Figure 16 shows the FB2 function. A switch (called FB2 switch) turns on to short the FB and the FB2 pins when the controller is in 2-phase mode. Capacitors C3.1 and C3.2 are in parallel, serving as part of the compensator. When the controller enters 1-phase mode, the FB2 switch turns off, removing C3.2 and leaving only C3.1 in the compensator. The compensator gain will increase with the removal of C3.2. By properly sizing C3.1 and C3.2, the compensator cab be optimal for both 2-phase mode and 1-phase mode. When the FB2 switch is off, C3.2 is disconnected from the FB pin. However, the controller still actively drives the FB2 pin voltage to follow the FB pin voltage such that C3.2 voltage always follows C3.1 voltage. When the controller turns on the FB2 switch, C3.2 will be reconnected to the compensator smoothly. The FB2 function ensures excellent transient response in both 2-phase mode and 1-phase mode. If one decides not to use the FB2 function, simply populate C3.1 only. Adaptive Body Diode Conduction Time Reduction In DCM, the controller turns off the low-side MOSFET when the inductor current approaches zero. During on-time of the low-side MOSFET, phase voltage is negative and the amount is the MOSFET rDS(ON) voltage drop, which is proportional to the inductor current. A phase comparator inside the controller monitors the phase voltage during on-time of the low-side MOSFET and compares it with a threshold to determine the zero-crossing point of the inductor current. If the inductor current 20 has not reached zero when the low-side MOSFET turns off, it’ll flow through the low-side MOSFET body diode, causing the phase node to have a larger voltage drop until it decays to zero. If the inductor current has crossed zero and reversed the direction when the low-side MOSFET turns off, it’ll flow through the high-side MOSFET body diode, causing the phase node to have a spike until it decays to zero. The controller continues monitoring the phase voltage after turning off the low-side MOSFET and adjusts the phase comparator threshold voltage accordingly in iterative steps such that the low-side MOSFET body diode conducts for approximately 40ns to minimize the body diode-related loss. When a load release occurs, the energy stored in the inductors will dump to the output capacitor, causing output voltage overshoot. The inductor current freewheels through the low-side MOSFET during this period of time. The overshoot reduction function turns off the low-side MOSFET during the output voltage overshoot, forcing the inductor current to freewheel through the low-side MOSFET body diode. Since the body diode voltage drop is much higher than MOSFET Rdson voltage drop, more energy is dissipated on the low-side MOSFET therefore the output voltage overshoot is lower. If the overshoot reduction function is enabled, the ISL62882 monitors the COMP pin voltage to determine the output voltage overshoot condition. The COMP voltage will fall and hit the clamp voltage when the output voltage overshoots. The ISL62882 will turn off LGATE1 and LGATE2 when COMP is being clamped. All the low-side MOSFETs in the power stage will be turned off. When the output voltage has reached its peak and starts to come down, the COMP voltage starts to rise and is no longer clamped. The ISL62882 will resume normal PWM operation. When PSI# is low, indicating a low power state of the CPU, the controller will disable the overshoot reduction function as large magnitude transient event is not expected and overshoot is not a concern. While the overshoot reduction function reduces the output voltage overshoot, energy is dissipated on the low-side MOSFET, causing additional power loss. The more frequent transient event, the more power loss dissipated on the low-side MOSFET. The MOSFET may face severe thermal stress when transient events happen at a high repetitive rate. User discretion is advised when this function is enabled. Key Component Selection RBIAS The ISL62882 uses a resistor (1% or better tolerance is recommended) from the RBIAS pin to GND to establish highly accurate reference current sources inside the IC. Refer to Table 2 to select the resistance according to desired configuration. Do not connect any other components to this pin. Do not connect any capacitor to the RBIAS pin as it will create instability. FN6890.4 June 21, 2011 ISL62882, ISL62882B Care should be taken in layout that the resistor is placed very close to the RBIAS pin and that a good quality signal ground is connected to the opposite side of the RBIAS resistor. Ris and Cis As Figures 1 thru 4 show, the ISL62882 needs the Ris - Cis network across the ISUM+ and the ISUM- pins to stabilize the droop amplifier. The preferred values are Ris = 82.5Ω and Cis = 0.01µF. Slight deviations from the recommended values are acceptable. Large deviations may result in instability. Inductor DCR Current-Sensing Network Phase1 Phase2 Rsum ISUM+ Rsum L L Rntcs Rp DCR DCR Cn Vcn Rntc Ro Ri ISUM- ( R ntcs + R ntc ) × R p R ntcnet = --------------------------------------------------R ntcs + R ntc + R p (EQ. 15) s 1 + -----ωL A cs ( s ) = ---------------------s 1 + -----------ω sns (EQ. 16) DCR ω L = -----------L (EQ. 17) 1 ω sns = -----------------------------------------------------R sum R ntcnet × -------------N ----------------------------------------- × C n R sum R ntcnet + -------------N (EQ. 18) where N is the number of phases. Transfer function Acs(s) always has unity gain at DC. The inductor DCR value increases as the winding temperature increases, giving higher reading of the inductor DC current. The NTC Rntc values decreases as its temperature decreases. Proper selections of Rsum, Rntcs, Rp and Rntc parameters ensure that VCn represent the inductor total DC current over the temperature range of interest. There are many sets of parameters that can properly temperature-compensate the DCR change. Since the NTC network and the Rsum resistors form a voltage divider, Vcn is always a fraction of the inductor DCR voltage. It is recommended to have a higher ratio of Vcn to the inductor DCR voltage, so the droop circuit has higher signal level to work with. Ro Io FIGURE 17. DCR CURRENT-SENSING NETWORK Figure 17 shows the inductor DCR current-sensing network for a 2-phase solution. An inductor current flows through the DCR and creates a voltage drop. Each inductor has two resistors in Rsum and Ro connected to the pads to accurately sense the inductor current by sensing the DCR voltage drop. The Rsum and Ro resistors are connected in a summing network as shown, and feed the total current information to the NTC network (consisting of Rntcs, Rntc and Rp) and capacitor Cn. Rntc is a negative temperature coefficient (NTC) thermistor, used to temperature-compensate the inductor DCR change. The inductor output side pads are electrically shorted in the schematic, but have some parasitic impedance in actual board layout, which is why one cannot simply short them together for the current-sensing summing network. It is recommended to use 1Ω~10Ω Ro to create quality signals. Since Ro value is much smaller than the rest of the current sensing circuit, the following analysis will ignore it for simplicity. The summed inductor current information is presented to the capacitor Cn. Equations 14 thru 18 describe the frequency-domain relationship between inductor total current Io(s) and Cn voltage VCn(s). ⎛ ⎞ R ntcnet ⎜ DCR⎟ -------------------------------------------------V Cn ( s ) = ⎜ × ⎟ × I ( s ) × A cs ( s ) N ⎟ o R sum ⎜ ⎝ R ntcnet + ------------⎠ N 21 A typical set of parameters that provide good temperature compensation are: Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ and Rntc = 10kΩ (ERT-J1VR103J). The NTC network parameters may need to be fine tuned on actual boards. One can apply full load DC current and record the output voltage reading immediately; then record the output voltage reading again when the board has reached the thermal steady state. A good NTC network can limit the output voltage drift to within 2mV. It is recommended to follow the Intersil evaluation board layout and current-sensing network parameters to minimize engineering time. VCn(s) also needs to represent real-time Io(s) for the controller to achieve good transient response. Transfer function Acs(s) has a pole ωsns and a zero ωL. One needs to match ωL and ωsns so Acs(s) is unity gain at all frequencies. By forcing ωL equal to ωsns and solving for the solution, Equation 19 gives Cn value. L C n = -----------------------------------------------------------R sum R ntcnet × -------------N ----------------------------------------- × DCR R sum R ntcnet + -------------N (EQ. 19) (EQ. 14) FN6890.4 June 21, 2011 ISL62882, ISL62882B io io iL Vo Vo RING BACK FIGURE 18. DESIRED LOAD TRANSIENT RESPONSE WAVEFORMS FIGURE 21. OUTPUT VOLTAGE RING BACK PROBLEM io ISUM+ Vo Rntcs Cn.1 Cn.2 Vcn Rp FIGURE 19. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO SMALL Rntc Rn OPTIONAL ISUM- Ri io Rip Cip OPTIONAL Vo FIGURE 20. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO LARGE For example, given N = 2, Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ, Rntc = 10kΩ, DCR = 0.88mΩ and L = 0.36µH, Equation 19 gives Cn = 0.294µF. Assuming the compensator design is correct, Figure 18 shows the expected load transient response waveforms if Cn is correctly selected. When the load current Icore has a square change, the output voltage Vcore also has a square response. If Cn value is too large or too small, VCn(s) will not accurately represent real-time Io(s) and will worsen the transient response. Figure 19 shows the load transient response when Cn is too small. Vcore will sag excessively upon load insertion and may create a system failure. Figure 20 shows the transient response when Cn is too large. Vcore is sluggish in drooping to its final value. There will be excessive overshoot if load insertion occurs during this time, which may potentially hurt the CPU reliability. 22 FIGURE 22. OPTIONAL CIRCUITS FOR RING BACK REDUCTION Figure 21 shows the output voltage ring back problem during load transient response. The load current io has a fast step change, but the inductor current iL cannot accurately follow. Instead, iL responds in first order system fashion due to the nature of current loop. The ESR and ESL effect of the output capacitors makes the output voltage Vo dip quickly upon load current change. However, the controller regulates Vo according to the droop current idroop, which is a real-time representation of iL; therefore it pulls Vo back to the level dictated by iL, causing the ring back problem. This phenomenon is not observed when the output capacitor have very low ESR and ESL, such as all ceramic capacitors. Figure 22 shows two optional circuits for reduction of the ring back. Cn is the capacitor used to match the inductor time constant. It usually takes the parallel of two (or more) capacitors to get the desired value. Figure 22 shows that two capacitors Cn.1 and Cn.2 are in parallel. Resistor Rn is an optional component to reduce the Vo ring back. At steady state, Cn.1 + Cn.2 provides the desired Cn capacitance. At the beginning of io change, the effective capacitance is less because Rn increases the impedance of the Cn.1 branch. As Figure 19 explains, Vo tends to dip when Cn is too small, and this effect will reduce the Vo ring back. This effect is more pronounced when Cn.1 is much larger than Cn.2. It is also more pronounced when Rn is bigger. However, the presence of Rn increases the ripple of the Vn signal if Cn.2 is too small. It is recommended to keep Cn.2 greater than 2200pF. Rn value usually is a few ohms. Cn.1, Cn.2 and Rn values should be determined through tuning the load transient response waveforms on an actual board. FN6890.4 June 21, 2011 ISL62882, ISL62882B Rip and Cip form an R-C branch in parallel with Ri, providing a lower impedance path than Ri at the beginning of io change. Rip and Cip do not have any effect at steady state. Through proper selection of Rip and Cip values, idroop can resemble io rather than iL, and Vo will not ring back. The recommended value for Rip is 100Ω. Cip should be determined through tuning the load transient response waveforms on an actual board. The recommended range for Cip is 100pF~2000pF. However, it should be noted that the Rip -Cip branch may distort the idroop waveform. Instead of being triangular as the real inductor current, idroop may have sharp spikes, which may adversely affect idroop average value detection and therefore may affect OCP accuracy. User discretion is advised. Resistor Current-Sensing Network PHASE1 Overcurrent Protection Refer to Equation 1 on page 16 and Figures 12, 17 and 23; resistor Ri sets the droop current Idroop. Table 4 shows the internal OCP threshold. It is recommended to design Idroop without using the Rcomp resistor. For example, the OCP threshold is 40µA for 2-phase solution. We will design Idroop to be 34.3µA at full load, so the OCP trip level is 1.16x of the full load current. For inductor DCR sensing, Equation 23 gives the DC relationship of Vcn(s) and Io(s). ⎛ ⎞ R ntcnet ⎜ DCR⎟ V Cn = ⎜ ----------------------------------------- × ------------⎟ × I o R sum N ⎟ ⎜ ⎝ R ntcnet + ------------⎠ N (EQ. 23) PHASE2 Substitution of Equation 23 into Equation 1 gives Equation 24: L L DCR DCR R ntcnet 2 DCR I droop = ----- × ----------------------------------------- × ------------ × I o R sum Ri N R ntcnet + -------------N Therefore: RSUM ISUM+ RSUM RSEN (EQ. 24) RSEN VCN RO CN RI ISUM- 2R ntcnet × DCR × I o R i = -------------------------------------------------------------------------------R sum N × ⎛ R ntcnet + --------------⎞ × I droop ⎝ N ⎠ Substitution of Equation 15 and application of the OCP condition in Equation 25 gives Equation 26: ( R ntcs + R ntc ) × R p 2 × --------------------------------------------------- × DCR × I omax R ntcs + R ntc + R p R i = ------------------------------------------------------------------------------------------------------------------------( R ⎛ ntcs + R ntc ) × R p R sum⎞ N × ⎜ --------------------------------------------------- + --------------⎟ × I droopmax N ⎠ ⎝ R ntcs + R ntc + R p RO IO FIGURE 23. RESISTOR CURRENT-SENSING NETWORK Figure 23 shows the resistor current-sensing network for a 2-phase solution. Each inductor has a series current-sensing resistor Rsen. Rsum and Ro are connected to the Rsen pads to accurately capture the inductor current information. The Rsum and Ro resistors are connected to capacitor Cn. Rsum and Cn form a filter for noise attenuation. Equations 20 thru 22 give VCn(s) expression R sen V Cn ( s ) = ------------ × I o ( s ) × A Rsen ( s ) N 1 A Rsen ( s ) = ---------------------s 1 + -----------ω sns (EQ. 25) (EQ. 26) where Iomax is the full load current, Idroopmax is the corresponding droop current. For example, given N = 2, Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ, Rntc = 10kΩ, DCR = 0.88mΩ, Iomax = 51A and Idroopmax = 34.3µA, Equation 26 gives Ri = 998Ω. For resistor sensing, Equation 27 gives the DC relationship of Vcn(s) and Io(s). R sen V Cn = ------------ × I o N (EQ. 27) (EQ. 20) Substitution of Equation 27 into Equation 1 gives Equation 28: (EQ. 21) 2 R sen I droop = ----- × ------------ × I o N Ri (EQ. 28) Therefore 1 ω Rsen = --------------------------R sum -------------- × C n N (EQ. 22) Transfer function ARsen(s) always has unity gain at DC. Currentsensing resistor Rsen value will not have significant variation over-temperature, so there is no need for the NTC network. The recommended values are Rsum = 1kΩ and Cn = 5600pF. 23 2R sen × I o R i = --------------------------N × I droop (EQ. 29) Substitution of Equation 29 and application of the OCP condition in Equation 25 gives Equation 30: 2R sen × I omax R i = -------------------------------------N × I droopmax (EQ. 30) FN6890.4 June 21, 2011 ISL62882, ISL62882B where Iomax is the full load current, Idroopmax is the corresponding droop current. For example, given N = 2, Rsen = 1mΩ, Iomax = 51A and Idroopmax = 34.3µA, Equation 30 gives Ri = 1.487kΩ. A resistor from COMP to GND can adjust the internal OCP threshold, providing another dimension of fine-tune flexibility. Table 4 shows the detail. It is recommended to scale Idroop such that the default OCP threshold gives approximately the desired OCP level, then use Rcomp to fine tune the OCP level if necessary. Load Line Slope Refer to Figure 12. For inductor DCR sensing, substitution of Equation 24 into Equation 2 gives the load line slope expression: 2R droop R ntcnet V droop DCR LL = ------------------ = ---------------------- × ----------------------------------------- × -----------N Io Ri R sum R ntcnet + -------------N (EQ. 31) For resistor sensing, substitution of Equation 28 into Equation 2 gives the load line slope expression: 2R sen × R droop V droop LL = ------------------ = ----------------------------------------N × Ri Io V Rimon × R droop R imon = -------------------------------------------3I o × LL (EQ. 37) For example, given LL = 1.9mΩ, Rdroop = 2.825kΩ, VRimon = 963mV at Iomax = 51A, Equation 37 gives Rimon = 9.358kΩ. A capacitor Cimon can be paralleled with Rimon to filter the IMON pin voltage. The RimonCimon time constant is the user’s choice. It is recommended to have a time constant long enough such that switching frequency ripples are removed. Compensator Figure 18 shows the desired load transient response waveforms. Figure 24 shows the equivalent circuit of a voltage regulator (VR) with the droop function. A VR is equivalent to a voltage source (= VID) and output impedance Zout(s). If Zout(s) is equal to the load line slope LL, i.e., constant output impedance, in the entire frequency range, Vo will have square response when Io has a square change. (EQ. 32) Zout(s) = LL Substitution of Equation 25 and rewriting Equation 31, or substitution of Equation 29 and rewriting Equation 32 give the same result in Equation 33: Io R droop = ---------------- × LL I droop Rewriting Equation 36 and application of full load condition gives Equation 37: VID VR i o LOAD V o (EQ. 33) One can use the full load condition to calculate Rdroop. For example, given Iomax = 51A, Idroopmax = 34.3µA and LL = 1.9mΩ, Equation 33 gives Rdroop = 2.825kΩ. It is recommended to start with the Rdroop value calculated by Equation 33, and fine tune it on the actual board to get accurate load line slope. One should record the output voltage readings at no load and at full load for load line slope calculation. Reading the output voltage at lighter load instead of full load will increase the measurement error. Current Monitor Refer to Equation 13 for the IMON pin current expression. Refer to Figures 1 and 2, the IMON pin current flows through Rimon. The voltage across Rimon is expressed in Equation 34: V Rimon = 3 × I droop × R imon (EQ. 34) Rewriting Equation 33 gives Equation 35: Io I droop = ------------------ × LL R droop (EQ. 35) Substitution of Equation 35 into Equation 34 gives Equation 36: 3I o × LL V Rimon = --------------------- × R imon R droop (EQ. 36) FIGURE 24. VOLTAGE REGULATOR EQUIVALENT CIRCUIT Intersil provides a Microsoft Excel-based spreadsheet to help design the compensator and the current sensing network, so the VR achieves constant output impedance as a stable system. Figure 27 shows a screenshot of the spreadsheet. A VR with active droop function is a dual-loop system consisting of a voltage loop and a droop loop which is a current loop. However, neither loop alone is sufficient to describe the entire system. The spreadsheet shows two loop gain transfer functions, T1(s) and T2(s), that describe the entire system. Figure 25 conceptually shows T1(s) measurement set-up and Figure 26 conceptually shows T2(s) measurement set-up. The VR senses the inductor current, multiplies it by a gain of the load line slope, then adds it on top of the sensed output voltage and feeds it to the compensator. T(1) is measured after the summing node, and T2(s) is measured in the voltage loop before the summing node. The spreadsheet gives both T1(s) and T2(s) plots. However, only T2(s) can be actually measured on an ISL62882 regulator. T1(s) is the total loop gain of the voltage loop and the droop loop. It always has a higher crossover frequency than T2(s) and has more meaning of system stability. T2(s) is the voltage loop gain with closed droop loop. It has more meaning of output voltage response. Design the compensator to get stable T1(s) and T2(s) with sufficient phase margin, and output impedance equal or smaller than the load line slope. 24 FN6890.4 June 21, 2011 ISL62882, ISL62882B Vo L Q1 Vin Q1 GATE Q2 DRIVER io Cout VIN GATE Q2 DRIVER COUT Ω 20 Ω COMP COMP VID CHANNEL B CHANNEL A ISOLATION TRANSFORMER CHANNEL A CHANNEL B NETWORK ANALYZER EXCITATION OUTPUT FIGURE 25. LOOP GAIN T1(s) MEASUREMENT SET-UP 25 20 EA MOD. EA MOD. I O LOAD LINE SLOPE LOAD LINE SLOPE LOOP GAIN = VO L CHANNEL B LOOP GAIN = CHANNEL A VID ISOLATION TRANSFORMER CHANNEL A CHANNEL B NETWORK ANALYZER EXCITATION OUTPUT FIGURE 26. LOOP GAIN T2(s) MEASUREMENT SET-UP FN6890.4 June 21, 2011 Compensation & Current Sensing Network Design for Intersil Multiphase R^3 Regulators for IMVP-6.5 Jia Wei, [email protected], 919-405-3605 Attention: 1. "Analysis ToolPak" Add-in is required. To turn on, go to Tools--Add-Ins, and check "Analysis ToolPak". 2. Green cells require user input Compensator Parameters Operation Parameters Controller Part Number: ISL6288x § s · § s · ¸ ¨1 ¸ KZi Zi ¨¨1 Phase Number: 2 2Sf z1 ¸¹ ¨© 2Sf z 2 ¸¹ © AV ( s ) Vin: 12 volts § s ·¸ §¨ s ·¸ Vo: 1.15 volts s ¨1 1 ¨ 2Sf p1 ¹¸ ©¨ 2Sf p 2 ¹¸ © Full Load Current: 50 Amps 26 87 3 470 4.5 0.6 30 10 3 3 300 0.36 0.9 1.9 33.1 % uF m: nH Recommended Value R1 2.870 k : R2 387.248 k : R3 0.560 k : C1 188.980 pF C2 498.514 pF C3 32.245 pF uF m: nH kHz uH m: m: uA User-Selected Value R1 2.87 k : R2 412 k : R3 0.562 k : C1 150 pF C2 390 pF C3 32 pF Use User-Selected Value (Y/N)? N Performance and Stability T1 Bandwidth: 190kHz T2 Bandwidth: 52kHz T1 Phase Margin: 63.4° T2 Phase Margin: 94.7° Changing the settings in red requires deep understanding of control loop design Place the 2nd compensator pole fp2 at: 1.9 xfs (Switching Frequency) Tune Ki to get the desired loop gain bandwidth Tune the compensator gain factor Ki: (Recommended Ki range is 0.8~2) Loop Gain, Gain Curve 7V 7V Recommended Value Cn 0.294 uF Ri 1014.245 : ( ( ( )UHTXHQF\+] ( Loop Gain, Phase Curve 7V 7V ( ( ( )UHTXHQF\+] ( ( ( ( ( )UHTXHQF\+] ( ( ( ( ( 3KDVHGHJUHH ( 3KDVHGHJUHH Output Impedance, Gain Curve 0DJQLWXGHPRKP *DLQG% 1.15 Output Impedance, Phase Curve ( ( ( ( )UHTXHQF\+] ( Operation Parameters Inductor DCR 0.88 m : Rsum 3.65 k : Rntc 10 k : Rntcs 2.61 k : Rp 11 k : ( FN6890.4 June 21, 2011 FIGURE 27. SCREENSHOT OF THE COMPENSATOR DESIGN SPREADSHEET User Selected Value Cn 0.294 uF Ri 1000 : ISL62882, ISL62882B Estimated Full-Load Efficiency: Number of Output Bulk Capacitors: Capacitance of Each Output Bulk Capacitor: ESR of Each Output Bulk Capacitor: ESL of Each Output Bulk Capacitor: Number of Output Ceramic Capacitors: Capacitance of Each Output Ceramic Capacitor: ESR of Each Output Ceramic Capacitor: ESL of Each Output Ceramic Capacitor: Switching Frequency: Inductance Per Phase: CPU Socket Resistance: Desired Load-Line Slope: Desired ISUM- Pin Current at Full Load: (This sets the over-current protection level) Current Sensing Network Parameters ISL62882, ISL62882B Optional Slew Rate Compensation Circuit For 1-Tick VID Transition where Cout is the total output capacitance. In the mean time, the Rvid-Cvid branch current Ivid time domain expression is: –t ------------------------------⎞ dV fb ⎛ R ×C I vid ( t ) = C vid × ------------ × ⎜ 1 – e vid vid⎟ ⎜ ⎟ dt ⎝ ⎠ Rdroop Vcore Rvid Cvid OPTIONAL FB It is desired to let Ivid(t) cancel Idroop_vid(t). So there are: Ivid dV fb C out × LL dV core C vid × ------------ = ------------------------ × -----------------dt dt R droop Idroop_vid E/A COMP Σ VDACDAC VIDs VID<0:6> RTN X1 INTERNAL TO IC (EQ. 39) VSSSENSE VSS (EQ. 40) and: (EQ. 41) R vid × C vid = C out × LL The result is expressed in Equation 42: (EQ. 42) R vid = R droop and: VID<0:6> dV core C out × LL ----------------dt C vid = ------------------------ × -----------------R droop dV fb -----------dt Vfb (EQ. 43) Ivid For example: given LL = 1.9mΩ, Rdroop = 2.87kΩ, Cout = 1710µF, dVcore/dt = 5mV/µs and dVfb/dt = 15mV/µs, Equation 42 gives Rvid = 2.87kΩ and Equation 43 gives Cvid = 377pF. Vcore It’s recommended to select the calculated Rvid value and start with the calculated Cvid value and tweak it on the actual board to get the best performance. Idroop_vid FIGURE 28. OPTIONAL SLEW RATE COMPENSATION CIRCUIT FOR1-TICK VID TRANSITION During a large VID transition, the DAC steps through the VIDs at a controlled slew rate. For example, the DAC may change a tick (12.5mV) per 2.5µs per, controlling output voltage Vcore slew rate at 5mV/µs. Figure 28 shows the waveforms of 1-tick VID transition. During 1-tick VID transition, the DAC output changes at approximately 15mV/µs slew rate, but the DAC cannot step through multiple VIDs to control the slew rate. Instead, the control loop response speed determines Vcore slew rate. Ideally, Vcore will follow the FB pin voltage slew rate. However, the controller senses the inductor current increase during the up transition, as the Idroop_vid waveform shows, and will droop the output voltage Vcore accordingly, making Vcore slew rate slow. Similar behavior occurs during the down transition. To control Vcore slew rate during 1-tick VID transition, one can add the Rvid-Cvid branch, whose current Ivid cancels Idroop_vid. When Vcore increases, the time domain expression of the induced Idroop change is –t -------------------------⎞ C out × LL dV core ⎛ C × LL⎟ ⎜ out I droop ( t ) = ------------------------ × ------------------ × 1 – e ⎜ ⎟ dt R droop ⎝ ⎠ 27 During normal transient response, the FB pin voltage is held constant, therefore is virtual ground in small signal sense. The Rvid - Cvid network is between the virtual ground and the real ground, and hence has no effect on transient response. Voltage Regulator Thermal Throttling 54µA 64µA VR_TT# SW1 NTC + VNTC - + RNTC Rs 1.24V SW2 1.20V INTERNAL TO ISL62882 FIGURE 29. CIRCUITRY ASSOCIATED WITH THE THERMAL THROTTLING FEATURE OF THE ISL62882 (EQ. 38) FN6890.4 June 21, 2011 ISL62882, ISL62882B Figure 29 shows the thermal throttling feature with hysteresis. An NTC network is connected between the NTC pin and GND. At low temperature, SW1 is on and SW2 connects to the 1.20V side. The total current flowing out of the NTC pin is 60µA. The voltage on NTC pin is higher than threshold voltage of 1.20V and the comparator output is low. VR_TT# is pulled up by the external resistor. When temperature increases, the NTC thermistor resistance decreases so the NTC pin voltage drops. When the NTC pin voltage drops below 1.20V, the comparator changes polarity and turns SW1 off and throws SW2 to 1.24V. This pulls VR_TT# low and sends the signal to start thermal throttle. There is a 6µA current reduction on NTC pin and 40mV voltage increase on threshold voltage of the comparator in this state. The VR_TT# signal will be used to change the CPU operation and decrease the power consumption. When the temperature drops down, the NTC thermistor voltage will go up. If NTC voltage increases to above 1.24V, the comparator will flip back. The external resistance difference in these two conditions is shown in Equation 44: Current Balancing Refer to Figures 1 and 2. The ISL62882 achieves current balancing through matching the ISEN pin voltages. Rs and Cs form filters to remove the switching ripple of the phase node voltages. It is recommended to use rather long RsCs time constant such that the ISEN voltages have minimal ripple and represent the DC current flowing through the inductors. Recommended values are Rs = 10kΩ and Cs = 0.22µF. Layout Guidelines Table 6 shows the layout considerations. The designators refer to the reference design shown in Figure 31. TABLE 6. LAYOUT CONSIDERATION PIN NAME LAYOUT CONSIDERATION EP GND Create analog ground plane underneath the controller and the analog signal processing components. Don’t let the power ground plane overlap with the analog ground plane. Avoid noisy planes/traces (e.g.: phase node) from crossing over/overlapping with the analog plane. 1 PGOOD No special consideration One needs to properly select the NTC thermistor value such that the required temperature hysteresis correlates to 2.96kΩ resistance change. A regular resistor may need to be in series with the NTC thermistor to meet the threshold voltage values. 2 PSI# No special consideration 3 RBIAS Place the RBIAS resistor (R16) in general proximity of the controller. Low impedance connection to the analog ground plane. For example, given Panasonic NTC thermistor with B = 4700, the resistance will drop to 0.03322 of its nominal at +105°C, and drop to 0.03956 of its nominal at +100°C. If the required temperature hysteresis is +105°C to +100°C, the required resistance of NTC will be as shown in Equation 45: 4 VR_TT# No special consideration 5 NTC The NTC thermistor (R9) needs to be placed close to the thermal source that is monitor to determine thermal throttling. Usually it’s placed close to phase-1 high-side MOSFET. 6 VW Place the capacitor (C4) across VW and COMP in close proximity of the controller Therefore, a larger value thermistor such as 470k NTC should be used. 7 COMP 8 FB Place the compensator components (C3, C5, C6 R7, R11, R10 and C11) in general proximity of the controller. At +105°C, 470kΩ NTC resistance becomes (0.03322 × 470kΩ) = 15.6kΩ. With 60µA on the NTC pin, the voltage is only (15.6kΩ × 60µA) = 0.937V. This value is much lower than the threshold voltage of 1.20V. Therefore, a regular resistor needs to be in series with the NTC. The required resistance can be calculated by Equation 46: 9 FB2 10 ISEN2 A capacitor (C9) decouples it to VSUM-. Place it in general proximity of the controller. 11 ISEN1 A capacitor (C10) decouples it to VSUM-. Place it in general proximity of the controller. 12 VSEN 13 RTN Place the VSEN/RTN filter (C12, C13) in close proximity of the controller for good decoupling. 1.24V 1.20V --------------- – --------------- = 2.96k 54μA 60μA (EQ. 44) 2.96kΩ ------------------------------------------------------- = 467kΩ ( 0.03956 – 0.03322 ) (EQ. 45) 1.20V --------------- – 15.6kΩ = 4.4kΩ 60μA (EQ. 46) 4.42k is a standard resistor value. Therefore, the NTC branch should have a 470k NTC and 4.42k resistor in series. The part number for the NTC thermistor is ERTJ0EV474J. It is a 0402 package. NTC thermistor will be placed in the hot spot of the board. 28 FN6890.4 June 21, 2011 ISL62882, ISL62882B TABLE 6. LAYOUT CONSIDERATION (Continued) TABLE 6. LAYOUT CONSIDERATION (Continued) PIN NAME LAYOUT CONSIDERATION PIN NAME LAYOUT CONSIDERATION 14 ISUM- 26 LGATE2 15 ISUM+ Place the current sensing circuit in general proximity of the controller. Place C82 very close to the controller. Place NTC thermistors R42 next to phase-1 inductor (L1) so it senses the inductor temperature correctly. Each phase of the power stage sends a pair of VSUM+ and VSUM- signals to the controller. Run these two signals traces in parallel fashion with decent width (>20mil). IMPORTANT: Sense the inductor current by routing the sensing circuit to the inductor pads. Route R63 and R71 to the phase-1 side pad of inductor L1. Route R88 to the output side pad of inductor L1. Route R65 and R72 to the phase-2 side pad of inductor L2. Route R90 to the output side pad of inductor L2. If possible, route the traces on a different layer from the inductor pad layer and use vias to connect the traces to the center of the pads. If no via is allowed on the pad, consider routing the traces into the pads from the inside of the inductor. The following drawings show the two preferred ways of routing current sensing traces. 27 VSSP2 Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing VSSP2 to the phase-2 lowside MOSFET (Q5 and Q1) source pins instead of general power ground plane for better performance. 28 PHASE2 29 UGATE2 30 BOOT2 Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 31~37 VID0~6 No special consideration. 38 VR_ON No special consideration. Inductor Inductor 39 40 Current-Sensing Traces DPRSLPVR No special consideration. CLK_EN# No special consideration. Other Phase Node Minimize phase node copper area. Don’t let the phase node copper overlap with/getting close to other sensitive traces. Cut the power ground plane to avoid overlapping with phase node copper. Other Vias Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing PHASE2 trace to the phase-2 high-side MOSFET (Q4 and Q10) source pins instead of general phase-2 node copper. Minimize the loop consisting of input capacitor, high-side MOSFETs and low-side MOSFETs (e.g., C27, C33, Q2, Q8, Q3 and Q9). Current-Sensing Traces 16 VDD A capacitor (C16) decouples it to GND. Place it in close proximity of the controller. 17 VIN A capacitor (C17) decouples it to GND. Place it in close proximity of the controller. 18 IMON Place the filter capacitor (C21) close to the CPU. 19 BOOT1 Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 20 UGATE1 21 PHASE1 Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing PHASE1 trace to the phase-1 high-side MOSFET (Q2 and Q8) source pins instead of general phase-1 node copper. 22 VSSP1 23 LGATE1a 24 LGATE1b 25 VCCP Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing VSSP1 to the phase-1 low-side MOSFET (Q3 and Q9) source pins instead of general power ground plane for better performance. A capacitor (C22) decouples it to GND. Place it in close proximity of the controller. 29 FN6890.4 June 21, 2011 8 7 IN C3 R10 15PF 100PF 10UF C27 10UF C33 C54 VCORE 22UF C55 22UF C56 22UF C40 22UF C41 22UF C59 C60 22UF OUT 22UF C61 Q9 0.88UH 2.3MOHM 22UF Q3 +5V IRF7832 C52 IN VCCP ISL62882HRZ COMP 1UF LGATE2 IRF7832 470UF 4MOHM VW L1 VSSP2 U6 NTC C LGATE1B LGATE1A FB2 VSSP1 ISEN2 2.37K 270PF R7 C24 PHASE2 VR_TT# C11 56UF UGATE2 RBIAS DNP DNP ------------ Q2 BOOT2 PSI# R16 ---- R12 499 ------- ------------ C4 1000PF C6 IRF7821 PGOOD 47.5K ------R8 R9 IN PHASE1 R11 422K 6.98K EP ISEN1 VSEN RTN ISUMISUM+ VDD VIN IMON BOOT UGATE1 C83 R110 DNP DNP -------- -------- ---- IN VIN C22 OUT IN FB OPTIONAL ---- 1 R20 0 R37 +5V VIN IN R40 IN 0 0.22UF R18 1UF C17 IN 1 C16 VSSSENSE C13 ---- B R56 C30 0 0.22UF OUT IMON IN VSSSENSE 0.01UF IN OPTIONAL ---- C21 10 R50 VCCSENSE R17 22.6K IN 330PF 1000PF ----- VCORE ----C12 B LAYOUT R63 10K 2.61K NTC R41 -----> R42 11K R38 DNP DNP -----------OPTIONAL 0.1UF ---- C20 R30 3.01K -----------C81 R109 ---- 0.15UF C18 0.056UF C82 R26 C15 A 0.01UF 82.5 10 3.65K ROUTE LGATE TRACE IN PARALLEL WITH THE VSSP TRACE GOING TO THE SOURCE OF Q3 PLACE NEAR L1 TITLE: ISL62881 DATE: ENGINEER: PAGE: GPU REFERENCE DESIGN 1-PHASE, DCR SENSING FIGURE 30. 1-PHASE GPU APPLICATON REFERENCE DESIGN 8 7 6 5 NOTE: ROUTE UGATE TRACE IN PARALLEL WITH THE PHASE TRACE GOING TO THE SOURCE OF Q2 4 3 JIA WEI 2 3/16/2009 1 OF 1 1 A ISL62882, ISL62882B ---- 10K DNP ------R6 ------R4 C 2 D OPTIONAL ----OPTIONAL ---- VR_TT# 3 IN CLK_EN# DPRSLPVR VR_ON VID6 VID5 VID4 VID3 VID2 VID1 VID0 30 PGOOD +1.1V 4 IN R19 D 5 IN IN IN IN IN IN IN 1.91K VID0 VID1 VID2 VID3 VID4 VID5 VID6 VR_ON DPRSLPVR +3.3V 6 FN6890.4 June 21, 2011 5 4 VIN 0.1UF ---- 1200PF 100 ----------OPTIONAL IMON C44 DNP C57 470UF C52 470UF C39 470UF 10UF 10UF 10UF 10UF C56 C64 C55 C63 C48 10UF 10UF 10UF 10UF C54 C62 C47 10UF 10UF 10UF 10UF C53 C61 C43 10UF 10UF 10UF 10UF C50 C60 C42 10UF 10UF 10UF 10UF C41 C40 10UF C49 10UF C59 10UF C65 10UF 1 VSUM- C70 C78 10UF C69 C75 C68 C74 10UF C67 C73 10UF C66 C72 10UF VSUM- C71 ISEN1 10UF 10K R90 ISEN2 R72 R65 3.65K VSUM+ VSUM+ 1 C B IN VSSSENSE IN VSUM+ ROUTE UGATE1 TRACE IN PARALLEL WITH THE PHASE1 TRACE GOING TO THE SOURCE OF Q2 AND Q8 ROUTE LGATE1 TRACE IN PARALLEL WITH THE VSSP1 TRACE GOING TO THE SOURCE OF Q3 AND Q9 SAME RULE APPLIES TO OTHER PHASES IN VSUM- PLACE NEAR L1 TITLE: ISL62882 REFERENCE DESIGN 2-PHASE, DCR SENSING ENGINEER: 8 10UF 10UF 10UF 10UF C33 C27 OUT 0.01UF R50 R41 -----> R42 R38 11K 1UF C17 0 +5V VIN C20 R26 IN R30 1K ----------C81 R109 ---- 10 C82 C16 IN R18 C15 ---- 0.01UF 82.5 ----C12 10 IN C13 VSSSENSE OPTIONAL ---- R17 IN R40 0.22UF 1 0.33UF R20 1000PF 330PF --------- IN VCCSENSE Q9 VCORE LAYOUT NOTE: R37 0 VCORE Q3 1UF EP C18 C10 C9 OPTIONAL ----ISEN2 IN ISEN1 IN 0.22UF 0.36UH IRF7832 PHASE1 C22 2.87K 0 IRF7832 VSSP1 ISEN2 R11 412K C30 LGATE1A FB2 10K 2.61K NTC 390PF +5V LGATE1B C21 562 IN VCCP ISL62882HRZ R56 10K R88 VW L1 VSSP2 LGATE2 9.31K C11 0.22UF R7 R10 0.22UF 10PF NTC R71 DNP PHASE2 U6 R63 DNP IRF7821 Q8 3.65K VR_TT# RBIAS 0.047UF C4 1000PF 8.06K DNP ------R6 ------R4 CLK_EN# DPRSLPVR VR_ON VID6 VID5 VID4 VID3 VID2 VID1 VID0 R19 R12 499 147K R9 IRF7821 Q2 OUT 560PF 2.87K ------------- 0.36UH IRF7832 Q11 OUT ------------C83 R110 IRF7832 Q5 OUT 7 6 5 4 FIGURE 31. 2-PHASE CPU APPLICATION REFERENCE DESIGN 3 JIA WEI 2 DATE: JULY 2009 PAGE: 1 OF 1 1 A ISL62882, ISL62882B R8 BOOT2 FB C6 A 0.22UF UGATE2 COMP 22PF C3 C31 0 OUT R16 PSI# C5 150PF R57 OUT B D OUT PGOOD VR_TT# OUT ---- OPTIONAL ---- IRF7821 Q10 OUT IN IN C ---- IRF7821 Q4 1.91K PGOOD OUT PSI# +1.1V C28 R23 IN ISEN1 VSEN RTN ISUMISUM+ VDD VIN IMON BOOT UGATE1 31 +3.3V 1 L2 1.91K D 2 IN C24 VID0 IN VID1 IN VID2 IN VID3 IN VID4 IN VID5 IN VID6 IN VR_ON IN DPRSLPVR IN CLK_EN# OUT 3 10UF C34 6 56UF 7 56UF C25 8 FN6890.4 June 21, 2011 ISL62882, ISL62882B 1-Phase GPU Application Reference Design Bill of Materials QTY REFERENCE VALUE DESCRIPTION MANUFACTURER 1 C11 270pF Multilayer Cap, 16V, 10% GENERIC H1045-00271-16V10 SM0603 1 C12 330pF Multilayer Cap, 16V, 10% GENERIC H1045-00331-16V10 SM0603 1 C13 1000pF Multilayer Cap, 16V, 10% GENERIC H1045-00102-16V10 SM0603 1 C15 0.01µF Multilayer Cap, 16V, 10% GENERIC H1045-00103-16V10 SM0603 2 C16,C22 1µF Multilayer Cap, 16V, 20% GENERIC H1045-00105-16V20 SM0603 1 C18 0.15µF Multilayer Cap, 16V, 10% GENERIC H1045-00154-16V10 SM0603 1 C20 0.1µF Multilayer Cap, 16V, 10% GENERIC H1045-00104-16V10 SM0603 3 C17, C21, C30 0.22µF Multilayer Cap, 25V, 10% GENERIC H1045-00224-25V10 SM0603 1 C24 56µF Radial SP Series Cap, 25V, 20% SANYO 25SP56M CASE-CC 2 C27,C33 10µF Multilayer Cap, 25V, 20% GENERIC H1065-00106-25V20 SM1206 1 C3 100pF Multilayer Cap, 16V, 10% GENERIC H1045-00101-16V10 SM0603 1 C52 470µF SPCAP, 2V, 4MΩ POLYMER CAP, 2.5V, 4.5MΩ PANASONIC EEXSX0D471E4 T520V477M2R5A(1)E4R5-6666 1 C4 1000pF Multilayer Cap, 16V, 10% 8 C40, C41, C54-C56, C59-C61 10µF 1 C6 15pF 1 C82 0 C81, C83 1 L1 KEMET PART NUMBER PACKAGE GENERIC H1045-00102-16V10 SM0603 Multilayer Cap, 6.3V, 20% MURATA PANASONIC TDK GRM21BR61C106KE15L ECJ2FB0J106K C2012X5R0J106K SM0805 Multilayer Cap, 16V, 10% GENERIC H1045-00150-16V10 SM0603 0.056µF Multilayer Cap, 16V, 10% GENERIC H1045-00563-16V10 SM0603 MPC1040LR88 10mmx10mm DNP 0.88µH Inductor, Inductance 20%, DCR 7% NEC-TOKIN 1 Q2 N-Channel Power MOSFET IR IRF7821 PWRPAKSO8 2 Q3, Q9 N-Channel Power MOSFET IR IRF7832 PWRPAKSO8 1 R10 2.37k Thick Film Chip Resistor, 1% GENERIC H2511-02371-1/16W1 SM0603 1 R11 6.98k Thick Film Chip Resistor, 1% GENERIC H2511-06981-1/16W1 SM0603 1 R16 47.5k Thick Film Chip Resistor, 1% GENERIC H2511-04752-1/16W1 SM0603 2 R17, R18 10 Thick Film Chip Resistor, 1% GENERIC H2511-00100-1/16W1 SM0603 1 R19 1.91k Thick Film Chip Resistor, 1% GENERIC H2511-01911-1/16W1 SM0603 1 R26 82.5 Thick Film Chip Resistor, 1% GENERIC H2511-082R5-1/16W1 SM0603 3 R20, R40, R56 0 Thick Film Chip Resistor, 1% GENERIC H2511-00R00-1/16W1 SM0603 1 R30 3.01k Thick Film Chip Resistor, 1% GENERIC H2511-03011-1/16W1 SM0603 1 R37 1 Thick Film Chip Resistor, 1% GENERIC H2511-01R00-1/16W1 SM0603 1 R38 11k Thick Film Chip Resistor, 1% GENERIC H2511-01102-1/16W1 SM0603 1 R41 2.61k Thick Film Chip Resistor, 1% GENERIC H2511-02611-1/16W1 SM0603 1 R42 PANASONIC ERT-J1VR103J SM0603 1 R50 GENERIC H2511-02262-1/16W1 SM0603 10k NTC Thermistor, 10k NTC 22.6k 32 Thick Film Chip Resistor, 1% FN6890.4 June 21, 2011 ISL62882, ISL62882B 1-Phase GPU Application Reference Design Bill of Materials (Continued) QTY REFERENCE VALUE DESCRIPTION MANUFACTURER PART NUMBER PACKAGE 1 R6 10k Thick Film Chip Resistor, 1% GENERIC H2511-01002-1/16W1 SM0603 1 R63 3.65k Thick Film Chip Resistor, 1% GENERIC H2511-03651-1/16W1 SM0805 1 R7 412k Thick Film Chip Resistor, 1% GENERIC H2511-04123-1/16W1 SM0603 0 R109, R110, R4, R8, R9 DNP 1 U6 IMVP-6.5 PWM Controller INTERSIL ISL62882HRTZ QFN-40 2-Phase CPU Application Reference Design Bill of Materials QTY REFERENCE VALUE 1 C11 390pF Multilayer Cap, 16V, 10% GENERIC H1045-00391-16V10 SM0603 1 C12 330pF Multilayer Cap, 16V, 10% GENERIC H1045-00331-16V10 SM0603 1 C13 1000pF Multilayer Cap, 16V, 10% GENERIC H1045-00102-16V10 SM0603 2 C15, C21 0.01µF Multilayer Cap, 16V, 10% GENERIC H1045-00103-16V10 SM0603 2 C16,C22 Multilayer Cap, 16V, 20% GENERIC H1045-00105-16V20 SM0603 1 C18 0.33µF Multilayer Cap, 16V, 10% GENERIC H1045-00334-16V10 SM0603 1 C20 Multilayer Cap, 16V, 10% GENERIC H1045-00104-16V10 SM0603 5 C9, C10, C17, C30, C31 0.22µF Multilayer Cap, 25V, 10% GENERIC H1045-00224-25V10 SM0603 2 C24,C25 56µF Radial SP Series Cap, 25V, 20% SANYO 25SP56M CASE-CC 4 C27,C28,C33,C34 10µF Multilayer Cap, 25V, 20% GENERIC H1065-00106-25V20 SM1206 1 C3 150pF Multilayer Cap, 16V, 10% GENERIC H1045-00151-16V10 SM0603 3 C39, C52, C57 470µF SPCAP, 2V, 4MΩ POLYMER CAP, 2.5V, 4.5MΩ PANASONIC KEMET EEXSX0D471E4 T520V477M2R5A(1)E4R5-6666 1 C4 1000pF Multilayer Cap, 16V, 10% GENERIC H1045-00102-16V10 SM0603 30 C40-C43, C47-C50, C53-C56, C59-C75, C78 10µF Multilayer Cap, 6.3V, 20% MURATA PANASONIC TDK GRM21BR61C106KE15L ECJ2FB0J106K C2012X5R0J106K SM0805 1 C5 22pF Multilayer Cap, 16V, 10% GENERIC H1045-00220-16V10 SM0603 1 C6 10pF Multilayer Cap, 16V, 10% GENERIC H1045-00100-16V10 SM0603 1 C81 1200pF Multilayer Cap, 16V, 10% GENERIC H1045-00122-16V10 SM0603 1 C82 0.047µF Multilayer Cap, 16V, 10% GENERIC H1045-00473-16V10 SM0603 1 C83 560pF Multilayer Cap, 16V, 10% GENERIC H1045-00561-16V10 SM0603 2 L1, L2 0.36µH Inductor, Inductance 20%, DCR 7% NEC-TOKIN PANASONIC MPCH1040LR36 ETQP4LR36AFC 10mmx10mm 4 Q2, Q4, Q8, Q10 N-Channel Power MOSFET IR IRF7821 PWRPAKSO8 4 Q3, Q5, Q9, Q11 N-Channel Power MOSFET IR IRF7832 PWRPAKSO8 1µF 0.1µF DESCRIPTION MANUFACTURER PART NUMBER PACKAGE 1 R10 562 Thick Film Chip Resistor, 1% GENERIC H2511-05620-1/16W1 SM0603 1 R109 100 Thick Film Chip Resistor, 1% GENERIC H2511-01000-1/16W1 SM0603 1 R11 2.87k Thick Film Chip Resistor, 1% GENERIC H2511-02871-1/16W1 SM0603 1 R110 2.87k Thick Film Chip Resistor, 1% GENERIC H2511-02871-1/16W1 SM0603 1 R12 499 Thick Film Chip Resistor, 1% GENERIC H2511-04990-1/16W1 SM0603 1 R16 147k Thick Film Chip Resistor, 1% GENERIC H2511-01473-1/16W1 SM0603 33 FN6890.4 June 21, 2011 ISL62882, ISL62882B 2-Phase CPU Application Reference Design Bill of Materials (Continued) QTY REFERENCE VALUE DESCRIPTION 2 R17, R18 10 Thick Film Chip Resistor, 1% GENERIC H2511-00100-1/16W1 SM0603 3 R19, R71, R72 10k Thick Film Chip Resistor, 1% GENERIC H2511-01002-1/16W1 SM0603 1 R23 1.91k Thick Film Chip Resistor, 1% GENERIC H2511-01911-1/16W1 SM0603 1 R26 82.5 Thick Film Chip Resistor, 1% GENERIC H2511-082R5-1/16W1 SM0603 4 R20, R40, R56, R57 0 Thick Film Chip Resistor, 1% GENERIC H2511-00R00-1/16W1 SM0603 1 R30 1k Thick Film Chip Resistor, 1% GENERIC H2511-01001-1/16W1 SM0603 3 R37, R88, R90 1 Thick Film Chip Resistor, 1% GENERIC H2511-01R00-1/16W1 SM0603 1 R38 11k Thick Film Chip Resistor, 1% GENERIC H2511-01102-1/16W1 SM0603 1 R4 DNP 1 R41 2.61k Thick Film Chip Resistor, 1% GENERIC H2511-02611-1/16W1 SM0603 1 R42 PANASONIC ERT-J1VR103J SM0603 1 R50 9.31k Thick Film Chip Resistor, 1% GENERIC H2511-09311-1/16W1 SM0603 1 R6 8.06k Thick Film Chip Resistor, 1% GENERIC H2511-08061-1/16W1 SM0603 2 R63, R65 3.65k Thick Film Chip Resistor, 1% GENERIC H2511-03651-1/16W1 SM0805 2 R8, R9 DNP 1 R7 412k Thick Film Chip Resistor, 1% GENERIC H2511-04123-1/16W1 SM0603 1 U6 IMVP-6.5 PWM Controller INTERSIL ISL62882HRTZ QFN-40 10k NTC Thermistor, 10k NTC 34 MANUFACTURER PART NUMBER PACKAGE FN6890.4 June 21, 2011 ISL62882, ISL62882B Typical Performance 92 1.10 90 1.08 1.06 86 1.04 84 VIN = 8V 82 VOUT (V) EFFICIENCY (%) 88 VIN = 12V 80 78 VIN = 19V 1.02 1.00 0.98 76 0.96 74 0.94 72 70 0.92 0 5 10 15 20 25 30 IOUT (A) 35 40 45 50 0 55 FIGURE 32. 2-PHASE CCM EFFICIENCY, VID = 1.075V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V 5 10 15 20 25 30 35 IOUT (A) 40 45 50 55 60 65 FIGURE 33. 2-PHASE CCM LOAD LINE, VID = 1.075V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V 90 85 0.885 VIN = 8V VIN = 12V VOUT (V) EFFICIENCY (%) 0.875 80 75 VIN = 19V 70 0.865 0.855 65 0.845 60 0.835 55 0.1 0.825 1 10 100 IOUT (A) 0 1 2 3 4 5 6 7 8 9 IOUT (A) 10 11 12 13 14 15 FIGURE 34. 1-PHASE DEM EFFICIENCY, VID = 0.875V, DPRSLPVR IS ASSERTED FOR IOUT < 3A, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V. SOLID LINES: ISL62882 EFFICIENCY, DOTTED LINES: WOULD-BE EFFICIENCY IF LGATE1b WAS NOT TURNED OFF IN DPRSLPVR MODE FIGURE 35. 1-PHASE DEM LOAD LINE, VID = 0.875V, DPRSLPVR IS ASSERTED FOR IOUT < 3A VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V FIGURE 36. 2-PHASE CPU MODE SOFT-START, VIN = 19V, IO = 0A, VID = 0.95V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2 FIGURE 37. 2-PHASE CPU MODESHUT DOWN, VIN = 19V, IO = 1A, VID = 0.95V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2 35 FN6890.4 June 21, 2011 ISL62882, ISL62882B Typical Performance (Continued) FIGURE 38. 2-PHASE CPU MODE CLK_EN# DELAY, VIN = 19V, IO = 2A, VID = 1.5V, Ch1: PHASE1, Ch2: VO, Ch4: CLK_EN# FIGURE 39. 2-PHASE CPU MODE PRE-CHARGED START UP, VIN = 19V, VID = 0.95V, Ch1: PHASE1, Ch2: VO, Ch4: VR_ON FIGURE 40. STEADY STATE, VIN = 19V, IO = 0A, VID = 1.075V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2 FIGURE 41. STEADY STATE, VIN = 19V, IO = 35A, VID = 1.075V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2 FIGURE 42. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 19V, VID = 1.075V, IO = 15A/50A, di/dt = “FASTEST” FIGURE 43. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 19V, VID = 1.075V, IO = 15A/50A, di/dt = “FASTEST” 36 FN6890.4 June 21, 2011 ISL62882, ISL62882B Typical Performance (Continued) FIGURE 44. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 19V, VID = 1.075V, IO = 15A/50A, di/dt = “FASTEST” FIGURE 45. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 19V, VID = 1.075V, IO = 15A/50A, di/dt = “FASTEST” FIGURE 46. 2-PHASE CPU MODE DEEPER SLEEP MODE ENTRY/EXIT, IO = 1.5A, HFM VID = 1.075V, LFM VID = 0.875V, DEEPER SLEEP VID = 0.875V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, CH4: DPRSLPVR FIGURE 47. 2-PHASE CPU MODE VID ON THE FLY, 1.075V/0.875V, 2-PHASE CONFIGURATION, PSI# = 1, DPRSLPVR = 0, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2 FIGURE 48. PHASE ADDING (PSI# TOGGLE), IO = 15A, VID = 1.075V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: N/A FIGURE 49. PHASE DROPPING (PSI# TOGGLE), IO = 15A, VID = 1.075V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: N/A 37 FN6890.4 June 21, 2011 ISL62882, ISL62882B Typical Performance (Continued) Phase Margin Gain FIGURE 51. 2-PHASE CPU MODE REFERENCE DESIGN LOOP GAIN T2(s) MEASUREMENT RESULT 1000 5.0 900 4.5 800 4.0 700 3.5 Z(f) (mΩ) IMON-VSSSENSE (mV) FIGURE 50. TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION ENABLED, VIN = 19V, VID = 0.95V, IO = 12A/51A, di/dt = “FASTEST”, Ch1: PHASE1, Ch2: VO, Ch3: N/A, Ch4: LGATE1 600 500 VIN = 12V SPEC 400 VIN = 8V 300 PSI# = 1, DPRSLPVR = 0, 2-Phase CCM 2.5 2.0 1.5 VIN = 19V 200 3.0 1.0 100 0.5 0 0 5 10 15 20 25 30 IOUT (A) 35 40 45 FIGURE 52. IMON, VID = 1.075 FIGURE 54. 1-PHASE GPU MODE SOFT-START, DPRSLPVR=0, VIN = 8V, IO = 0A, VID = 1.2375V, Ch1: PHASE1, Ch2: VO 38 50 0.0 PSI# = 0, DPRSLPVR = 0, 1-Phase DE 1k 1M 100k 10k FREQUENCY (Hz) FIGURE 53. REFERENCE DESIGN FDIM RESULT FIGURE 55. 1-PHASE GPU MODE SHUT DOWN, VIN = 8V, IO = 1A, VID = 1.2375V, Ch1: PHASE1, Ch2: VO FN6890.4 June 21, 2011 ISL62882, ISL62882B Typical Performance (Continued) FIGURE 56. 1-PHASE GPU MODE VID TRANSITION, DPRSLPVR = 0, IO = 2A, VID = 1.2375V/1.0375V, Ch2: VO, Ch3: VID4 39 FIGURE 57. 1-PHASE GPU MODE VID TRANSITION, DPRSLPVR = 1, IO = 2A, VID = 1.2375V/1.0375V, Ch2: VO, Ch3: VID4 FN6890.4 June 21, 2011 ISL62882, ISL62882B Revision History The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you have the latest Rev. DATE REVISION CHANGE 5/6/11 FN6890.4 Updated to most current Intersil template. Page 8, Electrical Spec table: Added min and max limits for IMON Output Current, Condition ISUM – pin current = 5µA, Min: 22 Max: 37.5 Page 8, Note 6: Updated over temp note in Min Max column of spec tables from "Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design." To: "Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested." 2/10/11 FN6890.3 Page 7, Electrical Spec table: removed min and max limits for IMON Output Current, Condition ISUM – pin current = 5µA Pg 2 - Updated Tape & Reel note in Ordering Information from "Add “-T” suffix for tape and reel." to new standard "Add “-T*” suffix for tape and reel." The "*" covers all possible tape and reel options. Removed -T FGs that are covered by "-T*" note (IRTZ-T, HRTZ-T, BHRTZ-T) Updated Intersil Trademark statement at bottom of page 1 per directive from Legal. Updated over temp note in Min Max column of spec tables from "Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested." to new standard "Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design." Electrical Spec table: Removed Note 7 "Limits established by characterization and are not production tested." and references to it in the table. Updated L40.5x5 POD to rev .1. Changes from rev 0: Added Note 7 (JEDEC reference drawing: MO-220WHHE-1) Added Note 4 callout to Dimension b in bottom view 12/3/09 FN6890.2 Removed ISL62882A device from data sheet. 11/4/09 FN6890.2 Converted to new Intersil template. On page 19, Modes of Operation section last paragraph Changed from "Rbias = 147kohm enables the overshoot reduction function and Rbias = 47kohm disables it" to "Rbias = 147kohm disables the overshoot reduction function and Rbias = 47kohm enables it". Applied Intersil Standards as follows: Ordering information with notes and links, Added bold verbiage to Electrical spec conditions for over-temp and bolded min and max value columns. Pin Descriptions placed in Table. 8/24/09 FN6890.1 8/18/09 - See attached .doc file for changes. 7/10/09: Updated Figures 1, 2, 10, 11 and 27. Per Jia, “All the drawings have updated the way ISEN capacitors are connected. They used to be connected to from ISEN to GND, now they are connected from ISEN to Vo. It’s an application patch that helps to avoid false IBAL fault during phase dropping due to an IC design error.” Changed “GND” to “VSUM-“ for pins 10 and 11 in table 5. Pin 10 now reads “A capacitor (C9) decouples it to VSUM-. Place it in general proximity of the controller.” Pin 11 now reads “A capacitor (C10) decouples it to VSUM-. Place it in general proximity of the controller.” 5/19/09: Changed under Recommended Operating Conditions- Battery Voltage VIN from "+5V to 21V" to "+5V to 25V" 04/01/09 FN6890.0 Initial Release to web Products Intersil Corporation is a leader in the design and manufacture of high-performance analog semiconductors. The Company's products address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks. Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a complete list of Intersil product families. *For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page on intersil.com: ISL62882, ISL62882B To report errors or suggestions for this datasheet, please go to www.intersil.com/askourstaff FITs are available from our website at http://rel.intersil.com/reports/search.php For additional products, see www.intersil.com/product_tree Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted in the quality certifications found at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 40 FN6890.4 June 21, 2011 ISL62882, ISL62882B Package Outline Drawing L40.5x5 40 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 1, 9/10 4X 3.60 5.00 A B 36X 0.40 6 PIN #1 INDEX AREA 3.50 5.00 6 PIN 1 INDEX AREA 0.15 (4X) 40X 0.4± 0 .1 BOTTOM VIEW TOP VIEW 0.20 b 4 0.10 M C A B PACKAGE OUTLINE 0.40 0.750 SEE DETAIL “X” SIDE VIEW 3.50 5.00 0.050 // 0.10 C C BASE PLANE SEATING PLANE 0.08 C (36X 0.40 0.2 REF (40X 0.20) C (40X 0.60) 5 0.00 MIN 0.05 MAX TYPICAL RECOMMENDED LAND PATTERN DETAIL "X" NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.27mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 indentifier may be 7. JEDEC reference drawing: MO-220WHHE-1 either a mold or mark feature. 41 FN6890.4 June 21, 2011 ISL62882, ISL62882B Package Outline Drawing L48.6x6 48 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 1, 4/07 4X 4.4 6.00 44X 0.40 A B 6 PIN 1 INDEX AREA 6 PIN #1 INDEX AREA 48 37 1 6.00 36 4 .40 ± 0.15 25 12 0.15 (4X) 13 24 0.10 M C A B 0.05 M C TOP VIEW 48X 0.45 ± 0.10 4 48X 0.20 BOTTOM VIEW SEE DETAIL "X" 0.10 C BASE PLANE MAX 0.80 ( SEATING PLANE 0.08 C ( 44 X 0 . 40 ) ( 5. 75 TYP ) C SIDE VIEW 4. 40 ) C 0 . 2 REF 5 ( 48X 0 . 20 ) 0 . 00 MIN. 0 . 05 MAX. ( 48X 0 . 65 ) DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 indentifier may be either a mold or mark feature. 42 FN6890.4 June 21, 2011