DATASHEET

DATASHEET
4-Phase Interleaved Boost PWM Controller with Light
Load Efficiency Enhancement
ISL78225
Features
The ISL78225 4-phase controller is targeted for applications
where high efficiency (>95%) and high power are required. The
multiphase boost converter architecture uses interleaved timing
to multiply channel ripple frequency and reduce input and output
ripple. Lower ripple results in fewer input/output capacitors and
therefore lower component cost and smaller implementation
area.
• Peak current mode PWM control with adjustable slope
compensation
The ISL78225 has a dedicated pin to initiate the phase dropping
scheme for higher efficiency at light load by dropping phases
based on the load current, so the switching and core losses in the
converter are reduced significantly. As the load increases, the
dropped phase(s) are added back to accommodate heavy load
transients and improve efficiency.
• Phase dropping facilitated with companion FET driver
ISL78420 (featuring tri-level input control)
• Precision resistor/DCR current sensing
• 2-, 3- or 4-phase operation
• Adjustable phase dropping/diode emulation/pulse skipping
for high efficiency at light load
• Adjustable switching frequency or external synchronization
from 75kHz up to 1MHz per phase
• Over-temperature/overvoltage protection
• 2V 1.0% internal reference
Input current is sensed continuously by measuring the voltage
across a dedicated current sense resistor or inductor DCR. This
current sensing provides precision channel-current balancing,
and per-phase overcurrent protection. A separate totalizing
current limit function provides overcurrent protection for all the
phases combined. This two-stage current protection provides
maximum performance and circuit reliability.
• Pb-free 44 Ld 10x10 EP-TQFP (RoHS compliant)
• -40°C to +125°C operating temperature range
• AEC-Q100 qualified
Applications
• Automotive power supplies
- Start/stop DC/DC converter
- Electronic power steering systems (EPAS)
- Fuel pumps
- Injection system
The ISL78225 can also provide for input voltage tracking via the
VREF2 pin. The comparison reference voltage will be the lower of
the VREF2 pin or the internal 2V reference. By using a resistor
network between VIN and VREF2 pin, the output voltage can track
input voltage to limit the output power during automotive cranking
conditions.
The ISL78225 can output a clock signal for expanding operation to
8 phases, which offers high system flexibility. The
threshold-sensitive enable input is available to accurately
coordinate the start-up of the ISL78225 with any other voltage rail.
0.98
• Telecom and industrial power supplies
WITH PHASE DROPPING
WITHOUT PHASE DROPPING
0.93
EFFICIENCY (%)
• Audio trunk amplifier power supplies
0.88
0.83
0.78
0.73
16V INPUT, 36V OUTPUT
SYNCHRONOUS BOOST
0
1
2
3
4
5
6
7
8
9
10
OUTPUT CURRENT (A)
FIGURE 1. EFFICIENCY vs OUTPUT CURRENT vs PHASE DROPPING MODE
September 5, 2014
FN7909.4
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
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ISL78225
Pin Configuration
FS
SS
VCC
GND
MODE
IOUT
VIN_SEN
VIN_OVB
VOUT_SEN
VOUT_OVB
DMAX
EN
PGOOD
ISL78225
(44 LD 10x10 EP-TQFP)
TOP VIEW
44 43 42 41 40 39 38 37 36 35 34
33
2
32
1
VIN
DNC
27
ISEN2N
PLL_COMP
8
26
DNC
SYNC
9
25
DNC
CLK_OUT
10
24
ISEN3P
PWM_INV
11
23
12 13 14 15 16 17 18 19 20 21 22
ISEN3N
ISEN1P
ISEN2P
7
DNC
28
SLOPE
ISEN1N
6
DNC
ISEN4N
GND
DRIVE_EN
29
PWM4
ISEN4P
5
PWM2
30
VREF2
P_COM
DNC
4
PWM3
31
FB
PWM1
3
PWM_TRI
COMP
Functional Pin Descriptions
PIN #
SYMBOL
1
FS
A resistor placed from FS to ground will set the PWM switching frequency.
DESCRIPTION
2
SS
Use this pin to set up the desired soft-start time. A capacitor placed from SS to ground will set up the soft-start
ramp rate and in turn determine the soft-start time.
3
COMP
4
FB
5
VREF2
The output of the transconductance amplifier. Place the compensation network between COMP and GND for
compensation loop design.
The inverting input of the transconductance amplifier. A resistor network should be placed between the FB pin and
output rail to set the output voltage.
External reference input to the transconductance amplifier. When the VREF2 pin voltage drops below 1.8V, the
internal reference will be shifted from 2V to VREF2. The VREF2 voltage can be programmed by connecting a
resistor divider network from VCC or VIN.
6
GND
7
SLOPE
8
PLL_COMP
9
SYNC
Frequency synchronization pin. Connecting the SYNC pin to an external square pulse waveform (typically 20% to
80% duty cycle) will synchronize the converter switching frequency to the fundamental frequency of the input
waveform. If SYNC function is not used, tie the SYNC pin to GND. A 500nA current source is connected internally
to pull-down the SYNC pin if it is left open.
10
CLK_OUT
This pin provides a clock signal to synchronize with another ISL78225. This provides scalability and flexibility. The
rising edge signal on the CLKOUT pin is in phase with the leading edge of the PWM1 signal.
11
PWM_INV
This pin determines the polarity of the PWM output signal. Pulling this pin to GND will force normal operation with
inverting MOSFET drivers. Pulling this pin to VCC will invert the PWM signal for operation with non-inverting
MOSFET drivers. This function provides the flexibility for the ISL78225 to work with different drivers.
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Bias and reference ground for the IC.
This pin programs the slope of the internal slope compensation. A resistor should be connected from the SLOPE
pin to GND. Please refer to “Adjustable Slope Compensation” on page 18 for how to choose the resistor value.
This pin serves as the compensation node for the PLL. A second order passive loop filter connected between
PLL_COMP pin and GND compensates the PLL feedback loop.
2
FN7909.4
September 5, 2014
ISL78225
Functional Pin Descriptions
(Continued)
PIN #
SYMBOL
DESCRIPTION
12
PWM_TRI
This pin enables the tri-level of the PWM output signal. Pulling this pin to GND forces the PWM output to be
traditional two level logic. Pulling the PWM_TRI pin to VCC will enable tri-level PWM signals, then the PWM output
can be at the 2.5V tri-level condition.
13, 14, 16,
17
PWM1, PWM3, PWM2,
PWM4
Pulse width modulation outputs. Connect these pins to the PWM input pins of the external driver ICs. The number
of active channels is determined by the state of PWM3, PWM4. For 2-phase operation, connect PWM3 to VCC;
similarly, connect PWM4 to VCC for 3-phase operation.
15
P_COM
19
DRIVE_EN
18, 20,
25, 26, 31,
32
DNC
PWM Compensation pin; connect this pin through resistor to VCC.
Driver enable output pin. This pin is connected to the enable pin of MOSFET drivers.
Do Not Connect – These pins must be left floating.
21, 22, 23, ISEN1N, ISEN1P, ISEN3N, The ISENxP and ISENxN pins are current sense inputs to individual differential amplifiers. The sensed current is
24, 27, 28, ISEN3P, ISEN2N, ISEN2P, used as a reference for current mode control and overcurrent protection. Inactive channels should have their
respective ISENxN pins connected to VIN and ISENxP pins left open or tied to VIN. The ISL78225 utilizes external
ISEN4N, ISEN4P
29, 30
sense resistor current sensing method or Inductor DCR sensing method.
33
VIN
Connect input rail to this pin. This pin is connected to the internal linear regulator, generating the power necessary
to operate the chip. It is recommended the DC voltage applied to the VIN pin does not exceed 40V.
34
VCC
This pin is the output of the internal linear regulator that supplies the bias and gate voltage for the IC. A minimum
4.7µF decoupling ceramic capacitor should be connected from VCC to GND. The controller starts to operate when
the voltage on this pin exceeds the rising POR threshold and shuts down when the voltage on this pin drops below
the falling POR threshold. This pin can be connected directly to a +5V supply if VIN falls below 5.6V.
35
GND
Bias and reference ground for the IC.
36
MODE
37
IOUT
IOUT is the current monitor pin with an additional OCP adjustment function. An RC network needs to be placed
between IOUT and GND to ensure the proper operation. The voltage at the IOUT pin will be proportional to the input
current. If the voltage on the IOUT pin is higher than 2V, ISL78225 will go into overcurrent protection mode and
the chip will latch off until the EN pin is toggled.
38
VIN_SEN
The VIN_SEN pin is used for sensing the VIN voltage. A resistor divider network is connected between this pin and
boost power stage input voltage rail. When the voltage on VIN_SEN is greater than 2.4V, the VIN_OVB pin will be
pulled low to indicate an input overvoltage condition. The threshold voltage can be programmed by changing the
divider ratios.
39
VIN_OVB
The VIN_OVB pin is an open-drain indicator of an overvoltage condition at the input. When the voltage on the
VIN_SEN pin is greater than the 2.4V threshold, the VIN_OVB pin will be pulled low.
40
VOUT_SEN
The VOUT_SEN pin is used for sensing the output voltage; a resistor divider network is connected between this pin
and output voltage rail. When the voltage on VOUT_SEN pin is greater than 2.4V, VOUT_OVB pin will be pulled low,
indicating an output overvoltage condition.
41
VOUT_OVB
The VOUT_OVB pin is an open-drain indicator of an overvoltage condition at the output. When the voltage on the
VOUT_SEN pin is greater than the 2.4V threshold, the VOUT_OVB pin will be pulled low and latched, toggling VIN
or EN will reset the latch.
42
DMAX
43
EN
This pin is a threshold-sensitive enable input for the controller. Connecting the power supply input to EN pin
through an appropriate resistor divider provides a means to synchronize power-up of the controller and the
MOSFET driver ICs. When EN pin is driven above 1.2V, the ISL78225 is active depending on status of the internal
POR, and pending fault states. Driving the EN pin below 1.1V will clear all fault states and the ISL78225 will
soft-start when re-enabled.
44
PGOOD
This pin is used as an indication of the end of soft-start and output regulation. It is an open-drain logic output that
is low impedance until the soft-start is completed. It will be pulled low again once the UV/OV/OC/OT conditions
are detected.
Mode selection pin. Pull this pin to logic HIGH for forced PWM mode; phase dropping/adding is inactive during
forced PWM mode. Connecting a resistor from MODE pin to GND will initialize phase dropping mode (PDM). In
PDM, a 5µA fixed reference current will flow out of the MODE pin, and the phase dropping threshold can be
programmed by adjusting the resistor value.
DMAX pin sets the maximum duty cycle of the PWM modulator. If the DMAX pin is connected to GND, the
maximum duty cycle will be set to 91.7%. Floating this pin will limit the duty cycle to 75% and connecting the
DMAX pin to VCC will limit the duty cycle to 83.3%.
Exposed Pad
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It is recommended to solder the Exposed Pad to the ground plane.
3
FN7909.4
September 5, 2014
ISL78225
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART
MARKING
ISL78225ANEZ
TEMP RANGE
(°C)
ISL78225 ANEZ
-40 to +125
PACKAGE
(Pb-free)
PKG.
DWG. #
44 Ld EP-TQFP
Q44.10x10A
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL78225. For more information on MSL please see techbrief TB363.
ISL78225 Block Diagram
VIN_OVB
OV_IN
VIN_SEN
VOUT_SEN
2.4V
2.4V
PGOOD
OC_ALL
OC_PH
VOUT_OVB
SYNC
OV_OUT
OV_IN
UV
OC
OT
SYNC
DETECT
REF
VIN
FAULT CONTROL
CIRCUITS
2V
5V LDO
VCC
S
Q
POR
2.4V
1.2V
CLK_OUT
DMAX
R
EN
OVER
TEMP
PLL_COMP
VCO
OV_OUT
DMAX
FB
FS
OT
UV
0.8Vref
SLOPE
COMPENSATION
5µA
SLOPE
SS
SOFT- START LOGIC
DRIVE_EN
ISEN1P
ISEN1
20k
CSA
ISEN1N
DUPLICATE FOR EACH CHANNEL
OC_PH
160µA
S
R1
2V
Gm
VREF2
DMAX
OT
OC
OV_OUT
FB
PH3
COMP
R2
IOUT1
ZCD
Q
(FOR PH1 &
PH2 ONLY)
PWM CONTROL
PWM1
PWM_TRI
PH4
PWM_INV
PHASE DROP CONTROL
IOUT1
MODE
ADDER
MODE
IOUT4
IOUT
OC_ALL
2V
GND
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4
P_COM
FN7909.4
September 5, 2014
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Typical Application 1: 4-Phase Synchronous Boost Converter with Sense Resistor Current
Sensing
VIN
+
EN
5
VCC
UGATE
PHASE
DRIVER
VIN
NC
NC
FB
ISEN4P
VREF2
ISEN4N
ISL78225
GND
ISEN2P
ISEN2N
SLOPE
NC
SYNC
NC
NC
ISEN1N
NC
PWM4
PWM2
ISEN3P
ISEN3N
ISEN1P
ISEN4P
ISEN4N
ISEN2P
ISEN2N
EN
ISEN3P
ISEN3N
PWM3
VOUT
Phase 2
Phase 3
LOAD
EN
ISEN4P
ISEN4N
PWM4
Phase 4
ISEN3P
ISEN3N
PWM4
P_COM
PWM2
PWM1
PWM3
PWM3
PWM1
PWM_INV
PWM_TRI
DRIVE_EN
PLL_COMP
CLK_OUT
EN
ISEN2P
ISEN2N
PWM2
VCC
GND
MODE
IOUT
Phase 1
VIN_SEN
VOUT_SEN
LGATE
COMP
VCC
PWM
VCC
ISEN1P
ISEN1N
Note: Please see ISL78420 for an Automotive Qualified 100V synchronous boost driver.
+
ISL78225
SS
VOUT_OVB
DMAX
PGOOD
FS
EN
VOUT_SEN
VIN_OVB
VOUT_SEN
PWM1
FN7909.4
September 5, 2014
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Typical Application 2: 4-Phase Standard Boost Converter with DCR Current Sensing
DCR
VIN
C
+
L
R
VCC
6
VOUT_SEN
EN
PWM1
VCC
GND
IOUT
MODE
VIN_SEN
EN
ISEN2P
ISEN2N
PWM2
VIN
NC
NC
FB
ISEN4P
VREF2
ISEN4N
ISL78225
GND
ISEN2N
NC
SYNC
NC
ISEN1N
NC
NC
PWM4
ISEN3P
ISEN3N
ISEN1P
PWM4
PWM2
P_COM
PWM2
PWM3
PWM1
PWM3
PWM1
DRIVE_EN
PLL_COMP
PWM_INV
PWM_TRI
ISEN4N
ISEN2P
ISEN2P
SLOPE
CLK_OUT
ISEN4P
VCC
ISEN1P
ISEN1N
ISEN2N
EN
ISEN3P
ISEN3N
PWM3
EN
ISEN4P
ISEN4N
PWM4
VOUT
Phase 2
Phase 3
LOAD
Phase 4
ISEN3P
ISEN3N
+
ISL78225
COMP
VCC
DRIVER
LGATE
Phase 1
VIN_OVB
VOUT_SEN
SS
VOUT_OVB
EN
PGOOD
FS
DMAX
VOUT_SEN
PWM
FN7909.4
September 5, 2014
ISL78225
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V to +45V
All ISEN_ Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VIN -5V to VIN +0.3V
VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .GND -0.3V to +6V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V to VCC +0.3V
ESD Rating
Human Body Model (Tested per JESD22-A114E) . . . . . . . . . . . . . . .2.5kV
Machine Model (Tested per JESD-A115-A) . . . . . . . . . . . . . . . . . . . 200V
Charge Device Model (Tested per AEC-Q100-11) . . . . . . . . . . . . . . 1.5kV
Latch Up (Tested per JESD78B, Class II, Level A) . . . . . . . . . . . . . . . 100mA
Thermal Resistance (Typical)
JA (°C/W) JC (°C/W)
44 Ld EP-TQFP Package (Notes 4, 5) . . . . . .
28
2.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see TB493
Operating Conditions
Voltage at VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5.6V to +40V
All ISEN_ Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VIN-5V to VIN+0.3V
Voltage at VCC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5%
Ambient Temperature (Auto) . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Operating Conditions: VIN = 12V, TA = -40°C to +125°C, unless otherwise specified. Typical specifications are
at TA = +25°C. Boldface limits apply across the operating temperature range, -40°C to +125°C.
PARAMETER
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNITS
5.6
12
40
V
8
12
mA
10
µA
SUPPLY INPUT
Input Voltage Range
Input Supply Current (Normal Mode)
VIN = 12V, RFS = 158kΩ(For fSW = 250kHz)EN = 5V
Input Supply Current (Shutdown Mode)
VIN = 12V, RFS = 158kΩ(For fSW = 250kHz)EN = 0V
INTERNAL LINEAR REGULATOR
LDO Output Voltage (VCC Pin)
VIN > 5.6V, CL = 4.7µF from VCC to GND, IVCC < 50mA
LDO Current Limit (VCC pin)
VCC = 3V, CL = 4.7µF from VCC to GND
4.75
5
5.25
200
(Note 7)
V
mA
POWER-ON RESET (POR) AND ENABLE
POR Threshold
VCC Rising
EN Threshold
4.4
4.5
4.6
V
VCC Falling
4.1
4.2
4.3
V
Rising
1.1
1.2
1.3
V
Hysteresis
70
mV
OSCILLATOR
Accuracy of Switching Frequency Setting
RFS = 158kΩfrom FS to GND
Adjustment Range of Switching Frequency
225
250
75
FS pin voltage
275
kHz
1000
kHz
1
V
SOFT-START
Soft-Start Current
CSS = 2.2nF from SS to GND
Soft-Start Prebias Voltage Range
4
5
0
Soft-Start Prebias Voltage Accuracy
VFB = 500mV
-25
Soft-Start Clamp Voltage
6
µA
2
V
25
mV
3.4
V
REFERENCE VOLTAGE
System Accuracy
-40°C to +125°C, measure at FB pin, VREF2 > 2.5V
FB Pin Input Bias Current
VFB = 2V, VREF2 > 2.5V
VREF2 Pin Input Bias Current
VREF2 = 1.6V
VREF2 External Reference Voltage Range
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7
1.98
2.02
V
1
µA
-1
1
µA
0.7
1.8
V
-1
2
FN7909.4
September 5, 2014
ISL78225
Electrical Specifications
Operating Conditions: VIN = 12V, TA = -40°C to +125°C, unless otherwise specified. Typical specifications are
at TA = +25°C. Boldface limits apply across the operating temperature range, -40°C to +125°C. (Continued)
PARAMETER
TEST CONDITIONS
VREF2 External Reference Voltage Accuracy
MIN
(Note 6)
TYP
MAX
(Note 6)
UNITS
%
-40°C to +125°C, measure at FB pin, VREF2 = 1.8V
-1
1
-40°C to +125°C, measure at FB pin, VREF2 = 0.7V
-1.5
1.5
ERROR AMPLIFIER
Transconductance Gain
2
mS
Output Impedance
5
MΩ
11
MHz
2.5
V/µs
±300
µA
Unity Gain Bandwidth
CCOMP = 100pF from COMP pin to GND
Slew Rate
CCOMP = 100pF from COMP pin to GND
Output Current Capability
Maximum Output Voltage
3.5
V
Minimum Output Voltage
0.5
V
6
%
PWM CORE
Duty Cycle Matching
IISENxP = 60µA, RSLOPE = 30.1k, fSW = 250kHz,
VCOMP = 2V, 4-phase, TA = +25°C
Zero Crossing Detection (ZCD) Threshold for
PWM1/PWM2
RSEN1, 2 = 750Ω
Leading Edge Blanking (Audio Mode)
VMODE = VCC, VPWM_TRI = VCC, VCOMP = 0.5V
Leading Edge Blanking (Other Mode)
VMODE <4V or VPWM_TRI = GND, VCOMP = 0.5V
SLOPE pin Voltage
-6
VISENxN = VISENxP, from VIN - 1V to VIN
ISENxN, ISENxP Common Mode Voltage
Range
VIN > 12V
mV
Ts/12
(Note 8)
ns
130
385
ISENxN Bias Current
3
515
ns
650
0.3
VIN-5
mV
µA
VIN
V
0.5
V
PWMx OUTPUT
PWMx Output Voltage LOW
IPWMx = -500µA
PWMx Output Voltage HIGH
IPWMx = +500µA
4.5
PWMx Tri-State Output Voltage
IPWMx = ±100µA
2.3
PWMx Pull-Down Current
During Phase Detection Time (t3 on Figure 14), VPWM = 1V
PWM3, PWM4 Disable Threshold
During Phase Detection Time (t3 on Figure 14)
3.5
MODE Pull-up Current
VMODE = 2.4V
4.2
5.1
5.6
VIOUT Threshold, 4-phase, Drop Phase 4
VMODE = 1.6V
1.175
1.2
1.225
V
VIOUT Threshold, 4-phase, Drop Phase 3
VMODE = 1.6V
0.775
0.8
0.825
V
VIOUT Threshold, 3-phase, Drop Phase 3
VMODE = 1.8V
1.175
1.2
1.225
V
2.5
2.7
50
V
µA
V
PHASE ADDING/DROPPING
VIOUT Threshold Hysteresis
Phase Drop Disable Threshold at MODE pin
3.5
µA
V
40
mV
4
V
160
µA
CURRENT SENSE AND OVERCURRENT PROTECTION
Peak Current Limit for Individual Channel
IOUT Current Tolerance
IISENxP = 60µA, 4-phase
173
Maximum Voltage Limit at IOUT Pin
187
200
2.0
µA
V
DMAX PIN
DMAX Threshold, High
3
V
DMAX Threshold, Low
2
V
DMAX Floating Voltage
During Phase Detection Time (t3 on Figure 14)
2.5
V
Max Duty Cycle, DMAX = GND
VCOMP = 3.5V
91.7
%
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8
FN7909.4
September 5, 2014
ISL78225
Electrical Specifications
Operating Conditions: VIN = 12V, TA = -40°C to +125°C, unless otherwise specified. Typical specifications are
at TA = +25°C. Boldface limits apply across the operating temperature range, -40°C to +125°C. (Continued)
PARAMETER
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNITS
Max Duty Cycle, DMAX = FLOAT
VCOMP = 3.5V
75
Max Duty Cycle, DMAX = VCC
VCOMP = 3.5V
83.3
%
DMAX Source/Sink Current
During t3 on Figure 14
50
µA
DMAX Source/Sink Current
After t3 on Figure 14
-1
%
1
µA
PWM_TRI, PWM_INV, SYNC PIN DIGITAL LOGIC
Input Leakage Current
EN < 1V
Input Pull-Down Current
EN > 2V, Pin Voltage = 2.1V
-1
0.4
Logic Input Low
Logic Input High
1
µA
1.5
µA
0.8
V
2
V
4.5
V
DRIVE_EN, CLK_OUT PIN
Output High Voltage
IDRIVE_EN = 500µA
Output Low Voltage
IDRIVE_EN = -500µA
0.5
V
VOUT SENSE PIN
Input Leakage Current
-1
Threshold Voltage
2.325
2.4
1
µA
2.475
V
1
µA
2.475
V
VIN SENSE PIN
Input Leakage Current
-1
Threshold Voltage
2.325
Hysteresis
2.4
110
mV
VOUT_OVB, VIN_OVB PIN
Leakage Current
VPIN = HIGH
1
µA
Low Voltage
IPIN = 0.5mA
0.2
V
1
µA
0.2
V
POWER GOOD MONITOR PIN
PGOOD Leakage Current
PGOOD = HIGH
PGOOD Low Voltage
IPGOOD = 0.5mA
Overvoltage Rising Trip Point
VFB/VREF, VREF2 > 2.5V
Overvoltage Rising Hysteresis
VFB/VREF, VREF2 > 2.5V
Undervoltage Rising Trip Point
VFB/VREF, VREF2 > 2.5V
Undervoltage Rising Hysteresis
VFB/VREF, VREF2 > 2.5V
117
120
123
5
77
80
5
%
%
83
%
%
OVER-TEMPERATURE PROTECTION
Over-Temperature Trip Point
160
°C
Over-Temperature Recovery Threshold
145
°C
NOTES:
6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise noted. Compliance to datasheet limits is assured by one or
more methods: production test, characterization and/or design.
7. Please refer to LDO current derating curve in “Internal 5V LDO Output Current Limit Derating Curves” on page 19 for IMAX vs VIN.
8. TS = Switching Period = 1/(switching frequency).
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FN7909.4
September 5, 2014
ISL78225
Typical Performance Curves
0.98
0.93
0.88
0.83
WITHOUT PHASE DROPPING
0.88
0.83
0.78
0.78
0.73
WITH PHASE DROPPING
0.93
WITHOUT PHASE DROPPING
EFFICIENCY (%)
EFFICIENCY (%)
0.98
WITH PHASE DROPPING
10V INPUT, 36V OUTPUT
SYNCHRONOUS BOOST
0
1
2
3
4
5
6
7
8
9
0.73
10
16V INPUT, 36V OUTPUT
SYNCHRONOUS BOOST
0
1
2
3
4
5
6
7
8
9
10
OUTPUT CURRENT (A)
FIGURE 2. 10V INPUT EFFICIENCY vs OUTPUT CURRENT vs PHASE
DROPPING MODE
FIGURE 3. 16V INPUT EFFICIENCY vs OUTPUT CURRENT vs PHASE
DROPPING MODE
36.5
36.5
36.4
36.4
36.3
36.3
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
OUTPUT CURRENT (A)
36.2
36.1
36.0
35.9
35.8
35.7
35.6
35.5
1
2
3
4
5
6
7
8
9
36.1
36.0
35.9
35.8
35.7
35.6
10V INPUT
0
36.2
10
35.5
10A OUTPUT
10
11
OUTPUT CURRENT (A)
FIGURE 4. OUTPUT VOLTAGE vs OUTPUT CURRENT
12
13
14
15
16
INPUT VOLTAGE (V)
FIGURE 5. OUTPUT VOLTAGE vs INPUT VOLTAGE
C1 = PHASE 1, 20V/DIV
C1 = PHASE 1, 20V/DIV
C4 = VOUT (AC-COUPLED), 100mV/DIV
C4 = VOUT (AC-COUPLED), 100mV/DIV
10V INPUT, 36V AT 10A OUTPUT
1µs/DIV
FIGURE 6. FULL LOAD OUTPUT RIPPLE
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10
10V INPUT, 0 TO 10A TO 0 STEP LOAD
2ms/DIV
FIGURE 7. FULL STEP LOAD TRANSIENT
FN7909.4
September 5, 2014
ISL78225
Typical Performance Curves
(Continued)
C1 = PWM1, 5V/DIV
C1 = PWM1, 5V/DIV
C2 = PWM2, 5V/DIV
C2 = PWM2, 5V/DIV
C3 = PWM3, 5V/DIV
C3 = PWM3, 5V/DIV
C4 = CLK_OUT, 5V/DIV
C4 = CLK_OUT, 5V/DIV
1µs/DIV
1µs/DIV
FIGURE 8. WAVEFORMS WITH PWM_INV = GND
FIGURE 9. WAVEFORMS WITH PWM_INV = VCC
C1 = PWM1, 5V/DIV
C1 = EN, 2V/DIV
C3 = PWM3, 5V/DIV
C2 = VCC, 5V/DIV
C4 = PWM4, 5V/DIV
C3 = PGOOD, 5V/DIV
10V INPUT, 36V AT 1A OUTPUT
C4 = VOUT, 20V/DIV
C2 = IL1, 5A/DIV
16V INPUT, 36V AT 10A OUTPUT
1µs/DIV
1µs/DIV
FIGURE 10. FULL LOAD WAVEFORMS
FIGURE 11. ENABLE/DISABLE WAVEFORMS
10V INPUT, 10A OUTPUT
C1 = VREF2, 1V/DIV
C4 = VOUT, 20V/DIV
20ms/DIV
FIGURE 12. MODULATING VREF2 INPUT
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11
FN7909.4
September 5, 2014
ISL78225
Operation Description
Multiphase Power Conversion
IL1 + IL2 + IL3
The technical challenges associated with producing a
single-phase converter that is both cost-effective and thermally
viable for high power applications have forced a change to the
cost-saving approach of multiphase solution. The ISL78225
controller helps reduce the complexity of implementation by
integrating vital functions and requiring minimal output
components.
IL3
PWM3
IL2
PWM2
Interleaving
IL1
The switching of each channel in a multiphase converter is timed
to be symmetrically out-of-phase with each of the other channels.
Take a 3-phase converter for example; each channel switches
1/3 cycle after the previous channel and 1/3 cycle before the
following channel. As a result, the three-phase converter has a
combined ripple frequency three times greater than the ripple
frequency of any one phase. In addition, the peak-to-peak
amplitude of the combined inductor current is reduced in
proportion to the number of phases (Equations 1 and 2). The
increased ripple frequency and the lower ripple amplitude mean
that the designer can use less per-channel inductance and lower
total input and output capacitance for any performance
specification.
Figure 13 illustrates the multiplicative effect on input ripple
current. The three channel currents (IL1, IL2, and IL3) combine to
form the AC ripple current and the DC input current. The ripple
component has three times the ripple frequency of each
individual channel current. Each PWM pulse is triggered 1/3 of a
cycle after the start of the PWM pulse of the previous phase.
To understand the reduction of the ripple current amplitude in the
multiphase circuit, examine the equation representing an
individual channel’s peak-to-peak inductor current.
In Equation 1, VIN and VOUT are the input and the output voltages
respectively, L is the single-channel inductor value, and fSW is the
switching frequency.
 V OUT – V IN  V IN
I P-P = ----------------------------------------------L f SW V
(EQ. 1)
OUT
The input capacitors conduct the ripple component of the
inductor current. In the case of multiphase converters, the
capacitor current is the sum of the ripple currents from each of
the individual channels. Compare Equation 1 to the expression for
the peak-to-peak current after the summation of N symmetrically
phase-shifted inductor currents in Equation 2. Peak-to-peak
ripple current decreases by an amount proportional to the
number of channels. Reducing the inductor ripple current allows
the designer to use fewer or less costly input capacitors.
 V OUT – N V IN  V IN
I C  P – P  = ----------------------------------------------------L f SW V
(EQ. 2)
OUT
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PWM1
TIME
FIGURE 13. PWM AND INDUCTOR-CURRENT WAVEFORMS FOR
3-PHASE CONVERTER
PWM Operations
The timing of each channel is set by the total number of active
channels. The default channel setting for the ISL78225 is 4, and
the switching cycle is defined as the time between PWM pulse
initiation signals of each channel. The cycle time of the pulse
initiation signal is the inversion of the switching frequency set by
the resistor between the FS pin and ground. The PWM signals
command the MOSFET drivers to turn on/off the channel
MOSFETs. Normal operation assumes PWM_INV is tied to GND
and inverting MOSFET drivers are used.
In the default 4-phase operation, the PWM2 pulse starts 1/4 of a
cycle after PWM1, the PWM3 pulse starts 1/4 of a cycle after
PWM2, and the PWM4 pulse starts 1/4 of a cycle after PWM3.
Phase Selection
The ISL78225 can work in 1, 2, 3 or 4-phase configuration.
Connecting the PWM4 to VCC selects 3-phase operation and the
pulse times are spaced in 1/3 cycle increments. Connecting the
PWM3 to VCC selects 2-phase operation and the pulse times are
spaced in 1/2 cycle increments. For the unused ISENxN and
ISENxP, a 1k resistor is recommended to connect ISENxN and
ISENxP, and connect ISENxN to VIN.
Modes of Operation
The different modes of operation will be determined by the
voltage combinations of the MODE pin and the PWM_TRI pin.
If automatic phase adding/dropping function is not needed, the
MODE pin should be tied to VCC (Logic HIGH). If higher light-load
efficiency is preferred, phase adding/dropping function could be
implemented by connecting the MODE pin through a resistor to
GND. A 5µA reference current will flow out of the MODE pin to
generate corresponding VMODE. VMODE is used to compare with
VIOUT to determine the phase adding/dropping level.
When PWM_TRI is tied to GND (Logic LOW), the PWM outputs will
be 2-levels (i.e., 0V and 5V). When PWM_TRI is pulled to VCC
(Logic HIGH), apart from generating the 0V and 5V PWM signals,
the PWM outputs can also generate 2.5V tri-level signal. The
external driver can identify this tri-level signal and turn off both
low side and high side output signals accordingly.
12
FN7909.4
September 5, 2014
ISL78225
The truth table regarding VMODE and VPWM_TRI for different
modes of applications is summarized in Table 1.
Figure 14 shows the ISL78225 internal circuit functions before
the soft-start begins.
TABLE 1. OPERATION MODE FOR DIFFERENT APPLICATIONS
EXTERNAL
DRIVER
IDENTIFY
2.5V TRI-LEVEL
SIGNAL?
CASE MODE PWM _TRI
A
B
C
D
1
Analog
1
Analog
1
1
0
0
Yes
Yes
No
No
EN
APPLICATIONS
Synchronous boost for audio
amplifier power supply. No
phase dropping. Forced
minimum ON pulses exists.
Applications that need
improving light load efficiency
(automatic phase dropping +
cycle-by-cycle diode emulation
+ pulse skipping).
Applications that the external
driver cannot identify tri-level
signal; no phase dropping.
Applications that the external
driver cannot identify tri-level
signal, with improved light load
efficiency (e.g., 6-phase
non synchronous boost with
phase dropping).
Considerations for Audio
Amplifier Power Supply
Application
For multiphase boost converters used in audio amplifier
applications, it is preferred to have the following features:
1. Automatic phase dropping function is NOT needed because
the load is fast changing.
2. In car audio amplifier applications, the switching frequency is
preferred to be fixed, such that it will not interfere with
FM/AM band.
3. For synchronous boost, diode emulation is needed during
start-up in order to prevent negative current dumping to the
input side.
4. For synchronous boost, a maximum duty cycle limitation on
the synchronous FET is preferred.
Based on the above mentioned “preferred features”, for audio
amplifier applications, it does not need phase dropping/adding,
but it needs a tri-state PWM signal if synchronous boost structure
is used. Also, in order to limit the maximum duty cycle of the
synchronous FET, the minimal turn on time of the active FET (Low
side FET for boost structure) will be changed from fixed 130ns to
variable time, which is 1/12 of the switching periods.
Operation Initialization Before
Soft-Start
Prior to converter initialization, proper conditions must exist on the
enable inputs (EN pin) and VCC pin. When both conditions are met,
the controller begins soft-start. Once the output voltage is within
the proper window of operation, VPGOOD is asserted logic high.
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13
CIRCUIT INITIALIZATION BEFORE SOFT- START
t
0
VCC
POR
t
0
t1
t2
t3
t4
THEN SOFT- START BEGINS
t5
PWM_DETECTION
t
0
PWM
0
t
FIGURE 14. CIRCUIT INITIALIZATION BEFORE SOFT-START
Figure 14 shows there are 5x intervals before the soft-start is
initialized. They are specified as t1, t2, t3, t4 and t5, respectively.
The descriptions for each time interval are as follows:
Time t1: The enable comparator holds the ISL78225 in shutdown
until the VEN rises above 1.2V at the beginning of t1 time period.
During t1, VVCC will gradually increase until it reaches the internal
power-on reset (POR) rising threshold. Then the system enters t2.
Time t2: During t2 time, the device initialization occurs. The time
duration for t2 is typically from 60µs to 100µs.
Time t3: After the self-calibration finishes, the internal PWM
detection signal will be asserted and the system enters the t3
period. During t3, the ISL78225 will detect the voltage on each
PWM pin to determine the active phase number. If PWM1 or
PWM2 is accidentally pulled to VCC, the chip will be latched off
and wait for power recycling. The time duration for t3 is fixed to
around 30µs.
Time t4: When the internal PWM detection signal is released, the
system enters t4 period. During t4 period, the ISL78225 will wait
until the internal PLL circuits are locked to the preset oscillator
frequency. When PLL locking is achieved, the oscillator will
generate output at the CLK_OUT pin. The time duration for t4 is
typically around 0.5ms, depending on the PLL_COMP pin
configuration.
Time t5: After the PLL locks the frequency, the system enters the
t5 period. During t5, the PWM outputs are held in a
high-impedance state (If VPWM_TRI = 1) or logic low (if
VPWM_TRI = 0), and the VDRIVE_EN is logic LOW to assure the
external drivers remain off. The ISL78225 has one unique
feature to prebias the VSS based on VFB information during this
time. The duration time for t5 is around 50µs.
After t5, the soft-start process will begin. The following section
will discuss the soft-start process in detail for different
applications.
FN7909.4
September 5, 2014
ISL78225
Soft-Start Process for Different
Modes (Refer to Table 1)
At the beginning of soft-start, the SS pin voltage will start
ramping up from a voltage equal to the FB voltage. The soft-start
period ends when the SS pin voltage reaches the lower
power-good threshold that is 80% of the lower value of VREF2
or 2V.
Case A (VMODE = VCC, VPWM_TRI = VCC)
Figure 15 shows the prebias start-up PWM waveform for Case A
in Table 1. The VPWM_TRI = VCC so the PWM can output a tri-level
signal, which the external drivers need to identify, and
VMODE = VCC to ban the automatic phase dropping function.
Time t4, t5: Same as the t4, t5 in Figure 14, soft-start has not
started yet. See “Operation Initialization Before Soft-Start” on
page 13 for a detailed description.
Time t6: At the beginning of t6, the SS pin has already been
pre-biased to a value very close to the VFB, so that the internal
reference signal will start from the voltage close to the FB pin.
This scheme will eliminate the internal delay for a non prebiased
application.
The DRIVE_EN pin, which is connected to the enable pins of the
external drivers, will be pulled high when first PWM toggles at the
beginning of t6; as a result external drivers will start working. The
PWM signals will switch between tri-level and low. The driver will
only turn on the lower MOSFET accordingly, and the duty cycle
will increase gradually from 0 to steady state. The synchronous
MOSFET (Upper FET for Boost converter) will never turn on during
this time, so diode emulation can be achieved during the start-up
and in turn prevent negative current flowing from output to input.
Time t7: Soft-start finishes at the beginning of t7. The PWMs will
change to a 2-level 0V to 5V switching signal and the
synchronous MOSFET will be turned on.
SOFT-START WAVEFORM (CASE A)
V
Vfb
0
V
5V
SEE (NOTE)
Vref
t4 t5
t6
DIODE EMULATION
t7
SYNCHRONOUS
OPERATION
LOWER FET TURN ON
(PWM_INV = 0)
2.5V
PWM
0
DIODE EMULATION
SYNCHRONOUS
OPERATION
5V
(PWM_INV = 1)
2.5V
PWM
0
5V
DRIVE_EN
0
NOTE: t4, t5 PERIOD ARE FROM FIGURE 5
FIGURE 15. SOFT-START WAVEFORM (CASE A)
Case B (VMODE < 4V, VPWM_TRI = VCC, Light
Load Condition)
The only difference between Case A and Case B start-up
waveforms is that at light load, Case B can drop phases and have
cycle-by-cycle diode emulation at PWM1 and PWM2.
For Case B applications, where good light-load efficiency is
always preferred, the ISL78225 provides three light-load
efficiency enhancement methods. When the load current
reduces, the ISL78225 will first assert the automatic phase
dropping function to reduce the active phase number according
to the load level. The minimum active phase number is two. If the
load current further reduces even when running at two-phase
operation, the ISL78225 will assert a second method by utilizing
cycle-by-cycle diode emulation. During this time the IC will sense
the inductor current, and when the current is approximately zero,
it will turn off the synchronous MOSFET. If the load current is
further reduced to deep light load operation, pulse skipping
function will kick in to optimize the overall efficiency.
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FN7909.4
September 5, 2014
ISL78225
SOFT-START WAVEFORM (CASE B, LIGHT LOAD)
V
Vfb
Vfb
0
V
5V
Vref
t4 t5
SOFT-START WAVEFORM (CASE C)
V
SEE (NOTE)
0
t7
t6
DIODE EMULATION
SYNCHRONOUS
OPERATION WITH
CYCLE-BY-CYCLE
DIODE EMULATION
SEE (NOTE)
t6
V
5V
(PWM_INV = 0)
LOWER FET TURN ON
Vref
t4 t5
LOWER FET TURN ON
(PWM_INV = 0)
PWM
2.5V
PWM
0
0
DIODE EMULATION
5V
SYNCHRONOUS
OPERATION WITH
CYCLE-BY-CYCLE
DIODE EMULATION
5V
(PWM_INV = 1)
(PWM_INV = 1)
PWM
2.5V
PWM
0
0
5V
5V
DRIVE_EN
DRIVE_EN
0
0
NOTE: t4, t5 PERIOD ARE FROM FIGURE 5
FIGURE 16. SOFT-START WAVEFORM (CASE B, LIGHT LOAD)
NOTE: t4, t5 PERIOD ARE FROM FIGURE 5
FIGURE 17. SOFT-START WAVEFORM (CASE C, LIGHT LOAD)
Case C (VPWM_TRI = 0)
Soft-Start Ramp Slew Rate
Calculation
Time t4, t5: Same as the t4, t5 in Figure 14, soft-start has not
started yet; see “Operation Initialization Before Soft-Start” on
page 13 for detailed description.
Time t6: At the beginning of t6, the PWM signal will start to
switch between 0V and 5V. The driver will turn on the lower and
upper MOSFETs accordingly, and the duty cycle for lower MOSFET
will increase gradually from 0 to steady state. DRIVE_EN will be
pulled high when the first PWM toggles at the beginning of t6 to
enable the external drivers.
The soft-start ramp slew rate SRSS is determined by the capacitor
value CSS from SS pin to GND. CSS can be calculated based on
Equation 3:
– 12
V
5X10
SR SS = -----------------------  ------
C SS  s
(EQ. 3)
Figure 18 shows the relationship between CSS and SRSS.
5.0
SOFT-START SLEW RATE (V/ms)
For applications that the driver cannot identify a tri-state PWM
signal, the VPWM_TRI should be connected to GND (Logic LOW),
such that the PWM signal will only be 2 levels between 0V and
5V. Then the DRIVE_EN pin can be connected to the EN pin of the
external drivers. DRIVE_EN will be asserted when the PWM first
toggles such that the prebias start-up capability can be achieved.
Detailed soft-start for Case C is shown in Figure 17.
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
1
10
100
CSS (nF)
FIGURE 18. SOFT-START CAPACITOR vs SLEW RATE
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15
FN7909.4
September 5, 2014
ISL78225
Oscillator and Synchronization
Current Sensing
The switching frequency is determined by the selection of the
frequency-setting resistor, RFS, connected from the FS pin to
GND. Equation 4 is provided to assist in selecting the correct
resistor value.
The ISL78225 senses the current continuously for fast response.
It supports both sense resistor and inductor DCR current sensing
methods. The sensed current for each active channel will be used
for loop control, phase current balance, individual channel
overcurrent protection and total average current protection. The
internal circuitry, (shown in Figures 20 and 21), represents a single
channel. This circuitry is repeated for each channel, but may not
be active depending on the status of the PWM3 and PWM4 pin
voltage.
R FS = 4X10
10  1
–8
---------- – 5X10 
f

(EQ. 4)
SW
Where fSW is the switching frequency of each phase. Figure 19
shows the relationship between RFS and switching frequency.
Peak current mode control is implemented by feeding back the
current output of the current sense amplifier (CSA) to the
regulator control loop. Individual channel peak current limit is
implemented by comparing the CSA output current with 160µA.
When the peak current limit comparator is tripped, the PWM
ON-pulse is terminated and the IC is latched off.
1000
900
800
fSW (kHz)
700
600
500
Sense Resistor Current Sensing
400
A sense resistor can be placed in series with the power inductor.
As shown in Figure 20, The ISL78225 acquires the channel
current information by sensing the voltage signal across the
sense resistor. Because the voltage on both the positive input
and the negative input of the current sense amplifier (CSA) are
forced to be equal, the voltage across RSET is equivalent to the
voltage drop across the RSEN resistor. The resulting current into
the ISENxP pin is proportional to the channel current, IL.
Equation 5 for ISEN is derived where IL is the channel current:
300
200
100
0
0
100
200
300
400
500
600
RFS (k)
FIGURE 19. RFS vs SWITCHING FREQUENCY
The maximum frequency at each PWM output is 1MHz. If the FS
pin is accidentally shorted to GND or connected to a low
impedance node, the internal circuits will detect this fault
condition and fold back the switching frequency to the 75kHz
minimal value.
The ISL78225 contains a phase lock loop (PLL) circuit and has
frequency synchronization capability by simply connecting the
SYNC pin to an external square pulse waveform (typically 20% to
80% duty cycle). In normal operation, the external SYNC
frequency needs to be at least 20% faster than the internal
oscillator frequency setting. The ISL78225 will synchronize its
switching frequency to the fundamental frequency of the input
waveform. The frequency synchronization feature will
synchronize the rising edge of the PWM1 clock signal with the
rising edge of the external clock signal at the SYNC pin.
The PLL is compensated with a series resistor-capacitor (Rc and
Cc) from the PLL_COMP pin to GND and a capacitor (Cp) from
PLL_COMP to GND. Typical values are Rc = 6.8kΩ, Cc = 6.8nF,
Cp = 1nF. The typical lock time is around 0.5ms.
The CLK_OUT pin provides a square pulse waveform at the
switching frequency. The amplitude is 5V with approximately
40% positive duty cycle, and the rising edge is synchronized with
the leading edge of PWM1.
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16
R SEN
I SEN = I L  ----------------R ISET
(EQ. 5)
VIN
VOUT
RSEN
ISEN
L
RSET
SENSE RESISTOR
CURRENT SENSING
ISEN
CSA
ISEN(n)P
ISEN(n)N
ISL78225 INTERNAL CIRCUITS
FIGURE 20. SENSE RESISTOR CURRENT SENSING
FN7909.4
September 5, 2014
ISL78225
Inductor DCR Sensing
An inductor’s winding is characteristic of a distributed resistance
as measured by the DCR (Direct Current Resistance) parameter.
IL
VIN
DCR
C
ISEN
L
VOUT
R
Adjustable Automatic Phase
Dropping/Adding at Light Load Condition
RSET
INDUCTOR DCR
CURRENT SENSING
ISEN
CSA
ISEN(n)P
ISEN(n)N
ISL78225 INTERNAL CIRCUITS
FIGURE 21. INDUCTOR DCR CURRENT SENSING
Consider the inductor DCR as a separate lumped quantity, as
shown in Figure 21. The channel current IL, flowing through the
inductor, will also pass through the DCR. Equation 6 shows the
s-domain equivalent voltage across the inductor VL.
V L = I L   s  L + DCR 
(EQ. 6)
A simple R-C network across the inductor extracts the DCR
voltage, as shown in Figure 21.
The voltage on the capacitor VC, can be shown to be proportional
to the channel current IL, see Equation 7.
L
 s  ------------+ 1   DCR  I L 
 DCR

V C = -------------------------------------------------------------------- s  RC + 1 
(EQ. 7)
If the R-C network components are selected such that the RC
time constant (= R*C) matches the inductor time constant
(= L/DCR), the voltage across the capacitor VC is equal to the
voltage drop across the DCR, i.e., proportional to the channel
current.
With the internal low-offset differential current sense amplifier,
the capacitor voltage VC is replicated across the sense resistor
RSET. Therefore, the current flow into the ISENxP pin is
proportional to the inductor current. Equation 8 shows that the
ratio of the channel current to the sensed current ISEN is driven
by the value of the sense resistor and the DCR of the inductor.
DCR
I SEN = I L  --------------R
(EQ. 8)
SET
Light Load Efficiency
Enhancement Schemes
For switching mode power supplies, the total loss is related to
both the conduction loss and the switching loss. At heavy load,
the conduction loss is dominant while the switching loss will take
charge at light load condition. So, if a multiphase converter is
running at a fixed phase number for the entire load range, we will
observe that below a certain load point, the total efficiency starts
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17
to drop heavily. The ISL78225 has automatic phase dropping,
cycle-by-cycle diode emulation and pulse skipping features to
enhance the light load efficiency. By observing the total input
current on-the-fly and dropping the active phase numbers
accordingly, the overall system can achieve optimized efficiency
over the entire load range. All the previously mentioned light load
enhancement features can be disabled by simply pulling the
MODE pin to VCC.
If the MODE pin is connected to a resistor to GND, and the voltage
on the MODE pin is lower than its disable threshold 4V, the
adjustable automatic phase dropping/adding mode will be
enabled. When the ISL78225 controller works in this mode, it
will automatically adjust the active phase number by comparing
the VMODE and VIOUT, which represents sensed total current
information. The VMODE sets the overall phase dropping
threshold, and the VIOUT is proportional to the input current,
which is in turn proportional to the load current. The smaller the
load current, the lower the voltage observed on the IOUT pin, and
the ISL78225 will drop phases in operation. Once the MODE pin
voltage is fixed, the threshold to determine how many phases are
in operation is dependent on two factors:
1. The maximum configured phase number.
2. The voltage on the IOUT pin (VIOUT).
For example, if the converter is working in 4-phase operation and
the MODE pin is set to 1.2V, the converter will monitor the VIOUT
and compared to 1.2V; if less than 900mV (75% of 1.2V), it will
drop to 3-phase; if less than 600mV (50% of 1.2V), it will drop to
2-phase. The detailed threshold setting is shown in the “Electrical
Specifications” on page 7.
If PWM_TRI is tied to VCC, the dropped phase will provide a 2.5V
tri-level signal at its PWM output. The external driver has to
identify this tri-state signal and turn off both the lower and upper
switches accordingly. For better transient response during phase
dropping, the ISL78225 will gradually reduce the duty cycle of
the phase from steady state to zero, typically within 15 switching
cycles. This gradual dropping scheme will help smooth the
change of the PWM signal and, in turn, will help to stabilize the
system when phase dropping happens.
The ISL78225 also has an automatic phase adding feature
similar to phase dropping, but when doing phase adding there
will not be 15 switching cycles gradually adding. It will add
phases instantly to take care of the increased load condition. The
phase adding scheme is controlled by three factors.
1. The maximum configured phase number
2. The voltage on the IOUT pin (VIOUT).
3. Individual phase current
Factors 1 and 2 are similar to the phase dropping scheme. If the
VIOUT is higher than the phase dropping threshold plus the
hysteresis voltage, the dropped phase will be added back one by
one instantly.
The previously mentioned phase-adding method can take care of
the condition that the load current increases slowly. However, if
the load is increasing quickly, the IC will use a different phase
adding scheme. The ISL78225 monitors the individual channel
FN7909.4
September 5, 2014
ISL78225
current for all active phases. During phase adding, the system
will bring down the preset channel current limit to 2/3 of its
original value (160µA). If any of the phase’s sensed current hit
the 2/3 of preset channel current limit threshold (i.e., 106.7µA),
all the phases will be added back instantly. After a fixed 1.5ms
delay, the phase dropping circuit will be activated and the system
will react to drop the phase number to the correct value.
Fault Monitoring and Protection
During phase adding, when either phase hits the preset channel
current limit, there will be 200µs blanking time such that
per-channel OCP will not be triggered during this blanking time.
PGOOD Signal
Diode Emulation at Very Light Load Condition
When phase dropping is asserted and the minimum phase
operation is 2 phases, if the load is still reducing and
synchronous boost structure is used, the ISL78225 controller will
enter into forced cycle-by-cycle diode emulation mode. The PWM
output will be tri-stated when the inductor current falls to zero,
such that the synchronous MOSFET can be turned off accordingly
cycle-by-cycle for forced diode emulation. This cycle-by-cycle
diode emulation scheme will only be asserted when two
conditions are met:
1. The PWM_TRI pin voltage is logic HIGH.
2. Only two phases are running either by phase dropping or
initial configuration.
By utilizing the cycle-by-cycle diode emulation scheme in this
way, negative current is prevented and the system can still
optimize the efficiency even at very light load conditions.
Pulse Skipping at Deep Light Load Condition
If the converter enters diode emulation mode and the load is still
reducing, eventually pulse skipping will occur to increase the
deep light-load efficiency.
Adjustable Slope Compensation
For a boost converter working in current mode control, slope
compensation is needed when steady state duty cycle is larger
than 50%. When slope compensation is too low, the converter
can suffer from jitter or oscillation. On the other hand, over
compensation of the slope will cause the reduction of the phase
margin. Therefore, proper design of the slope compensation is
needed.
The ISL78225 features adjustable slope compensation by
setting the resistor value RSLOPE from the SLOPE pin to GND.
This function will ease the compensation design and provide
more flexibility in choosing the external components.
For current mode control, typically we need the compensation
slope mA to be 50% of the inductor current down ramp slope mB
when the lower MOSFET is off. Equation 9 shows how to choose
the suitable resistor value.
6
1.136x10 xLxR SET
R SLOPE = ----------------------------------------------------------   
 V OUT – V IN   R SEN 
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(EQ. 9)
The ISL78225 actively monitors input/output voltage and current to
detect fault conditions. Fault monitors trigger protective measures
to prevent damage to the load. Common power-good indicator pin
(PGOOD pin) and VIN_OVB, VOUT_OVB pins are provided for linking
to external system monitors.
The PGOOD pin is an open-drain logic output to indicate that the
soft-start period is completed and the output voltage is within the
specified range. This pin is pulled low during soft-start and
releases high after a successful soft-start. PGOOD will be pulled
low when a UV/OV/OC/OT fault occurs.
Input Overvoltage Detection
The ISL78225 utilizes VIN_SEN and VIN_OVB pins to deal with a
high input voltage. The VIN_SEN pin is used for sensing the input
voltage. A resistor divider network is connected between this pin
and the boost power stage input voltage rail. When the voltage
on VIN_SEN is higher than 2.4V, the open-drain output VIN_OVB
pin will be pulled low to indicate an input overvoltage condition,
The VIN overvoltage sensing threshold can be programmed by
changing the resistor values, and hysteresis voltage of the
internal comparator is fixed to be 100mV.
Output Undervoltage Detection
The undervoltage threshold is set at 80% of the internal voltage
reference. When the output voltage at the FB pin is below the
undervoltage threshold minus the hysteresis, PGOOD is pulled
low. When the output voltage comes back to 80% of the
reference voltage, PGOOD will return back to high.
Output Overvoltage Detection/Protection
The ISL78225 overvoltage detection circuit is active after time t2
in Figure 14 on page 13. The OV trip point is set to 120% of the
internal reference level. Once an overvoltage condition is
detected, the PGOOD will be pulled low but the controller will
continue to operate.
The ISL78225 also provides the flexibility for output overvoltage
protection by utilizing the VOUT_SEN and VOUT_OVB pins. The
VOUT_SEN pin is used for sensing the output voltage. A resistor
divider network is connected between this pin and the boost
power stage output voltage rail. When the voltage on VOUT_SEN
is higher than 2.4V, the open-drain output VOUT_OVB will be
pulled low, and the ISL78225 IC will be latched off to indicate an
output overvoltage condition. The VOUT overvoltage sensing
threshold can be programmed by changing the resistor values.
Overcurrent Protection
ISL78225 has two levels of overcurrent protection. Each phase is
protected from an overcurrent condition by limiting its peak
current, and the combined total current is protected on an
average basis.
For the individual channel overcurrent protection, the ISL78225
continuously compares the CSA output current of each channel
with a 160µA reference current. If any channel’s current trips the
current limit comparator, the ISL78225 will be shut down.
FN7909.4
September 5, 2014
ISL78225
However, during the phase adding period, the individual channel
current protection function will be blanked for 200µs, in order to
give other phases the chance to take care of the current.
R SEN
V IOUT = 0.75I IN ---------------- R IOUT
R
(EQ. 10)
SET
When the VIOUT is higher than 2V for a consecutive 100µs, the
ISL78225 IC will be triggered to shut down. This provides
additional safety for the voltage regulator.
Equation 11 can be used to calculate the value of the resistor RIOUT
based on the desired OCP level IAVG, OCP2.
2
R IOUT = ------------------------------I AVG OCP2
(EQ. 11)
The total average overcurrent protection scheme will not be
asserted until the soft-start pin voltage VSS reaches its clamped
value (approximately 3.5V). During the soft-start time, the system
does not latch-off if per-channel or overall OC limit is reached.
Instead, the individual channel current will run at its preset peak
current limit level.
Thermal Protection
The ISL78225 will be disabled if the die junction temperature
reaches a nominal of +160°C. It will recover when the junction
temperature falls below a +15°C hysteresis. The +15°C
hysteresis insures that the device will not be re-enabled until the
junction temperature has dropped to below about +145°C.
Internal 5V LDO Output Current
Limit Derating Curves
ISL78225 contains an internal 5V/200mA LDO, and the input of
LDO (VIN pin) can go as high as 40V. Based on the junction to
ambient thermal resistance RJA of the package, we need to
guarantee that the maximum junction temperature should be
below +125°C TMAX. Figure 22 shows the relationship between
maximum allowed LDO output current and input voltage. The curve
is based on +35°C/W thermal resistance RJA for the package.
Each curve represents different ambient temperature, TA.
TA = +25°C
180
160
ILDO(MAX) (mA)
The IOUT pin serves for both input current monitoring and total
average current OCP functions. The CSA output current for each
channel is scaled and summed together at this pin. An RC
network should be connected between the IOUT pin and GND,
such that the ripple current signal can be filtered out and
converted to a voltage signal to represent the averaged total
input current. The relationship between total input current IIN and
VIOUT can be calculated as Equation 10 (see Figure 20 on
page 16 for RSEN and RSET positions):
200
TA = +50°C
140
TA = +75°C
120
100
80
60
40
TA = +100°C
20
0
6
8 10 12 14 16 18 20 22 24 26 28 30 32 34 36 38 40
VIN (V)
FIGURE 22. ILDO(MAX) vs VIN
Dedicated VREF2 Pin for Input
Voltage Tracking
A second reference input pin, VREF2, is added to the input of the
transconductance amplifier. The ISL78225 internal reference will
automatically change to VREF2 when it is pulled below 1.8V. The
VREF2 pin can be connected to VIN through resistor network to
implement the automatic input voltage tracking function. This
function is very useful under car battery voltage cranking
conditions (such as when the car is parked and the driver is
listening to the stereo), where the full load power is typically not
needed. In this case, the ISL78225 can limit the output power by
allowing the output voltage to track the input voltage. If VREF2 is
not used, the pin should be connected to VCC.
Configurations for Dual IC
Operations
For high power applications, two ISL78225 ICs can be easily
configured to support 8-phase operation. The IC that provides the
CLK_OUT signal is called master IC, and the IC that received the
CLK_OUT signal is called slave IC. Note that the two PWM1
signals are synchronized and the net effect is 4-phase operation
with double the output current.
SYSTEM
DRIVE_EN
DRIVE_EN
DRIVE_EN
CLK_OUT
SYNC
MASTER IC
COMP
FB
SLAVE IC
SS
COMP
FB
SS
FIGURE 23. CONFIGURATIONS FOR DUAL IC OPERATION
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FN7909.4
September 5, 2014
ISL78225
Figure 23 shows the step-by-step setup as follows:
1. Connect the CLK_OUT pin of the master IC to the SYNC pin of
the slave IC.
2. Set the master IC’s switching frequency as desired frequency;
set the slave IC’s switching frequency 20% below the master IC’s.
3. Connect both IC’s COMP, SS and FB pins together.
4. Both IC’s DRIVE_EN pin should be ANDed together to provide
the system’s driver enable signal.
5. Since PGOOD, VOUT_OVB and VIN_OVB pins are open-drain
structure, both IC’s PGOOD, VOUT_OVB and VIN_OVB pins can
be tied together and use one pull-up resistor to connect to VCC.
6. If Phase dropping function is needed, tie both IC’s IOUT and
MODE pins together.
ISL78225EVAL1Z Evaluation
Board
The ISL78225EVAL1Z evaluation board is designed for
automotive trunk audio application to deliver 36V at 10A over a
10 to 16V input range. The ISL78420 100V inverting MOSFET
drivers with tri-state input capability facilitate synchronous boost
rectification and phase-dropping for high efficiency over the
entire load range. See AN1727 on the Intersil website,
www.intersil.com, for more information on ISL78225EVAL1Z.
Submit Document Feedback
20
FN7909.4
September 5, 2014
ISL78225
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you
have the latest revision.
DATE
REVISION
CHANGE
September 5, 2014 FN7909.4 Page 7- Changed the spec reference for Charge Device Model from “JESD22-C101C” to “AEC-Q100-11”
Page 21- Updated the “About Intersil” verbiage.
February 11, 2014
FN7909.3 Page 21
- 2nd line of the disclaimer changed from:
"Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted"
to:
"Intersil Automotive Qualified products are manufactured, assembled and tested utilizing TS16949 quality systems
as noted"
Page 12- Updated the "About Intersil" verbiage
December 6, 2012
FN7909.2 Page 4 - Ordering Information table - changed part number from: ISL78225ANEZ-T to: ISL78225ANEZ and changed
Note 1 from: “Please refer to TB347 for details on reel specifications.” to: “Add “-T*” suffix for tape and reel. Please
refer to TB347 for details on reel specifications.”
Page 12, “Phase Selection” section: -Changed first sentence from: “The ISL78225 can work in 2, 3, or 4-phase
configuration”. to: “The ISL78225 can work in 1, 2, 3 or 4-phase configuration.”
Changed last sentence from “Unused current sense inputs must be left floating.” to: “For the unused ISENxN and
ISENxP, a 1k resistor is recommended to connect ISENxN and ISENxP, and connect ISENxN to VIN.”
Page 13, Table 1 - added comment to cell of row A and column APPLICATIONs: “Forced minimum ON pulses exists.”
Page 14, added sentence under section title “Soft-Start Process for Different Modes”: “At the beginning of soft-start,
the SS pin voltage will start ramping up from a voltage equal to the FB voltage. The soft-start period ends when the
SS pin voltage reaches the lower power-good threshold that is 80% of the lower value of VREF2 or 2V.”
August 2, 2012
FN7909.1 Changed the symbols of the main switching devices from IGBT to N-MOSFET in “Typical Application 1: 4-Phase
Synchronous Boost Converter with Sense Resistor Current Sensing” on page 5, “Typical Application 2: 4-Phase
Standard Boost Converter with DCR Current Sensing” on page 6, Figure 20 on page 16 and Figure 21 on page 17.
July 17, 2012
Updated guidance for unused current sense inputs, specified normal operation with inverting MOSFET drivers, and
added pointer to ISL78225EVAL1Z evaluation board.
December 15, 2011 FN7909.0 Initial release.
About Intersil
Intersil Corporation is a leading provider of innovative power management and precision analog solutions. The company's products
address some of the largest markets within the industrial and infrastructure, mobile computing and high-end consumer markets.
For the most updated datasheet, application notes, related documentation and related parts, please see the respective product
information page found at www.intersil.com.
You may report errors or suggestions for improving this datasheet by visiting www.intersil.com/ask.
Reliability reports are also available from our website at www.intersil.com/support
For additional products, see www.intersil.com/en/products.html
Intersil Automotive Qualified products are manufactured, assembled and tested utilizing TS16949 quality systems as noted
in the quality certifications found at www.intersil.com/en/support/qualandreliability.html
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time
without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be
accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
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21
FN7909.4
September 5, 2014
ISL78225
Package Outline Drawing
Q44.10x10A
44 LEAD THIN PLASTIC QUAD FLATPACK PACKAGE WITH EXPOSED PAD (EP-TQFP)
Rev 2, 12/10
4
10.00
12.00
5
D 3
3
A
12.00
10.00
4
5
4.50±0.1
B
3
0.80
EXPOSED PAD
4X
0.20 C A-B D
4X
0.20 H A-B D
4.50±0.1
TOP VIEW
BOTTOM VIEW
1.20 MAX
11/13°
7
0.05
0.20 M C A-B D
/ / 0.10 C
WITH LEAD FINISH
0.37 +0.08/-0.07
C
SIDE VIEW
0.10
SEE DETAIL "A"
0.09/0.20
0.09/0.16
0° MIN.
0.35 ±0.05
H
BASE METAL
2
1.00 ±0.05
0.05/0.15
(10.00)
0.08
R. MIN.
0.20 MIN.
DETAIL "A"
(0.45) TYP
SCALE: NONE
0.25
GAUGE
PLANE
0.60 ±0.15
0-7°
(1.00)
NOTES:
1. All dimensioning and tolerancing conform to ANSI Y14.5-1982.
2. Datum plane H located at mold parting line and coincident
with lead, where lead exits plastic body at bottom of parting line.
3. Datums A-B and D to be determined at centerline between
leads where leads exit plastic body at datum plane H.
10.00
(4.50)
(1.50) TYP
(4.50)
TYPICAL RECOMMENDED LAND PATTERN
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22
4. Dimensions D1 and E1 do not include mold protrusion.
Allowable mold protrusion is 0.254mm on D1 and E1
dimensions.
5. These dimensions to be determined at datum plane H.
6. Package top dimensions are smaller than bottom dimensions
and top of package will not overhang bottom of package.
7. Dimension b does not include dambar protrusion. Allowable
dambar protrusion shall be 0.08mm total in excess of the
b dimension at maximum material condition. Dambar cannot
be located on the lower radius or the foot.
8. Controlling dimension: millimeter.
9. This outline conforms to JEDEC publication 95 registration
MS-026, variation ACB.
10. Dimensions in ( ) are for reference only.
11. The corners of the exposed heatspreader may appear different
due to the presence of the tiebars.
FN7909.4
September 5, 2014
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