DATASHEET

ISL8500
®
Data Sheet
December 10, 2007
FN6611.0
2A Standard Buck PWM Regulator
Features
The ISL8500 is a high-performance, simple output controller
that provides a single, high frequency power solution for a
variety of point-of-load applications. The ISL8500 integrates
a 2A standard buck PWM controller and switching MOSFET.
• Standard Buck Controller with Integrated Switching Power
MOSFET
The PWM controller in the ISL8500 drives an internal
switching N-Channel power MOSFET and requires an
external Schottky diode to generate an output voltage from
0.6V to 19V. The integrated power switch is optimized for
excellent thermal performance up to 2A of output current.
The standard buck input voltage range supports a fixed 5V
or variable 5.5V to 25V range. The PWM regulator switches
at a fixed frequency of 500kHz and utilizes simple voltage
mode control with input voltage feed forward to provide
flexibility in component selection and minimize solution size.
Protection features include overcurrent, undervoltage and
thermal overload protection integrated into the IC. The
ISL8500 power good signal output indicates loss of
regulation on the PWM output.
ISL8500 is available in a small 4mmx3mm Dual Flat No-Lead
(DFN) package.
• Integrated Boot Diode
• Input Voltage Range
- Fixed 5V ±10%
- Variable 5.5V to 25V
• PWM Output Voltage Adjustable from 0.6V to 19V with
Continuous Output Current up to 2A
• ±1% VFB Tolerance
• Voltage Mode Control with Voltage Feed Forward
• Fixed 500kHz Switching Frequency
• Externally Adjustable Soft-Start Time
• Output Undervoltage Protection
• Enable Inputs
• PGOOD Output
• Overcurrent Protection
• Thermal Overload Protection
Ordering Information
PART
NUMBER
(Note)
ISL8500IRZ
PART
TEMP.
MARKING RANGE (°C)
• Internal 5V LDO regulator
PACKAGE
(Pb-free)
PKG.
DWG. #
Applications
• WLAN Cards-PCMCIA, Cardbus32, MiniPCI Cards-Compact
Flash Cards
500Z
-40 to +85
12 Ld DFN
L12.4x3
ISL8500IRZ-T* 500Z
-40 to +85
12 Ld DFN
L12.4x3
• General Purpose
*Please refer to TB347 for details on reel specifications.
• Hand-Held Instruments
NOTE: These Intersil Pb-free plastic packaged products employ
special Pb-free material sets; molding compounds/die attach
materials and 100% matte tin plate PLUS ANNEAL - e3 termination
finish, which is RoHS compliant and compatible with both SnPb and
Pb-free soldering operations. Intersil Pb-free products are MSL
classified at Pb-free peak reflow temperatures that meet or exceed
the Pb-free requirements of IPC/JEDEC J STD-020.
• LCD Panel
• Set-top Box
Pinout
ISL8500
(12 LD DFN)
TOP VIEW
FB
1
12 VIN
COMP
2
11
SS
3
VIN
10 PHASE
DFN 4x3
EN
4
PG
5
GND
1
6
GND
9
PHASE
8
BOOT
7
VDD
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2007. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL8500
Typical Application Schematic
R3
301
C3
100pF
VOUT
C2
2.2nF
R4
3.16k
FB
SS
C1
10pF
COMP
R2
51.1k
C5
0.1μF
R1
10k
VIN
5.5V TO 25V
C9
10uF
EN
PG
ISL8500
L
10μH
PHASE
BOOT
VOUT = 2.5V
C10
0.1μF
C11
100μF
D
B340LB
GND
VDD
C13
1µF
FIGURE 1. VIN RANGE FROM 5.5V TO 25V
2
FN6611.0
December 10, 2007
ISL8500
BOOT
FB
COMP
Functional Block Diagram
VDD
VDD
SOFT-START
CONTROL
VIN (x2)
30µA
OC
MONITOR
PWM
EA
+
-
VOLTAGE
MONITOR
+
-
SS
0.6V
REFERENCE
FAULT
MONITOR
EN
THERMAL
MONITOR
+150°C
RAMP
GENERATOR
VIN
GATE
DRIVE
PHASE (x2)
OSCILLATOR
OC
MONITOR
POR
VIN
LDO
POWER-ON
RESET
MONITOR
VDD
VDD
GND
PG
EPAD GND
3
FN6611.0
December 10, 2007
ISL8500
Absolute Maximum Ratings (Note 1)
Thermal Information
VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 26V
BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 33V
BOOT to PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.03V to 6V
VDD, FB, EN, COMP, PG, SS . . . . . . . . . . . . . . . . . . . . -0.3V to 6V
Thermal Resistance
Recommended Operating Conditions
VIN Supply Voltage Range . . . . . . . . . . . . . . . . . . . . . . 4.5V to 25V
Load Current Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0A to 2A
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . -40°C to +85°
θJA (°C/W)
θJC (°C/W)
QFN Package (Notes 1, 2). . . . . . . . . .
39
3
Ambient Temperature Range. . . . . . . . . . . . . . . . . . .-40°C to +85°C
Junction Temperature Range. . . . . . . . . . . . . . . . . .-40°C to +125°C
Storage Temperature Range . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379 for details.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. See Tech Brief TB379 for details.
3. Test Condition: VIN = 15V, FB forced above regulation point (0.6V), no switching, and power MOSFET gate charging current not included.
4. Excluding the blanking time.
5. Specifications at -40°C to +85°C are established by +25°C test with margin limits.
Electrical Specifications
Unless Otherwise Noted, All Parameter Limits are Established Over the Recommended Operating Conditions
and the Typical Specifications are Measured at the Following Conditions: TA = -40°C To +85°C (Note 5),
VIN = 5.5V to 25V, Unless Otherwise Noted. Typical Values are at TA = +25°C.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
5.5
-
25
V
4.5
5.0
5.5
V
SUPPLY VOLTAGE
VIN Voltage Range
VIN
VIN connected to VDD
VIN Operating Supply Current
IOP
Note 3
-
2
2.5
mA
VIN Shutdown Supply Current
ISD
VIN = 15V, EN = GND
-
80
100
µA
4.00
4.15
4.30
V
-
275
-
mV
4.5
5.00
5.5
V
VIN = 5.5V to 25V, IREF = 0
0.594
0.6
0.606
V
FB Line Regulation
IOUT = 0mA, VIN = 5.5V to 25V
-0.05
-
0.05
%
FB Leakage Current
VFB = 0.6V
-50
0
50
nA
450
500
550
kHz
0.65
0.75
0.95
V/V
-
1.3
-
V
0.75
0.8
0.85
V
80
-
-
%
POWER-ON RESET
VDD POR Threshold
Rising Edge
Hysteresis
INTERNAL VDD LDO
VDD Output Voltage Range
VIN = 5.5V to 25V, IVDD = 0mA to 30mA
REFERENCE
Reference Voltage
VFB
STANDARD BUCK PWM REGULATOR
OSCILLATOR AND PWM MODULATOR
Nominal Switching Frequency
fSW
Modulator Gain
AMOD
VIN = 12V (AMOD = 8/VIN)
Peak-to-Peak Sawtooth Amplitude
VRAMP
VIN = 12V (VP-P = VIN/8)
PWM Ramp Offset Voltage
VOFFSET
Maximum Duty Cycle
DCmax
4
COMP > 4V
FN6611.0
December 10, 2007
ISL8500
Electrical Specifications
Unless Otherwise Noted, All Parameter Limits are Established Over the Recommended Operating Conditions
and the Typical Specifications are Measured at the Following Conditions: TA = -40°C To +85°C (Note 5),
VIN = 5.5V to 25V, Unless Otherwise Noted. Typical Values are at TA = +25°C. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
-
88
-
dB
-
15
-
MHz
-
5
-
V/µs
1.2
1.7
2.2
V
-
400
-
mV
-1
-
1
µA
Rising Threshold
-
150
-
°C
THYS
Hysteresis
-
15
-
°C
VUV
Referred to Nominal VOUT
70
75
80
%
-
270
-
ns
2.56
3.2
3.94
A
-
100
-
ns
Lower Level, Falling Edge, with typically 15mV
hysteresis
85
88
91
%
Upper Level, Rising Edge, with typically 15mV
hysteresis
108
112
116
%
-
9
-
µs
ERROR AMPLIFIER
Open-Loop Gain
Gain Bandwidth Product
GBWP
Slew Rate
SR
COMP = 10pF
ENABLE SECTION
EN Threshold
Rising Edge
Hysteresis
EN Logic Input Current
FAULT PROTECTION
Thermal Shutdown Temperature
TSD
PWM UV Trip Level
PWM UVP Propagation Delay
PWM OCP Threshold
VIN = VDD = 5V, Note 4
OCP Blanking Time
POWER GOOD
PG Trip Level Referred to Nominal VOUT
PG Propagation Delay
PG Low Voltage
ISINK = 4mA
-
0.05
0.3
V
PG Leakage Current
VPG = 5.5V, VFB = 0.6V, VDD = 5.5V
-1
-
1
µA
Soft-Start Threshold to Enable Buck
0.9
1
1.1
V
Soft-Start Threshold to Enable PG
2.5
3.0
3.5
V
-
3.45
-
V
20
30
40
µA
VSS = 3.0V
-
25
-
mA
IOUT = 100mA, Die Resistance
-
120
200
mΩ
SOFT-START SECTION
Soft-Start Voltage High
Soft-Start Charging Current
Soft-Start Pull-down
POWER MOSFET
rDS(ON)
5
FN6611.0
December 10, 2007
ISL8500
Pin Descriptions
BOOT (Pin 8)
FB (Pin 1) and COMP (Pin 2)
The standard buck regulator employs a single voltage
control loop. FB is the negative input to the voltage loop error
amplifier. COMP is the output of the error amplifier. The
output voltage is set by an external resistor divider
connected to FB. With a properly selected divider, the output
voltage can be set to any voltage between the power rail
(reduced by converter losses) and the 0.6V reference.
Connecting an AC network across COMP and FB provides
loop compensation to the amplifier.
In addition, the PWM regulator power good and
undervoltage protection circuitry use FB to monitor the
regulator output voltage.
Floating bootstrap supply pin for the power MOSFET gate
driver. The bootstrap capacitor provides the necessary
charge to turn and hold on the internal N-Channel MOSFET.
Connect an external capacitor from this pin to PHASE.
PHASE (Pins 9, 10)
Switch node connections to internal power MOSFET source,
external output inductor and external diode cathode.
VIN (Pins 11, 12)
The input supply for the PWM regulator power stage and the
source for the internal linear regulator that provides bias for
the IC. Place a ceramic capacitor from VIN to GND, close to
the IC for decoupling (typical 10µF).
SS (Pin 3)
Program pin for soft-start duration. A regulated 30µA pull-up
current source charges a capacitor connected from the pin to
GND. The output voltage of the converter follows the
ramping voltage on the SS pin.
EN (Pin 4)
PWM controller enable input. The PWM converter output is
held off when the pin is pulled to ground. When the voltage
on this pin rises above 1.7V, the chip is enabled.
PG (Pin 5)
PWM converter power good output. Open drain logic output
that is pulled to ground when the output voltage is outside
regulation limits. Connect a 100kΩ resistor from this pin to
VDD. Pin is low when the buck regulator output voltage is
not within 10% of the respective nominal voltage, or during
the soft-start interval. Pin is high impedance when the output
is within regulation.
GND (Pin 6)
Ground connect for the IC and thermal relief for the package.
The exposed pad must be connected to GND and soldered
to the PCB. All voltage levels are measured with respect to
this pin.
VDD (Pin 7)
Internal 5V linear regulator output provides bias to all the
internal control logic. The ISL8500 may be powered directly
from a 5V (±10%) supply at this pin. When used as a 5V supply
input, this pin must be externally connected to VIN. The VDD
pin must always be decoupled to GND with a ceramic bypass
capacitor (minimum 1µF) located close to the pin.
TABLE 1. INPUT SUPPLY CONFIGURATION
INPUT
PIN CONFIGURATION
5.5V to 25V
Connect the input supply to the VIN pin only. The
VDD pin will provide a 5V output from the internal
linear regulator.
5V ±10%
Connect the input supply to the VIN and VDD pins.
6
FN6611.0
December 10, 2007
ISL8500
Typical Performance Curves
Unless otherwise noted, operating conditions are: TA = +25°C, VIN = 12V, EN = VDD, L = 10µH,
COUT = 100µF, C2 = 2x22µF, IOUT = 0A to 2A. See “The input supply for the PWM regulator
power stage and the source for the internal linear regulator that provides bias for the IC. Place a
ceramic capacitor from VIN to GND, close to the IC for decoupling (typical 10µF).” on page 6.
100
2.5VOUT
90
100
1.8VOUT
80
70
1.2VOUT
60
1.5VOUT
50
40
3.3VOUT
30
70
60
50
1.5VOUT
1.2VOUT
40
0.5
1.0
1.5
20
0.0
2.0
0.5
OUTPUT LOAD (A)
EFFICIENCY (%)
POWER DISSIPATION (W)
1.4
5VOUT
70
60
1.2VOUT
1.5VOUT
2.5VOUT
1.8VOUT
30
20
0.0
0.5
1.0
1.5
1.2
1.0
0.6
12VIN
0.4
5VIN
0.2
0.0
0.0
2.0
25VIN
0.8
0.5
FIGURE 4. EFFICIENCY vs LOAD, 25VIN
1.5
2.0
FIGURE 5. POWER DISSIPATION vs LOAD, 2.5VOUT
1.510
1.206
1.509
1.205
25VIN
12VIN
1.204
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
1.0
OUTPUT LOAD (A)
OUTPUT LOAD (A)
1.203
1.202
1.201
5VIN
1.200
1.199
1.198
0.0
2.0
1.6
80
40
1.5
FIGURE 3. EFFICIENCY vs LOAD, 12VIN
12VOUT
50
1.0
OUTPUT LOAD (A)
FIGURE 2. EFFICIENCY vs LOAD, 5VIN
90
1.8VOUT
30
20
0.0
100
5VOUT
80
EFFICIENCY (%)
EFFICIENCY (%)
2.5VOUT 3.3VOUT
90
25VIN
12VIN
1.508
1.507
1.506
1.505
5VIN
1.504
1.503
0.5
1.0
1.5
OUTPUT LOAD (A)
FIGURE 6. VOUT REGULATION vs LOAD, 1.2VOUT
7
2.0
1.502
0.0
0.5
1.0
1.5
2.0
OUTPUT LOAD (A)
FIGURE 7. VOUT REGULATION vs LOAD, 1.5VOUT
FN6611.0
December 10, 2007
ISL8500
Typical Performance Curves
Unless otherwise noted, operating conditions are: TA = +25°C, VIN = 12V, EN = VDD, L = 10µH,
COUT = 100µF, C2 = 2x22µF, IOUT = 0A to 2A. See “The input supply for the PWM regulator
power stage and the source for the internal linear regulator that provides bias for the IC. Place a
ceramic capacitor from VIN to GND, close to the IC for decoupling (typical 10µF).” on page 6.
(Continued)
2.506
1.814
1.812
2.505
25VIN
12VIN
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
1.813
1.811
1.810
1.809
5VIN
1.808
2.503
12VIN
2.502
2.501
2.500
5VIN
2.499
1.807
1.806
0.0
25VIN
2.504
0.5
1.0
1.5
2.498
0.0
2.0
0.5
OUTPUT LOAD (A)
FIGURE 8. VOUT REGULATION vs LOAD, 1.8VOUT
4.98
25VIN
12VIN
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
2.0
4.99
3.328
3.324
3.322
3.320
7VIN
3.318
3.316
3.314
0.0
1.5
FIGURE 9. VOUT REGULATION vs LOAD, 2.5VOUT
3.330
3.326
1.0
OUTPUT LOAD (A)
25VIN
4.97
4.96
4.95
7VIN
4.94
4.93
12VIN
4.92
0.5
1.0
1.5
2.0
4.91
0.0
0.5
FIGURE 10. VOUT REGULATION vs LOAD, 3.3VOUT
1.0
1.5
2.0
OUTPUT LOAD (A)
OUTPUT LOAD (A)
FIGURE 11. VOUT REGULATION vs LOAD, 5VOUT
PHASE 10V/DIV
PHASE 5V/DIV
VOUT RIPPLE
20mV/DIV
IL 0.5A/DIV
FIGURE 12. STEADY STATE OPERATION AT NO LOAD
(5µs/DIV)
8
VOUT RIPPLE
20mV/DIV
IL 1A/DIV
FIGURE 13. STEADY STATE OPERATION AT FULL LOAD
(1µs/DIV)
FN6611.0
December 10, 2007
ISL8500
Typical Performance Curves
Unless otherwise noted, operating conditions are: TA = +25°C, VIN = 12V, EN = VDD, L = 10µH,
COUT = 100µF, C2 = 2x22µF, IOUT = 0A to 2A. See “The input supply for the PWM regulator
power stage and the source for the internal linear regulator that provides bias for the IC. Place a
ceramic capacitor from VIN to GND, close to the IC for decoupling (typical 10µF).” on page 6.
(Continued)
PHASE 10V/DIV
EN 5V/DIV
VOUT 2V/DIV
VOUT RIPPLE
100mV/DIV
IL 0.5mA/DIV
PG 5V/DIV
IL 1A/DIV
SS 5V/DIV
FIGURE 14. LOAD TRANSIENT (200µs/DIV)
FIGURE 15. SOFT-START AT NO LOAD (2ms/DIV)
EN 5V/DIV
EN 5V/DIV
VOUT 2V/DIV
VOUT 2V/DIV
IL 2A/DIV
IL 1A/DIV
PG 5V/DIV
PG 5V/DIV
SS 5V/DIV
FIGURE 16. SOFT-START AT FULL LOAD (2ms/DIV)
FIGURE 17. SHUT DOWN CIRCUIT (100µs/DIV)
PHASE 10V/DIV
PHASE 10V/DIV
VOUT 1V/DIV
VOUT 2V/DIV
IL 2A/DIV
IL 2A/DIV
PG 5V/DIV
PG 5V/DIV
FIGURE 18. OUTPUT SHORT CIRCUIT (5µs/DIV)
9
FIGURE 19. OUTPUT SHORT CIRCUIT RECOVERY (1ms/DIV)
FN6611.0
December 10, 2007
ISL8500
Detailed Description
Power-On Reset and Undervoltage Lockout
The ISL8500 combines a standard buck PWM controller with
an integrated switching MOSFET. The buck controller drives
an internal N-Channel MOSFET and requires an external
diode to deliver load current up to 2A. A Schottky diode is
recommended for improved efficiency and performance over
a standard diode. The standard buck regulator can operate
from either an unregulated DC source, such as a battery,
with a voltage ranging from +5.5V to +25V, or from a
regulated system rail of +5V. When operating from +5.5V or
greater, the controller is biased from an internal +5V LDO
voltage regulator. The converter output is regulated down to
0.6V from either input source. These features make the
ISL8500 ideally suited for FPGA and wireless chipset power
applications.
The PWM portion of the ISL8500 automatically initializes
upon receipt of input power. The power-on reset (POR)
function continually monitors the VDD voltage. While below
the POR thresholds, the controller inhibits switching off the
internal power MOSFET. Once exceeded, the controller
initializes the internal soft-start circuitry. If either input supply
drops below their falling POR threshold during soft-start or
operation, the buck regulator latches off.
The PWM control loop uses a single output voltage loop with
input voltage feed forward, which simplifies feedback loop
compensation and rejects input voltage variation. External
feedback loop compensation allows flexibility in output filter
component selection. The regulator switches at a fixed 500kHz.
The buck regulator is equipped with a lossless current limit
scheme. The current limit in the buck regulator is achieved
by monitoring the drain-to-source voltage drop of the internal
switching power MOSFET. The current limit threshold is
internally set at 3.5A. The part also features undervoltage
protection by latching the switching MOSFET driver to the
OFF-state during an overcurrent, when the output voltage is
lower than 70% of the regulated output. This helps minimize
power dissipation during a short-circuit condition. Due to
only the switching power MOSFET integration, there is no
overvoltage protection feature for this part.
+5V Internal Bias Supply (VDD)
Voltage applied to the VIN pin with respect to GND is
regulated to +5V DC by an internal LDO regulator. The output
of the LDO, VDD, is the bias voltage used by all the internal
control and protection circuitry. The VDD pin requires a
ceramic capacitor connected to GND. The capacitor serves to
stabilize the LDO and to decouple load transients.
The input voltage range for the ISL8500 is specified as
+5.5V to +25V or +5V ±10%. In the case of an unregulated
supply case, the power supply is connected to VIN only.
Once enabled, the linear regulator will turn-on and rise to
+5V on VDD. In the +5V supply case, the VDD and VIN pins
must be tied together to bypass the LDO. The external
decoupling capacitor is still required in this mode.
Operation Initialization
The power-on reset circuitry and enable inputs prevent false
start-up of the PWM regulator output. Once all the input
criteria are met, the controller soft-starts the output voltage
to the programmed level.
10
Enable and Disable
All internal power devices are held in a high-impedance
state, which ensures they remain off while in shutdown
mode. Typically, the enable input for a specific output is
toggled high after the input supply to that regulator is active
and the internal LDO has exceeded it’s POR threshold.
The EN pin enables the buck controller portion of the
ISL8500. When the voltage on the EN pin exceeds the POR
rising threshold, the controller initiates the soft-start function
for the PWM regulator. If the voltage on the EN pin drops
below the POR falling threshold, the buck regulator shuts
down.
Pulling the EN pin low simultaneously put the output into
shutdown mode and supply current drops to 100µA typical.
Soft-Start
Once the input supply latch and enable threshold are met, the
soft-start function is initialized. The soft-start circuitry begins
sourcing 30µA, from an internal current source, which
charges the external soft-start capacitor. The voltage on SS
begins ramping linearly from ground until the voltage across
the soft-start capacitor reaches 3.0V. This linear ramp is
applied to the non-inverting input of the internal error amplifier
and overrides the nominal 0.6V reference. The output voltage
reaches its regulation value when the soft-start capacitor
voltage reaches 1.6V. Connect a capacitor from SS pin to
ground. This capacitor (along with an internal 30µA current
source) sets the soft-start interval of the converter, TSS.
C SS [ μF ] = 50 ⋅ T SS [ s ]
(EQ. 1)
Upon disable, the SS pin voltage will discharge to zero voltage.
Power Good
PG is an open-drain output of a window comparator that
continuously monitors the buck regulator output voltage. PG is
actively held low when EN is low and during the buck regulator
soft-start period. After the soft-start period terminates, PG
becomes high impedance as long as the output voltage is
within ±12% of the nominal regulation voltage set by FB. When
VOUT drops 12% below or rises 12% above the nominal
regulation voltage, the ISL8500 pulls PG low. Any fault
condition forces PG low until the fault condition is cleared by
attempts to soft-start. For logic level output voltages, connect
an external pull-up resistor between PG and VDD. A 100kΩ
resistor works well in most applications.
FN6611.0
December 10, 2007
ISL8500
Output Voltage Selection
The regulator output voltages can be programmed using
external resistor dividers that scale the voltage feedback
relative to the internal reference voltage. The scaled voltage
is fed back to the inverting input of the error amplifier; refer to
Figure 20.
The output voltage programming resistor, R4, will depend on
the value chosen for the feedback resistor, R1, and the desired
output voltage, VOUT, of the regulator; see Equation 2. The
value for the feedback resistor is typically between 1kΩ and
10kΩ.
R 1 × 0.6V
R 4 = ---------------------------------V OUT – 0.6V
cycles, the overcurrent fault counter overflows, indicating an
overcurrent fault condition exists. The regulator is shut down
and power good goes low. If the overcurrent condition clears
prior to the counter reaching four consecutive cycles, the
internal flag and counter are reset.
The protection circuitry attempts to recover from the
overcurrent condition after waiting 4 soft-start cycles. The
internal overcurrent flag and counter are reset. A normal
soft-start cycle is attempted and normal operation continues
if the fault condition has cleared. If the overcurrent fault
counter overflows during soft-start, the converter shuts down
and this hiccup mode operation repeats.
(EQ. 2)
There is 100ns blanking time for noise immunity. It is
recommended to operate the duty cycle higher than the
blanking time to insure proper overcurrent protection.
If the output voltage desired is 0.6V, then RP is left
unpopulated.
VOUT
R1
+
EA
R4
0.6V
REFERENCE
FIGURE 20. EXTERNAL RESISTOR DIVIDER
The buck output can be programmed as high as 19V. Proper
heatsinking must be provided to insure that the junction
temperature does not exceed +125°C.
When the output is set greater than 2.7V, it is recommended to
pre-load at least 10mA and make sure that the input rise time is
>> faster than the VOUT1 rise time. This allows the BOOT
capacitor adequate time to charge for proper operation.
Protection Features
The ISL8500 limits current in the power devices to limit on-chip
power dissipation. Overcurrent limits on the regulator protect
the internal power device from excessive thermal damage.
Undervoltage protection circuitry on the buck regulator provides
a second layer of protection for the internal power device under
high current condition.
Undervoltage Protection
If the voltage detected on the buck regulator FB pin falls 25%
below the internal reference voltage, the undervoltage fault
condition flag is set. The regulator is shutdown. The
controller enters a recovery mode similar to the overcurrent
hiccup mode. No action is taken for 4 soft-start cycles and
the internal undervoltage counter and fault condition flag are
reset. A normal soft-start cycle is attempted and normal
operation continues if the fault condition has cleared. If the
undervoltage counter overflows during soft-start, the
converter is shut down and this hiccup mode operation
repeats.
Thermal Overload Protection
Thermal overload protection limits total power dissipation in
the ISL8500. There is a sensor on the chip to monitor the
junction temperature of the internal LDO and PWM switching
power N-Channel MOSFET. When the junction temperature
(TJ) of the sensor exceeds +150°C, the thermal sensor
sends a signal to the fault monitor.
The fault monitor commands the buck regulator to shut down.
The buck regulator soft-starts turn on again after the IC’s
junction temperature cools by +20°C. The buck regulator
experiences hiccup mode operation during continuous thermal
overload conditions. For continuous operation, do not exceed
the +125°C junction temperature rating.
Buck Regulator Overcurrent Protection
During the PWM on-time, the current through the internal
switching MOSFET is sampled and scaled through an
internal pilot device. The sampled current is compared to a
nominal 3.5A overcurrent limit. If the sampled current
exceeds the overcurrent limit reference level, an internal
overcurrent fault counter is set to 1 and an internal flag is
set. The internal power MOSFET is immediately turned off
and will not be turned on again until the next switching cycle.
The protection circuitry continues to monitor the current and
turns off the internal MOSFET as described. If the
overcurrent condition persists for eight sequential clock
11
Application Guidelines
Operating Frequency
The ISL8500 operates at a fixed switching frequency of
500kHz.
Buck Regulator Output Capacitor Selection
An output capacitor is required to filter the inductor current
and supply the load transient current. The filtering
requirements are a function of the switching frequency and
the ripple current. The load transient requirements are a
function of the slew rate (di/dt) and the magnitude of the
FN6611.0
December 10, 2007
ISL8500
transient load current. These requirements are generally met
with a mix of capacitors and careful layout.
Embedded processor systems are capable of producing
transient load rates above 1A/ns. High frequency capacitors
initially supply the transient and slow the current load rate
seen by the bulk capacitors. The bulk filter capacitor values
are generally determined by the ESR (Effective Series
Resistance) and voltage rating requirements rather than
actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by Equation 3:
ΔI =
VIN - VOUT
Fs x L
x
VOUT
VIN
ΔVOUT = ΔI x ESR
(EQ. 3)
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL8500 will provide either 0% or 80% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval, the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
12
The response time to a transient is different for the
application of load and the removal of load. Equation 4 gives
the approximate response time interval for application and
removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
VOUT
(EQ. 4)
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check Equation 4 at the minimum and
maximum output levels for the worst case response time.
Rectifier Selection
Current circulates from ground to the junction of the MOSFET
and the inductor when the high-side switch is off. As a
consequence, the polarity of the switching node is negative
with respect to ground. This voltage is approximately -0.5V (a
Schottky diode drop) during the off-time. The rectifier's rated
reverse breakdown voltage must be at least equal to the
maximum input voltage, preferably with a 20% derating factor.
The power dissipation is shown in Equation 5:
V OUT⎞
⎛
P D [ W ] = I OUT ⋅ V D ⋅ ⎜ 1 – ----------------⎟
V IN ⎠
⎝
(EQ. 5)
where VD is the voltage of the Schottky diode = 0.5V to 0.7V
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time the switching
MOSFET turns on. Place the small ceramic capacitors
physically close to the MOSFET VIN pins (switching
MOSFET drain) and the Schottky diode anode.
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. Their voltage rating should be
at least 1.25 times greater than the maximum input voltage,
while a voltage rating of 1.5 times is a conservative guideline.
For most cases, the RMS current rating requirement for the
input capacitor of a buck regulator is approximately 1/2 the
DC load current.
The maximum RMS current required by the regulator may be
closely approximated through Equation 6:
I RMS
MAX
=
V OUT ⎛
V IN – V OUT V OUT 2
2
1
-------------- × I OUT
+ ------ × ⎛ ----------------------------- × --------------⎞ ⎞
⎝
V IN
V IN ⎠ ⎠
12 ⎝ L × f s
MAX
(EQ. 6)
For a through hole design, several electrolytic capacitors
may be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
FN6611.0
December 10, 2007
ISL8500
regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge-current at
power-up. Some capacitor series available from reputable
manufacturers are surge current tested.
VIN
DRIVER
OSC
PWM
COMPARATOR
VDDQ
PHASE
5. Place 2ND Pole at Half the Switching Frequency.
ESR
(PARASITIC)
VE/A
6. Check Gain against Error Amplifier’s Open-Loop Gain.
7. Estimate Phase Margin - Repeat if Necessary.
ZIN
+
Compensation Break Frequency Equations
REFERENCE
DETAILED COMPENSATION COMPONENTS
ZFB
C1
C2
C3
R3
1
F Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3
1
F P2 = -----------------------------------2π x R 3 x C 3
(EQ. 8)
R1
COMP
FB
+
ISL8500
1
F P1 = --------------------------------------------------------⎛ C 1 x C 2⎞
2π x R 2 x ⎜ ----------------------⎟
⎝ C1 + C2 ⎠
VOUT
ZIN
R2
1
F Z1 = -----------------------------------2π x R 2 x C 2
R4
REFERENCE
FIGURE 21. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN AND OUTPUT
VOLTAGE SELECTION
Feedback Compensation
Figure 21 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The error
amplifier output (VE/A) is compared with the oscillator (OSC)
triangular wave to provide a pulse-width modulated (PWM)
wave with an amplitude of VIN at the PHASE node. The
PWM wave is smoothed by the output filter (LO and CO).
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR . The DC Gain of
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage ΔVOSC .
Modulator Break Frequency Equations
100
FZ1 FZ2
FP1
FP2
80
OPEN LOOP
ERROR AMP GAIN
60
40
20
20LOG
(R2/R1)
20LOG
(VIN/ΔVOSC)
0
COMPENSATION
GAIN
MODULATOR
GAIN
-20
CLOSED LOOP
GAIN
-40
FLC
1
F ESR = -------------------------------------------2π x ESR x C O
-60
(EQ. 7)
The compensation network consists of the error amplifier
(internal to the ISL8500) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180°. Equation 8 relates the compensation network’s poles,
13
Figure 22 shows an asymptotic plot of the DC/DC
converter’s gain vs frequency. The actual Modulator Gain
has a high gain peak due to the high Q factor of the output
filter and is not shown in Figure 22. Using the previously
mentioned guidelines should give a Compensation Gain
similar to the curve plotted. The open loop error amplifier
gain bounds the compensation gain. Check the
compensation gain at FP2 with the capabilities of the error
amplifier. The Closed Loop Gain is constructed on the graph
of Figure 4 by adding the Modulator Gain (in dB) to the
Compensation Gain (in dB). This is equivalent to multiplying
the modulator transfer function to the compensation transfer
function and plotting the gain.
GAIN (dB)
ERROR
AMP
1
F LC = ------------------------------------------2π x L O x C O
3. Place 2ND Zero at Filter’s Double Pole.
4. Place 1ST Pole at the ESR Zero.
CO
D
ZFB
1. Pick Gain (R2/R1) for desired converter bandwidth.
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).
LO
+
ΔVOSC
zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and
C3) in Figure 22. Use the following guidelines for locating the
poles and zeros of the compensation network:
10
100
1k
FESR
10k
100k
1M
10M
FREQUENCY (Hz)
FIGURE 22. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW) overall
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45°.
Include worst case component variations when determining
phase margin.
FN6611.0
December 10, 2007
ISL8500
A more detailed explanation of voltage mode control of a
buck regulator can be found in TB417, entitled “Designing
Stable Compensation Networks for Single Phase Voltage
Mode Buck Regulators.”
VIN
VIN
CIN
ISL8500
Layout Considerations
L
As an example, consider the turn-off transition of the upper
MOSFET. Prior to turn-off, the MOSFET is carrying the full load
current. During turn-off, current stops flowing in the MOSFET
and is picked up by the Schottky diode. Any parasitic
inductance in the switched current path generates a large
voltage spike during the switching interval. Careful component
selection, tight layout of the critical components, and short, wide
traces minimizes the magnitude of voltage spikes.
There are two sets of critical components in the ISL8500
switching converter. The switching components are the most
critical because they switch large amounts of energy, and
therefore tend to generate large amounts of noise. Next are
the small signal components, which connect to sensitive
nodes or supply critical bypass current and signal coupling.
A multi-layer printed circuit board is recommended. Figure 23
shows the connections of the critical components in the
converter. Note that capacitors CIN and COUT could each
represent numerous physical capacitors. Dedicate one solid
layer, usually a middle layer of the PC board, for a ground
plane and make all critical component ground connections
with vias to this layer. Dedicate another solid layer as a power
plane and break this plane into smaller islands of common
voltage levels. Keep the metal runs from the PHASE terminals
to the output inductor short. The power plane should support
the input power and output power nodes. Use copper filled
polygons on the top and bottom circuit layers for the phase
nodes. Use the remaining printed circuit layers for small signal
wiring.
VDD
5V
D
CBP1
VOUT1
PHASE
LOAD
Layout is very important in high frequency switching
converter design. With power devices switching efficiently
between 100kHz and 600kHz, the resulting current
transitions from one device to another cause voltage spikes
across the interconnecting impedances and parasitic circuit
elements. These voltage spikes can degrade efficiency,
radiate noise into the circuit, and lead to device overvoltage
stress. Careful component layout and printed circuit board
design minimizes these voltage spikes.
COUT1
GND
COMP
C2
C1
R2
R1
FB
R4
C3
R3
GND PAD
KEY
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT AND/OR POWER PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 23. PRINTED CIRCUIT BOARD POWER PLANES AND
ISLANDS
possible. Position the output inductor and output capacitors
between the upper and Schottky diode and the load.
The critical small signal components include any bypass
capacitors, feedback components, and compensation
components. Place the PWM converter compensation
components close to the FB and COMP pins. The feedback
resistors should be located as close as possible to the FB
pin with vias tied straight to the ground plane as required.
In order to dissipate heat generated by the internal LDO and
MOSFET, the ground pad, pin 13, should be connected to
the internal ground plane through at least four vias. This
allows the heat to move away from the IC and also ties the
pad to the ground plane through a low impedance path.
The switching components should be placed close to the
ISL8500 first. Minimize the length of the connections
between the input capacitors, CIN, and the power switches
by placing them nearby. Position both the ceramic and bulk
input capacitors as close to the upper MOSFET drain as
14
FN6611.0
December 10, 2007
ISL8500
Dual Flat No-Lead Plastic Package (DFN)
L12.4x3
12 LEAD DUAL FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-229-VGED-4 ISSUE C)
2X
0.15 C A
A
D
MILLIMETERS
2X
0.15 C B
SYMBOL
0.80
A1
-
A3
E
6
INDEX
AREA
b
D2
B
//
A
SIDE VIEW
C
SEATING
PLANE
0.10
0.08
A3
7
8
-
-
0.05
-
0.23
0.30
5,8
4.00 BSC
3.15
3.30
3.40
7,8
3.00 BSC
1.55
e
1.70
1.80
7,8
0.50 BSC
-
k
0.20
-
-
-
L
0.30
0.40
0.50
8
N
12
2
Nd
6
3
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd refers to the number of terminals on D.
2
4. All dimensions are in millimeters. Angles are in degrees.
NX k
(DATUM A)
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
E2
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
E2/2
NX L
N
N-1
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
NX b
e
8. Nominal dimensions are provided to assist with PCB Land
Pattern Design efforts, see Intersil Technical Brief TB389.
5
(Nd-1)Xe
REF.
BOTTOM VIEW
NX (b)
C
NOTES
1.00
NOTES:
D2/2
1
C
MAX
0.90
Rev. 1 2/05
D2
(DATUM B)
8
0.18
E
E2
NOMINAL
0.20 REF
D
TOP VIEW
6
INDEX
AREA
MIN
A
0.10
M C A B
CL
(A1)
L
5
e
SECTION "C-C" TERMINAL TIP
FOR EVEN TERMINAL/SIDE
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
15
FN6611.0
December 10, 2007
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