HIP6028 Data Sheet May 1998 Advanced PWM and Dual Linear Power Control with Integrated ACPI Support Interface The HIP6028 provides the power control and protection for three output voltages in high-performance microprocessor and computer applications. The IC integrates a PWM controller, a linear regulator and a linear controller as well as the monitoring and protection functions into a single package. The PWM controller regulates the microprocessor core voltage with a synchronous-rectified buck converter. The linear controller regulates power for the GTL bus and the linear regulator provides power for the clock driver circuit. The HIP6028 includes an Intel-compatible, TTL 5-input digital-to-analog converter (DAC) that adjusts the core PWM output voltage from 2.1VDC to 3.5VDC in 0.1V increments and from 1.3VDC to 2.05VDC in 0.05V steps. The precision reference and voltage-mode control provide ±1% static regulation. The linear regulator uses an internal pass device to provide a fixed 2.5V ±2.5%. The linear controller drives an external N-channel MOSFET to provide a fixed 1.5V ±2.5%. The HIP6028 monitors all the output voltages. A single Power Good signal is issued when the core is within ±10% of the DAC setting and the other levels are above their undervoltage levels. Additional built-in over-voltage protection for the core output uses the lower MOSFET to prevent output voltages above 115% of the DAC setting. The PWM overcurrent function monitors the output current by using the voltage drop across the upper MOSFET’s rDS(ON), eliminating the need for a current sensing resistor. The HIP6028 offers integrated ACPI S3 shutdown state support. Through the SD1&3 pin, the microprocessor core and GTL bus supplies can be shut down when entering power-saving standby operation mode. HIP6028CB HIP6028EVAL1 Features • Provides 3 Regulated Voltages - Microprocessor Core, Clock and GTL Power • Integrated ACPI S3-State Shutdown Support • Drives Low Cost Transistors - PWM Controller Drives N-MOSFETs - Linear Controller Compatible with Both MOSFETs and NPN Bipolar Transistors • Operates from +3.3V, +5V and +12V Inputs • Simple Control Design - Single-Loop Voltage-Mode PWM Control - Fixed 1.5V GTL Output Voltage - Fixed 2.5V Clock Output Voltage • Fast Transient Response - High-Bandwidth Error Amplifier - Full 0% to 100% Duty Ratio • Excellent Output Voltage Regulation - Core PWM Output: ±1% Over Temperature - Other Outputs: ±2.5% Over Temperature • TTL-compatible 5-Bit Digital-to-Analog Core Output Voltage Selection - Wide Range . . . . . . . . . . . . . . . . . . . 1.3VDC to 3.5VDC • Power-Good Output Voltage Monitor • Microprocessor Core Voltage Protection Against Shorted MOSFET • Over-Voltage and Over-Current Fault Monitors - Does Not Require Extra Current Sensing Element, Uses MOSFET’s rDS(ON) • Small Converter Size - Constant Frequency Operation; 200kHz Free-Running Oscillator; Programmable from 50kHz to 1MHz Applications • Full Motherboard Power Regulation for Computers • Low-Voltage Distributed Power Supplies HIP6028 (SOIC) TOP VIEW TEMP. RANGE (oC) 0 to 70 PACKAGE 24 Ld SOIC Evaluation Board PKG. NO. M24.3 VCC 1 24 UGATE VID4 2 23 PHASE VID3 3 22 LGATE VID2 4 21 PGND VID1 5 20 OCSET VID0 6 19 VSEN1 PGOOD 7 18 FB FAULT 8 17 COMP SS 9 16 VSEN3 RT 10 SD1&3 11 VIN2 12 2-311 4630 Pinout Ordering Information PART NUMBER File Number 15 DRIVE3 14 GND 13 VOUT2 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. http://www.intersil.com or 407-727-9207 | Copyright © Intersil Corporation 1999 2-312 SD1&3 SS FAULT VOUT2 VIN2 DRIVE3 0.23A - + 3.3V CLIM2 - + + - - + 0.3V 1.26V + VSEN3 4V 11µA VCC OV - + + - - + - + FB OC1 COMP ERROR AMP 115% 90% 110% FIGURE 1. DACOUT SOFTSTART & FAULT LOGIC TTL D/A CONVERTER (DAC) POR LUV VID4 VID0 VID2 VID1 VID3 - + - + LINEAR UNDERVOLTAGE VSEN1 PWM INHIBIT RT OSCILLATOR PWM COMP - + - + 200µA OCSET LOWER DRIVE GATE CONTROL UPPER DRIVE RESET (POR) POWER-ON VCC VCC VCC 3.3V GND PGND LGATE PHASE UGATE PGOOD HIP6028 Block Diagram HIP6028 Simplified Power System Diagram +5VIN Q1 +3.3VIN LINEAR REGULATOR VOUT2 PWM CONTROLLER VOUT1 HIP6028 Q2 LINEAR CONTROLLER Q3 VOUT3 FIGURE 2. Typical Application +12VIN +5VIN CIN VCC OCSET VIN2 +3.3VIN POWERGOOD PGOOD VOUT2 VOUT2 2.5V UGATE COUT2 Q1 PHASE LGATE Q3 Q2 COUT1 PGND DRIVE3 HIP6028 VOUT3 VSEN3 VSEN1 1.5V COUT3 FB SD1&3 COMP VID0 VID1 VID2 FAULT VID3 RT VID4 SS GND FIGURE 3. 2-313 VOUT1 1.3V TO 3.5V LOUT1 CSS HIP6028 Absolute Maximum Ratings Thermal Information Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +15V PGOOD, RT, FAULT, DRIVE3, LGATE, and UGATE Voltage . . . . . . . . . GND - 0.3V to VCC + 0.3V Other Input, Output or I/O Voltage . . . . . . . . . . . . . GND -0.3V to 7V Thermal Resistance (Typical, Note 1) θJA (oC/W) SOIC Package. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75 Maximum Junction Temperature (Plastic Package) . . . . . . . 150oC Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 3 00oC (SOIC - Lead Tips Only) Operating Conditions Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10% Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . 0oC to 70oC Junction Temperature Range . . . . . . . . . . . . . . . . . . . 0oC to 125oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTE: 1. θJA is measured with the component mounted on an evaluation PC board in free air. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Figures 1, 2 and 3 PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS - 8 - mA VCC SUPPLY CURRENT Nominal Supply ICC UGATE, DRIVE3, LGATE, and VOUT2 Open POWER-ON RESET Rising VCC Threshold VOCSET = 4.5V 8.6 - 10.4 V Falling VCC Threshold VOCSET = 4.5V 8.2 - 10.2 V 2.45 2.55 2.65 V VIN2 Under-Voltage Hysteresis - 0.5 - V Rising VOCSET Threshold - 1.25 - V Rising VIN2 Under-Voltage Threshold OSCILLATOR Free Running Frequency RT = OPEN 185 200 215 kHz Total Variation 6kΩ < RT to GND < 200kΩ -15 - +15 % - 1.9 - VP-P DAC(VID0-VID4) Input Low Voltage - - 0.8 V DAC(VID0-VID4) Input High Voltage 2.0 - - V DACOUT Voltage Accuracy -1.0 - +1.0 % 2.437 2.500 2.563 V - 1.875 2.175 V - 0.150 - V 180 230 - mA 1.462 1.500 1.538 V - 1.125 1.305 V - 0.090 - V 20 40 - mA - 88 - dB - 15 - MHz ∆VOSC Ramp Amplitude RT = Open REFERENCE and DAC LINEAR REGULATOR Regulation 10mA < IVOUT2 < 150mA Under-Voltage Level VOUT2UV VOUT2 Rising Under-Voltage Hysteresis Over-Current Protection (Current-Limiting) LINEAR CONTROLLER Regulation VSEN3 = DRIVE3 Under-Voltage Level VSEN3UV VSEN3 Rising Under-Voltage Hysteresis DRIVE3 Source Current VSEN3 = 1.4V, DRIVE3 = 2V, VIN2 = 3.3V PWM CONTROLLER ERROR AMPLIFIER DC Gain Gain-Bandwidth Product GBWP 2-314 HIP6028 Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Figures 1, 2 and 3 (Continued) PARAMETER SYMBOL Slew Rate SR TEST CONDITIONS MIN TYP MAX UNITS COMP = 10pF - 6 - V/µs PWM CONTROLLER GATE DRIVER Upper Drive Source IUGATE VCC = 12V, VUGATE (or VGATE2) = 6V - 1 - A Upper Drive Sink RUGATE VUGATE-PHASE = 1V - 1.7 3.5 Ω Lower Drive Source ILGATE VCC = 12V, VLGATE = 1V - 1 - A Lower Drive Sink RLGATE VLGATE = 1V - 1.4 3.0 Ω VSEN1 Rising 112 115 118 % VFAULT = 10V 10 14 - mA VOCSET = 4.5VDC 170 200 230 µA - 11 - µA VOUT1 and VOUT3 Disable Low Voltage - - 0.8 V VOUT1 and VOUT3 Disable High Voltage 2.0 - - V PROTECTION AND CONTROL VOUT1 Over-Voltage Trip FAULT Sourcing Current IOVP OCSET Current Source IOCSET Soft-Start Current ISS POWER GOOD VOUT1 Upper Threshold VSEN1 Rising 108 - 110 % VOUT1 Under Voltage VSEN1 Rising 92 - 94 % VOUT1 Hysteresis (VSEN1 / DACOUT) Upper/Lower Threshold - 2 - % IPGOOD = -4mA - - 0.5 V PGOOD Voltage Low VPGOOD Typical Performance Curves 100 CGATE = 4800pF CUGATE = CLGATE = CGATE VVCC = 12V, VIN = 5V 80 RT PULLUP TO +12V ICC (mA) RESISTANCE (kΩ) 1000 100 60 CGATE = 3600pF 40 10 CGATE = 1500pF RT PULLDOWN TO VSS 20 CGATE = 660pF 10 100 SWITCHING FREQUENCY (kHz) 1000 0 100 200 300 400 500 600 700 800 900 SWITCHING FREQUENCY (kHz) FIGURE 4. RT RESISTANCE vs FREQUENCY 2-315 FIGURE 5. BIAS SUPPLY CURRENT vs FREQUENCY 1000 HIP6028 Functional Pin Description UGATE (Pin 24) VSEN1 (Pin 19) Connect UGATE pin to the PWM converter’s upper MOSFET gate. This pin provides the gate drive for the upper MOSFET. This pin is connected to the PWM converter’s output voltage. The PGOOD and OVP comparator circuits use this signal to report output voltage status and for over voltage protection. OCSET (Pin 20) PGND (Pin 21) This is the power ground connection. Tie the PWM converter’s lower MOSFET source to this pin. Connect a resistor (ROCSET) from this pin to the drain of the upper MOSFET. ROCSET, an internal 200µA current source (IOCSET), and the upper MOSFET on-resistance (rDS(ON)) set the PWM converter over-current (OC) trip point according to the following equation: LGATE (Pin 22) I OCSET × R OCSET I PEAK = ---------------------------------------------------r DS ( ON ) Provide a 12V bias supply for the IC to this pin. This pin also provides the gate bias charge for all the MOSFETs controlled by the IC. An over-current trip cycles the soft-start function. Sustaining an over-current for 2 soft-start intervals shuts down the controller. RT (Pin 10) SS (Pin 9) Connect a capacitor from this pin to ground. This capacitor, along with an internal 11µA (typically) current source, sets the soft-start interval of the PWM converter and GTL controller. VID0, VID1, VID2, VID3, VID4 (Pins 6, 5, 4, 3 and 2) VID0-4 are the input pins to the 5-bit DAC. The states of these five pins program the internal voltage reference (DACOUT). The level of DACOUT sets the core converter output voltage. It also sets the core PGOOD and OVP thresholds. Connect LGATE to the PWM converter’s lower MOSFET gate. This pin provides the gate drive for the lower MOSFET. VCC (Pin 1) This pin provides oscillator switching frequency adjustment. By placing a resistor (RT) from this pin to GND, the nominal 200kHz switching frequency is increased according to the following equation: 6 5 × 10 Fs ≈ 200kHz + --------------------R T ( kΩ ) (RT to GND) Conversely, connecting a pull-up resistor (RT) from this pin to VCC reduces the switching frequency according to the following equation: 7 4 × 10 Fs ≈ 200kHz – --------------------R T ( kΩ ) (RT to 12V) COMP and FB (Pins 17 and 18) FAULT (Pin 8) COMP and FB are the available external pins of the PWM error amplifier. The FB pin is the inverting input of the error amplifier and the COMP pin is the error amplifier output. These pins are used to compensate the voltage-control feedback loop of the PWM converter. This pin is low during normal operation, but it is pulled to VCC in the event of an over-voltage or over-current condition. GND (Pin 14) Signal ground for the IC. All voltage levels are measured with respect to this pin. PGOOD (Pin 7) PGOOD is an open collector output used to indicate the status of the output voltages. This pin is pulled low when the core output is not within ±10% of the DACOUT reference voltage and the other outputs are below their under-voltage thresholds. The PGOOD output is open for “11111” VID code combination. PHASE (Pin 23) Connect the PHASE pin to the PWM converter’s upper MOSFET source. This pin is used to monitor the voltage drop across the upper MOSFET for over-current protection. 2-316 DRIVE3 (Pin 15) Connect this pin to the gate of an external N-MOSFET or the base of an external NPN bipolar transistor. This pin provides the drive for the linear controller’s pass transistor. VSEN3 (Pin 16) Connect this pin to the linear controller’s pass MOSFET source. The voltage at this pin is regulated to 1.5V. VOUT2 (Pin 13) Output of the linear regulator. The voltage at this pin is regulated to 2.5V for output currents up to 230mA (typically). SD1&3 (Pin 11) A TTL-compatible, high level signal applied to this pin disables the PWM (VOUT1) and linear controller (VOUT3) outputs. Leaving this pin unconnected or connecting it to ground enables these two outputs. HIP6028 VIN2 (Pin 12) This pin supplies power to the internal regulator. Connect this pin to a suitable 3.3V source. Additionally, this pin is used to monitor the 3.3V supply. If, following a startup cycle, the voltage drops below 2.05V (typically), the chip shuts down. A new soft-start cycle is initiated upon return of the 3.3V supply above the undervoltage threshold. Description Operation The HIP6028 monitors and precisely controls 3 output voltage levels (Refer to Figures 1, 2, and 3). It is designed for microprocessor computer applications with 3.3V and 5V power, and 12V bias input from an ATX power supply. The IC has one PWM controller, a linear controller, and a linear regulator. The PWM controller is designed to regulate the microprocessor core voltage (VOUT1) by driving 2 MOSFETs (Q1 and Q2) in a synchronous-rectified buck converter configuration. The core voltage is regulated to a level programmed by the 5-bit digital-to-analog converter (DAC). An integrated linear regulator supplies the 2.5V clock power (VOUT2). The linear controller drives an external MOSFET or bipolar NPN (Q3) to supply the 1.5V GTL bus power (VOUT3). transition to the final set voltage. Additionally, the linear controller’s reference input is clamped to a voltage proportional to the SS pin voltage. This method provides a rapid and controlled output voltage rise. Figure 6 shows the soft-start sequence for the typical application. At time T0 the POR is released, SS voltage rapidly increases to approximately 1V, and VOUT2 rapidly ramps up to 2.5V. At T1, the SS pin and error amplifier output voltage reach the valley of the oscillator’s triangle wave. The oscillator’s triangular waveform is compared to the clamped error amplifier output voltage. As the SS pin voltage increases, the pulse-width on the PHASE pin increases. The interval of increasing pulse-width continues until each output reaches sufficient voltage to transfer control to the input reference clamp. If we consider the 2.0V output (VOUT1) in Figure 3, this time occurs at T2. During the interval between T2 and T3, the error amplifier reference ramps to the final value and the converter regulates the output to a voltage proportional to the SS pin voltage. At T3 the input clamp voltage exceeds the reference voltage and the output voltage is in regulation. POR 0V PGOOD (2V/DIV) Initialization The HIP6028 automatically initializes upon receipt of input power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the input supply voltages. The POR monitors the bias voltage (+12VIN) at the VCC pin, the 5V input voltage (+5VIN) on the OCSET pin, and the 3.3V input voltage (+3.3VIN) on the VIN2 pin. The normal level on OCSET is equal to +5VIN less a fixed voltage drop (see over-current protection). The POR function initiates soft-start operation after all three input supply voltages exceed their POR thresholds. 0V SOFT-START (1V/DIV) VOUT2 AT COUT2 = 47µF ( = 2.5V) 0V VOUT1 (DAC = 2V) VOUT3 ( = 1.5V) OUTPUT VOLTAGES (0.5V/DIV) Soft-Start The POR function initiates the soft-start sequence. As soon as POR is released, the linear regulator output voltage VOUT2 (2.5V) quickly ramps up across the output capacitor. The ramp dV/dt is determined by the internal 230mA current limit and the value of the output capacitor. Simultaneously, the voltage on the SS pin rapidly increases to approximately 1V (this minimizes the soft-start interval). Then an internal 11µA current source charges an external capacitor (CSS) on the SS pin to 4V. The PWM error amplifier reference input (+ terminal) and output (COMP pin) is clamped to a level proportional to the SS pin voltage. As the SS pin voltage slews from 1V to 4V, the output clamp generates PHASE pulses of increasing width that charge the output capacitor(s). After this initial stage, the reference input clamp slows the output voltage rate-of-rise and provides a smooth 2-317 0V T0 T1 T2 TIME T3 T4 FIGURE 6. SOFT-START INTERVAL VOUT3 follows a ramp similar to that of the soft-start. The PGOOD signal toggles ‘high’ when all output voltage levels have exceeded their under-voltage levels. See the Soft-Start Interval section under Applications Guidelines for a procedure to determine the soft-start interval. Fault Protection All three outputs are monitored and protected against extreme overload. A sustained overload on any regulator output or an HIP6028 OVER CURRENT LATCH INHIBIT S Q OC1 S R 0.15V + COUNTER - R SS + 4V FAULT LATCH VCC S Q UP - R FAULT POR SD1&3 OV FIGURE 7. FAULT LOGIC - SIMPLIFIED SCHEMATIC Figure 7 shows a simplified schematic of the fault logic. An over-voltage detected on VSEN1 immediately sets the fault latch. A sequence of three over-current fault signals also sets the fault latch. A comparator indicates when CSS is fully charged (UP signal), such that an under-voltage event on either linear output (FB2 or FB3) is ignored until after the soft-start interval (T4 in Figure 6). At startup, this allows VOUT3 to slew up without generating a fault. Cycling the bias input voltage (+12VIN on the VCC pin) off then on resets the counter and the fault latch. Shutting down VOUT1 and VOUT3 also resets the counter and latch. Over-Voltage Protection During operation, a short on the upper PWM MOSFET (Q1) causes VOUT1 to increase. When the output exceeds the over-voltage threshold of 115% (typical) of DACOUT, the over-voltage comparator trips to set the fault latch and turns Q2 on as required in order to regulate VOUT1 to 1.15 x DACOUT. This blows the input fuse and reduces VOUT1. The fault latch raises the FAULT pin close to VCC potential. A separate over-voltage circuit provides protection during the initial application of power. For voltages on the VCC pin below the power-on reset (and above ~4V), VOUT1 is monitored for voltages exceeding 1.26V. Should VSEN1 exceed this level, the lower MOSFET (Q2) is driven on as needed to regulate VOUT1 to 1.26V. Over-Current Protection All outputs are protected against excessive over-currents. The PWM controller uses the upper MOSFET’s onresistance, rDS(ON) to monitor the current for protection against shorted outputs. The linear regulator restricts the current through the integrated power device to 230mA (typically). The linear regulator and the linear controller monitor VOUT2 and VSEN3, respectively, for under-voltage to protect against excessive currents. 2-318 FAULT/RT LUV Figures 8 and 9 illustrate the over-current protection with an overload on VOUT1. The overload is applied at T0 and the current increases through the output inductor (LOUT1). At time T1, the OVER-CURRENT1 comparator trips when the voltage across Q1 (ID • rDS(ON)) exceeds the level programmed by ROCSET. This inhibits all outputs, discharges the soft-start capacitor (CSS) with an 11µA current sink, and increments the counter. CSS recharges at T2 and initiates a soft-start cycle with the error amplifiers clamped by soft-start. With OUT1 still overloaded, the inductor current increases to trip the over-current comparator. Again, this inhibits all outputs, but the soft-start voltage continues increasing to 4V before discharging. The counter increments to 2. The soft-start cycle repeats at T3 and trips the over-current comparator. The SS pin voltage increases to 4V at T4 and the counter increments to 3. This sets the fault latch to disable the converter. The fault is reported on the FAULT pin. INDUCTOR CURRENTSOFT-START over-voltage on the PWM output disables all converters and drives the FAULT pin to VCC. FAULT REPORTED 10V 0V COUNT =1 COUNT =2 COUNT =3 4V 2V 0V OVERLOAD APPLIED 0A T0 T1 T2 T3 T4 TIME FIGURE 8. OVER-CURRENT OPERATION The linear regulator and linear controller monitor the outputs for an under-voltage. Should excessive currents cause VOUT2 or VSEN3 to fall below the linear under-voltage threshold, the LUV signal sets the over-current latch if CSS is fully charged. Blanking the LUV signal during the CSS charge interval allows the linear outputs to build above the undervoltage threshold during normal start-up. Cycling the bias input power off then on resets the counter and the fault latch. Resistor ROCSET programs the over-current trip level for the PWM converter. As shown in Figure 9, the internal 200µA current sink develops a voltage across ROCSET (VSET) that is referenced to VIN. The DRIVE signal enables the overcurrent comparator (OVER-CURRENT1). When the voltage across the upper MOSFET (VDS(ON)) exceeds VSET, the over-current comparator trips to set the over-current latch. Both VSET and VDS are referenced to VIN and a small capacitor across ROCSET helps VOCSET track the variations HIP6028 of VIN due to MOSFET switching. The over-current function will trip at a peak inductor current (IPEAK) determined by: TABLE 1. VOUT1 VOLTAGE PROGRAM VID4 VID3 VID2 VID1 VID0 NOMINAL OUT1 VOLTAGE DACOUT 0 1 1 1 1 1.30 0 1 1 1 0 1.35 0 1 1 0 1 1.40 0 1 1 0 0 1.45 0 1 0 1 1 1.50 + 0 1 0 1 0 1.55 VDS 0 1 0 0 1 1.60 0 1 0 0 0 1.65 0 0 1 1 1 1.70 0 0 1 1 0 1.75 0 0 1 0 1 1.80 0 0 1 0 0 1.85 0 0 0 1 1 1.90 0 0 0 1 0 1.95 0 0 0 0 1 2.00 0 0 0 0 0 2.05 1 1 1 1 1 INHIBIT 2. The minimum IOCSET from the specification table. 1 1 1 1 0 2.1 3. Determine IPEAK for IPEAK > IOUT(MAX) + (∆I)/2, where ∆I is the output inductor ripple current. 1 1 1 0 1 2.2 1 1 1 0 0 2.3 1 1 0 1 1 2.4 1 1 0 1 0 2.5 OUT1 Voltage Program 1 1 0 0 1 2.6 The output voltage of the PWM converter is programmed to discrete levels between 1.3VDC and 3.5VDC . This output is designed to supply the microprocessor core voltage. The voltage identification (VID) pins program an internal voltage reference (DACOUT) through a TTL-compatible 5-bit digitalto-analog converter. The level of DACOUT also sets the PGOOD and OVP thresholds. Table 1 specifies the DACOUT voltage for the different combinations of connections on the VID pins. The VID pins can be left open for a logic 1 input, since they are internally pulled up to +5V by a 10µA (typically) current source. Changing the VID inputs during operation is not recommended. The sudden change in the resulting reference voltage could toggle the PGOOD signal and exercise the over-voltage protection or cause an over-current event. ‘11111’ VID pin combination disables the IC and drives ‘on’ the otherwise open-collector at the PGOOD pin. 1 1 0 0 0 2.7 1 0 1 1 1 2.8 1 0 1 1 0 2.9 1 0 1 0 1 3.0 1 0 1 0 0 3.1 1 0 0 1 1 3.2 1 0 0 1 0 3.3 1 0 0 0 1 3.4 1 0 0 0 0 3.5 I OCSET × R OCSET I PEAK = ---------------------------------------------------r DS ( ON ) OVER-CURRENT TRIP: VDS > VSET VIN = +5V (iD • rDS(ON) > IOCSET • ROCSET) OCSET IOCSET 200µA ROCSET VSET + ID VCC UGATE DRIVE OC1 + PHASE - OVERCURRENT1 PWM VCC LGATE GATE CONTROL VPHASE = VIN - VDS VOCSET = VIN - VSET PGND PIN NAME HIP6028 FIGURE 9. OVER-CURRENT DETECTION The OC trip point varies with MOSFET’s temperature. To avoid over-current tripping in the normal operating load range, determine the ROCSET resistor from the equation above with: 1. The maximum rDS(ON) at the highest junction temperature. For an equation for the output inductor ripple current see the section under component guidelines titled ‘Output Inductor Selection’. 2-319 NOTE: 0 = connected to GND or VSS, 1 = open or connected to 5V through pull-up resistors HIP6028 Application Guidelines Toggling the SD1&3 pin does not affect VOUT2 (2.5V), which remains operational as long as the input voltages are above their POR levels and output current rating is not exceeded. Soft-Start Interval Initially, the soft-start function clamps the error amplifier’s output of the PWM converter. After the output voltage increases to approximately 80% of the set value, the reference input of the error amplifier is clamped to a voltage proportional to the SS pin voltage. The linear controller output follows a similar start-up sequence. The integrated linear regulator’s soft-start is independent of CSS, its ramp-up time being dependent on the 230mA current limit and the size of the output capacitor. The resulting output voltage sequence is shown in Figure 6. The soft-start function controls the output voltage rate of rise to limit the current surge at start-up. The soft-start interval is programmed by the soft-start capacitor, CSS. Programming a faster soft-start interval increases the peak surge current. The peak surge current occurs during the initial output voltage rise to 80% of the set value. Shutdown The PWM output does not switch until the soft-start voltage exceeds the oscillator’s valley voltage. Additionally, the linear controller and PWM’s error amplifiers inputs are clamped to the soft-start voltage. Applying a logic ‘high’ signal on the SD1&3 pin turns off the GTL (VOUT3) and core PWM (VOUT1) regulators and discharges the soft-start capacitor. Releasing, or applying a logic ‘low’ on the SD1&3 pin allows the GTL and core voltages to undergo a soft-start ramp-up to their preset levels. Figure 10 exemplifies such output sequencing resulting from SD1&3 pin toggling (S3 powersaving sleep mode cycling). SD1&3 (5V/DIV) 0V PGOOD (2V/DIV) 0V SOFT-START (2V/DIV) 0V VOUT2 ( = 2.5V) VOUT1 (DAC = 2V) VOUT3 ( = 1.5V) OUTPUT VOLTAGES (0.5V/DIV) 0V TIME FIGURE 10. POWER-SAVING SHUTDOWN SEQUENCE DETAIL 2-320 The ‘11111’ VID code resulting in an INHIBIT, as shown in Table 1, disables the entire IC (all outputs). Layout Considerations MOSFETs switch very fast and efficiently. The speed with which the current transitions from one device to another causes voltage spikes across the interconnecting impedances and parasitic circuit elements. The voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device over-voltage stress. Careful component layout and printed circuit design minimizes the voltage spikes in the converter. Consider, as an example, the turn-off transition of the upper PWM MOSFET. Prior to turn-off, the upper MOSFET is carrying the full load current. During the turn-off, current stops flowing in the upper MOSFET and is picked up by the lower MOSFET (and/or parallel Schottky diode). Any inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide circuit traces minimize the magnitude of voltage spikes. Contact Intersil for evaluation board drawings of the component placement and printed circuit board. There are two sets of critical components in a DC-DC converter using a HIP6028 controller. The power components are the most critical because they switch large amounts of energy. The critical small signal components connect to sensitive nodes or supply critical by-passing current. The power components should be placed first. Locate the input capacitors close to the power switches. Minimize the length of the connections between the input capacitors and the power switches. Locate the output inductor and output capacitors between the MOSFETs and the load. Locate the PWM controller close to the MOSFETs. The critical small signal components include the bypass capacitor for VCC and the soft-start capacitor, CSS. Locate these components close to their connecting pins on the control IC. Minimize any leakage current paths from SS node because the internal current source is only 11µA. HIP6028 VIN +5VIN OSC +3.3VIN CIN +12V CVCC VCC VIN2 VOUT3 COCSET GND DRIVE3 LOUT1 - SS LGATE LOAD COUT2 PHASE CO ESR (PARASITIC) CR1 PGND VE/A ZIN ERROR AMP CSS VOUT2 VOUT ZFB COUT1 LOAD LOAD VOUT2 Q2 DRIVER VOUT1 PHASE HIP6028 LO + ROCSET OCSET Q1 UGATE Q3 ∆ VOSC DRIVER PWM COMP + REFERENCE DETAILED FEEDBACK COMPENSATION KEY ZFB ISLAND ON POWER PLANE LAYER VOUT C2 ISLAND ON CIRCUIT PLANE LAYER VIA CONNECTION TO GROUND PLANE C1 ZIN C3 R2 FIGURE 11. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS R3 R1 COMP A multi-layer printed circuit board is recommended. Figure 11 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the control IC to the MOSFET gate and source should be sized to carry 1A currents. The traces for VOUT2 need only be sized for 0.2A. Locate COUT2 close to the HIP6028 IC. PWM Controller Feedback Compensation Both PWM controllers use voltage-mode control for output regulation. This section highlights the design consideration for a voltage-mode controller. Apply the methods and considerations to both PWM controllers. Figure 12 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage is regulated to the reference voltage level. The reference voltage level is the DAC output voltage for the PWM controller. The error amplifier output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). The modulator transfer function is the small-signal transfer function of VOUT/VE/A. This function is dominated by a DC gain and the output filter, with a double pole break frequency at FLC and a zero at FESR. The DC gain of the modulator is simply the input voltage, VIN, divided by the peak-to-peak oscillator voltage, ∆VOSC . 2-321 + HIP6028 FB REFERENCE FIGURE 12. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN Modulator Break Frequency Equations 1 F LC = ---------------------------------------2π × L O × C O 1 F ESR = ----------------------------------------2π × ESR × C O The compensation network consists of the error amplifier internal to the HIP6028 and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with an acceptable 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The equations below relate the compensation network’s poles, zeros and gain to the components (R1, R2, R3, C1, C2, and C3) in Figure 12. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth 2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC) 3. Place 2ND Zero at Filter’s Double Pole 4. Place 1ST Pole at the ESR Zero 5. Place 2ND Pole at Half the Switching Frequency 6. Check Gain against Error Amplifier’s Open-Loop Gain 7. Estimate Phase Margin - Repeat if Necessary HIP6028 Compensation Break Frequency Equations 1 F P1 = ------------------------------------------------------C1 × C2 2π × R 2 × ---------------------- C1 + C2 1 F P2 = ----------------------------------2π × R 3 × C3 1 F Z1 = ----------------------------------2π × R 2 × C1 1 F Z2 = ------------------------------------------------------2π × ( R1 + R3 ) × C3 Figure 13 shows an asymptotic plot of the DC-DC converter’s gain vs. frequency. The actual modulator gain has a peak due to the high Q factor of the output filter at FLC, which is not shown in Figure 13. Using the above guidelines should yield a compensation gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The closed loop gain is constructed on the log-log graph of Figure 13 by adding the modulator gain (in dB) to the compensation gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. 100 FZ1 FZ2 FP1 FP2 80 OPEN LOOP ERROR AMP GAIN GAIN (dB) 60 40 20 20LOG (R2/R1) 0 20LOG (VIN/∆VOSC) MODULATOR GAIN -20 COMPENSATION GAIN CLOSED LOOP GAIN -40 FLC -60 10 100 1K FESR 10K 100K 1M 10M FREQUENCY (Hz) PWM Output Capacitors Modern microprocessors produce transient load rates above 10A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and ESL (effective series inductance) parameters rather than actual capacitance. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching regulator applications for the bulk capacitors. The bulk capacitor’s ESR determines the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select suitable components. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. For a given transient load magnitude, the output voltage transient response due to the output capacitor characteristics can be approximated by the following equation: dI TRAN V TRAN = ESL × --------------------- + ESR × I TRAN dt FIGURE 13. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN Linear Output Capacitors The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth loop. A stable control loop has a 0dB gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. The output capacitors for the linear regulator and the linear controller provide dynamic load current. The linear controller uses dominant pole compensation integrated in the error amplifier and is insensitive to output capacitor selection. Capacitor, COUT3 should be selected for transient load regulation. Component Selection Guidelines The output capacitor for the linear regulator provides loop stability. The linear 2.5V regulator requires an output capacitor characteristic shown in Figure 14 The upper line plots the 45 phase margin with 150mA load and the lower line is the 45 phase margin limit with a 10mA load. Select a COUT2 capacitor with characteristic between the two limits. Output Capacitor Selection The output capacitors for each output have unique requirements. In general the output capacitors should be selected to meet the dynamic regulation requirements. Additionally, the PWM converters require an output capacitor to filter the current ripple. The linear regulator is internally compensated and requires an output capacitor that meets the stability requirements. The load transient for the microprocessor core requires high quality capacitors to supply the high slew rate (di/dt) current demands. 2-322 Output Inductor Selection The PWM converter requires an output inductor. The output inductor is selected to meet the output voltage ripple requirements and sets the converter’s response time to a load transient. The inductor value determines the converter’s HIP6028 ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: V IN – V OUT V OUT ∆I = -------------------------------- × ---------------V IN FS × LO ∆V OUT = ∆I × ESR Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. Input Capacitor Selection The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use ceramic capacitance for the high frequency decoupling and bulk capacitors to supply the RMS current. Small ceramic capacitors should be placed very close to the upper MOSFET to suppress the voltage induced in the parasitic circuit impedances. 0.7 0.6 0.5 ESR (Ω) 0.4 LE AB ON ST ATI ER OP 0.3 0.2 0.1 10 100 CAPACITANCE (µF) 1000 FIGURE 14. COUT2 OUTPUT CAPACITOR One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the HIP6028 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time interval required to slew the inductor current from an initial current value to the post-transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitors. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: L O × I TRAN t RISE = -------------------------------V IN – V OUT L O × I TRAN t FALL = ------------------------------V OUT where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. With a +5V input source, the worst case response time can be either at the application or removal of load, and dependent upon the output voltage setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. 2-323 For a through hole design, several electrolytic capacitors (Panasonic HFQ series or Nichicon PL series or Sanyo MVGX or equivalent) may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. The TPS series available from AVX, and the 593D series from Sprague are both surge current tested. MOSFET Selection/Considerations The HIP6028 requires 3 power transistors. Two N-channel MOSFETs are used in the synchronous-rectified buck topology of the PWM converter. The linear controller drives a MOSFET or a bipolar NPN as a pass transistor. These components should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. PWM MOSFET Selection and Considerations In high-current PWM applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. These losses are distributed between the upper and lower MOSFETs according to duty factor (see the equations below). The conduction loss is the only component of power dissipation for the lower MOSFET. Only the upper MOSFET has switching losses, since the lower device turns on into near zero voltage. The equations below assume linear voltage-current transitions and do not model power loss due to the reverserecovery of the lower MOSFET’s body diode. The gatecharge losses are proportional to the switching frequency (FS) and are dissipated by the HIP6028, thus not contributing to the MOSFETs’ temperature rise. However, large gate charge increases the switching interval, tSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by HIP6028 calculating the temperature rise according to package thermal resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. Linear Controller Pass Transistor Selection The main criteria for selection of a pass transistor for the linear regulator is package selection for efficient removal of heat. The power dissipated in a linear regulator is: 2 I O × r DS ( ON ) × V OUT I O × V IN × t SW × F S P UPPER = ------------------------------------------------------------ + ---------------------------------------------------V IN 2 2 I O × r DS ( ON ) × ( V IN – V OUT ) P LOWER = --------------------------------------------------------------------------------V IN Select a package and heatsink that maintains the junction temperature below the maximum rating while operating at the highest expected ambient temperature. The rDS(ON) is different for the two previous equations even if the type device is used for both. This is because the gate drive applied to the upper MOSFET is different than the lower MOSFET. Figure 15 shows the gate drive where the upper gate-to-source voltage is approximately VCC less the input supply. For +5V main power and +12VDC for the bias, the gate-to-source voltage of Q1 is 7V. The lower gate drive voltage is +12VDC. A logic-level MOSFET is a good choice for Q1 and a logic-level MOSFET can be used for Q2 if its absolute gate-to-source voltage rating exceeds the maximum voltage applied to VCC . +12V +5V OR LESS VCC HIP6028 UGATE Q1 PHASE - + LGATE PGND NOTE: VGS ≈ VCC -5V Q2 CR1 NOTE: VGS ≈ VCC GND FIGURE 15. OUTPUT GATE DRIVERS Rectifier CR1 is a clamp that catches the negative inductor voltage swing during the dead time between the turn off of the lower MOSFET and the turn on of the upper MOSFET. The diode must be a Schottky type to prevent the lossy parasitic MOSFET body diode from conducting. It is acceptable to omit the diode and let the body diode of the lower MOSFET clamp the negative inductor swing, but efficiency might drop one or two percent as a result. The diode's rated reverse breakdown voltage must be greater than twice the maximum input voltage. 2-324 P LINEAR = I O × ( V IN – V OUT ) Additionally, if selecting a bipolar NPN transistor, insure the gain (hfe) at the minimum operating temperature and given collector-to-emitter voltage is sufficiently high as to deliver the worst-case steady state current required by the GTL output, when the transistor is driven with the minimum guaranteed DRIVE3 output current. For example, operating at ‘T’ junction temperature, 3.3V input, and 1.5V output (VCE = 1.8V), the NPN’s gain should satisfy the following equation: I GTL ( steady – state ) h fe > ----------------------------------------------------------I DRIVE3 ( min ) HIP6028 HIP6028 DC-DC Converter Application Circuit Figure 16 shows an application circuit of a power supply for a microprocessor computer system. The power supply provides the microprocessor core voltage (VOUT1), the GTL bus voltage (VOUT3) and clock generator voltage (VOUT2) from +3.3VDC, +5VDC and +12VDC. For detailed information +12VIN on the circuit, including a Bill-of-Materials and circuit board description, see Application Note AN9833. Also see Intersil’s web page (http://www.intersil.com) or Intersil AnswerFAX (407-724-7800) document # 99833 for the latest information. L1 F1 +5VIN 1µH 15A C1-7 + 7x680µF GND C14, 15 2x1µF C16 1µF C18 VCC 1000pF R2 1 VIN2 +3.3VIN 20 OCSET 12 1K POWERGOOD + 7 C19 1000µF 24 23 Q3 RFD3055 DRIVE3 VOUT3 VSEN3 R25 1K L3 Q1 HUF76143 UGATE PHASE VOUT1 (1.3 to 3.5V) 3.5µH 15 22 16 (1.5V) + PGOOD 21 C43 1000µF LGATE C24-30 + 7x1000µF Q2 HUF76143 PGND R4 4.99K VSEN1 19 HIP6028 VOUT2 VOUT2 13 18 FB (2.5V) + C47 270µF 17 COMP R8 C40 2.21K 0.68µF C41 10pF C42 SD1&3 S3 (ACPI CONTROLLER) VID0 VID0 VID1 VID1 VID2 VID2 VID3 VID3 VID4 VID4 R10 2200pF 160K 11 8 6 5 10 4 9 750K FAULT RT SS 3 2 R9 C48 0.039µF 14 GND FIGURE 16. APPLICATION CIRCUIT All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification. Intersil semiconductor products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see web site http://www.intersil.com 2-325 HIP6028 2-326