DATASHEET

ISL6253
®
Data Sheet
August 2004
Highly Integrated Battery Charger for
Notebook Computers
Features
• ±0.5% Voltage Accuracy
The ISL6253 is a highly integrated battery charger controller
for Li-Ion/Li-Polymer batteries. High Efficiency is achieved by
a synchronous buck topology and the use of a MOSFET,
instead of a diode, for selecting power from the adapter or
battery. The low side MOSFET emulates a diode at light
loads to improve the light load efficiency and prevent system
bus boosting.
The constant output voltage can be selected for 2, 3 or 4
series Li-Ion cells with 0.5% accuracy over temperature. It
can also be programmed between 4.2V +5% per cell and
4.2V -5% per cell to optimize battery capacity. When
supplying the load and battery charger simultaneously, the
input current limit for the AC adapter is programmable to
within 3% accuracy to avoid overloading the AC adapter,
and to allow the system to make efficient use of available
adapter power for charging. It also has programmable
charging current with 4% accuracy. The ISL6253 provides
outputs that are used to monitor the current drawn from the
AC adapter, and to monitor for the presence of an AC
adapter. The ISL6253 automatically transitions from
regulating current to regulating voltage. A conditioning
charge feature provides approximately 10% of full scale
charge current to safely charge deeply discharged lithiumion (LI+) battery packs when the battery voltage is below
3.0V/cell.
ISL6253 has a feature of automatic power source selection
by switching to the battery when the AC adapter is removed
or switching to the AC adapter when the AC adapter is
available. It also supports aircraft power applications while
powering the system and not charging the battery.
Ordering Information
PART #
TEMP.
RANGE (°C)
PACKAGE
FN9117.2
PKG.
DWG. #
ISL6253HRZ
(Note 1)
-10 to 100
28 Ld 5x5 QFN
(Pb-free)
L28.5x5
ISL6253HAZ
(Note 1)
-10 to 100
28 Ld QSOP
(Pb-free)
M28.15
• ±3% Input Current Limit Accuracy
• ±4% Accurate Battery Charge Current Limit
• Programmable Charge Limit Current, Adaptor Current
Limit and Charge Voltage
• Fixed 300kHz PWM Synchronous Buck Controller with
Diode Emulation at Light Load
• Output for Current Drawn from the AC Adapter
• AC Adapter Present Indicator
• Fast Input Current Limit Response
• Input Voltage Range 7V to 25V
• Up to 17.6V Battery-Voltage Set Point
• Trickle Charge Mode When Battery Voltage is below
3.0V/Cell
• Supports 2, 3 and 4 Cell Battery Packs
• Control Adapter Power Source Select MOSFET
• Thermal Shutdown
• Aircraft Power Capable
• DC Adapter Present Indicator
• Battery Discharge MOSFET Control
• Less than 10µA Battery Leakage Current
• QFN Package
- Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat
No Leads - Package Outline
- Near Chip Scale Package footprint, which improves
PCB efficiency and has a thinner profile
• Pb-free Available
Applications
• Notebook, Desknote and Sub-notebook Computers
• Personal Digital Assistants
NOTES:
1. Intersil Pb-free products employ special Pb-free material sets;
molding compounds/die attach materials and 100% matte tin
plate termination finish, which is compatible with both SnPb and
Pb-free soldering operations. Intersil Pb-free products are MSL
classified at Pb-free peak reflow temperatures that meet or
exceed the Pb-free requirements of IPC/JEDEC J Std-020B.
2. Add “-T” for Tape and Reel.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2004. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6253
Pinouts
EN
ACSET
VDD
DCIN
DCPRN
ACPRN
CSON
ISL6253 (28 LD QSOP)
TOP VIEW
DCSET
ISL6253 (28 LD QFN)
TOP VIEW
DCIN
28
27
26
25
24
23
22
ACSET
21
1
CSOP
CELLS
2
20
CSIN
ICOMP
3
19
CSIP
VCOMP
ICM
18
4
17
5
SGATE
BGATE
UGATE
8
9
10
11
12
13
14
BOOT
15
VDDP
7
LGATE
CHLIM
PGND
PHASE
GND
16
VADJ
6
ACLIM
VREF
2
1
28 DCPRN
VDD
2
27 ACPRN
3
26 CSON
DCSET
4
25 CSOP
EN
5
24 CSIN
CELLS
6
23 CSIP
ICOMP
7
22 SGATE
VCOMP
8
21 BGATE
ICM
9
20 PHASE
VREF
10
19 UGATE
CHLIM
11
18 BOOT
ACLIM
12
17 VDDP
VADJ
13
16 LGATE
GND
14
15 PGND
ISL6253
Absolute Maximum Ratings
Thermal Information
DCIN, CSIP, DCPRN, ACPRN, CSON to GND . . . . . . -0.3V to +28V
CSIP-CSIN, CSOP-CSON . . . . . . . . . . . . . . . . . . . . . -0.3V to +0.3V
CSIP-SGATE, CSIP-BGATE . . . . . . . . . . . . . . . . . . . . . -0.3V to 16V
PHASE to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -7V to 28V
BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +35V
BOOT-PHASE, VDD-GND, VDDP-PGND . . . . . . . . . . . . -0.3V to 7V
ICM, ICOMP, VCOMP . . . . . . . . . . . . . . . . . . . . -0.3V to VDD+0.3V
ACSET and DCSET to GND (Note 3) . . . . . . . . -0.8V to VDD+0.3V
VDDP, ACLIM, CHLIM, VREF, CELLS. . . . . . . . -0.3V to VDD+0.3V
EN, VADJ, PGND to GND . . . . . . . . . . . . . . . . . -0.3V to VDD+0.3V
UGATE. . . . . . . . . . . . . . . . . . . . . . . . . PHASE-0.3V to BOOT+0.3V
LGATE . . . . . . . . . . . . . . . . . . . . . . . . . . PGND-0.3V to VDDP+0.3V
Thermal Resistance
θJA (°C/W) θJC (°C/W)
QFN Package (Notes 4, 6). . . . . . . . . .
39
9.5
QSOP Package (Note 5) . . . . . . . . . . .
80
NA
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Level 1
Junction Temperature Range. . . . . . . . . . . . . . . . . .-10°C to +150°C
Operating Temperature Range . . . . . . . . . . . . . . . .-10°C to +100°C
Storage Temperature. . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Lead Temperature (soldering, 10s).. . . . . . . . . . . . . . . . . . . . +300°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress ratings only and operation of the
device at these or any other conditions above those indicated in the operational sections of the specifications is not implied.
NOTES:
3. When the voltage across ACSET and DCSET is below 0V, the current through ACSET and DCSET should be limited less than 1mA.
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
5. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
6. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
DCIN = CSIP = CSIN = 18V, CSOP = CSON = 12V, ACSET = DCSET = 1.5V, VREF = ACLIM = CHLIM,
VADJ = Floating, EN = VDD = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDD = 1µF IVDD = 0mA,
TA = -10°C to +100°C, TJ ≤ 125°C, unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
25
V
SUPPLY AND BIAS REGULATOR
DCIN Input Voltage Range
7
DCIN Quiescent Current
EN = VDD, 7V < DCIN < 25V
1.6
4
mA
DCIN Quiescent Current in Shutdown mode
EN = 0, 7V < DCIN < 25V
1.3
3
mA
Battery Leakage Current (Note 5)
DCIN = 0, no load
2
10
µA
VDD Output Voltage/Regulation
7V < DCIN < 25 V, 0 < IVDD < 30mA
4.925
5.075
5.225
V
VDD Undervoltage Lockout Trip Point
Rising
4.2
4.4
4.6
V
Hysteresis
100
250
400
mV
V
O < IVREF < 300µA
Reference Output Voltage VREF
Battery Charge Voltage Accuracy
2.365
2.390
2.415
CSON = 16.8V, CELLS = VDD, VADJ = Float
-0.5
0
0.5
CSON = 12.6V, CELLS = GND, VADJ = Float
-0.55
0
0.55
CSON = 8.4V, CELLS = FLOAT, VADJ = Float
-0.55
0
0.55
CSON = 17.64V, CELLS = VDD, VADJ = VREF
-0.6
0
0.6
CSON = 13.23V, CELLS = GND, VADJ = VREF
-0.6
0
0.6
CSON = 8.82V, CELLS = FLOAT, VADJ = VREF
-0.6
0
0.6
CSON = 15.96V, CELLS = VDD, VADJ = GND
-0.6
0
0.6
CSON = 11.97V, CELLS = GND, VADJ = GND
-0.6
0
0.6
CSON = 7.98V, CELLS = FLOAT, VADJ = GND
-0.6
0
0.6
1.235
1.26
1.285
V
2
3.4
4.8
µA
%
TRIP POINTS
ACSET Threshold
ACSET Input Bias Current Hysteresis
ACSET Input Bias Current
ACSET > 1.26V
2
3.4
4.8
µA
ACSET Input Bias Current
ACSET < 1.26V
-1
0
1
µA
1.235
1.26
1.285
V
DCSET Threshold
3
ISL6253
Electrical Specifications
DCIN = CSIP = CSIN = 18V, CSOP = CSON = 12V, ACSET = DCSET = 1.5V, VREF = ACLIM = CHLIM,
VADJ = Floating, EN = VDD = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDD = 1µF IVDD = 0mA,
TA = -10°C to +100°C, TJ ≤ 125°C, unless otherwise noted. (Continued)
PARAMETER
TEST CONDITIONS
DCSET Input Bias Current Hysteresis
MIN
TYP
MAX
UNITS
2
3.4
4.8
µA
DCSET Input Bias Current
DCSET ≥ 1.26V
2
3.4
4.8
µA
DCSET Input Bias Current
DCSET < 1.26V
-1
0
1
µA
OSCILLATOR
Frequency
240
300
360
kHz
CSIP = 18V
1.4
1.55
1.7
V
CSIP = 11V
0.8
0.9
1.0
V
Maximum Duty Cycle
300kHz
97
99
99.6
%
UGATE Pull-up Resistance
BOOT-PHASE = 5V, 500mA source current
1.8
3.0
Ω
1.8
Ω
PWM Ramp Voltage (peak-peak)
SYNCHRONOUS BUCK REGULATOR
UGATE Source Current
BOOT-PHASE = 5V, BOOT-UGATE = 2.5V
1.0
UGATE Pull-down Resistance
BOOT-PHASE = 5V, 500mA sink current
0.9
UGATE Sink Current
BOOT-PHASE = 5V, UGATE-PHASE = 2.5V
1.8
LGATE Pull-up Resistance
VDDP-PGND = 5V, 500mA source current
1.8
LGATE Source Current
VDDP-PGND = 5V, VDDP-LGATE = 2.5V
1.0
LGATE Pull-down Resistance
VDDP-PGND = 5V, 500mA sink current
0.9
LGATE Sink Current
VDDP-PGND = 5V, LGATE = 2.5V
1.8
A
A
3.0
Ω
1.8
Ω
A
A
CHARGING CURRENT SENSING AMPLIFIER
Input Common Mode Range
0
Input Offset Voltage
Guarantee by design
-3
0
18
V
3
mV
Input Bias Current at CSOP
0 < CSOP < 18V
10
20
µA
Input Bias Current at CSON
0 < CSON < 18V
300
425
µA
CSOP to CSON Input Full Scale Sense Voltage
CHLIM = 3.3V
123
127
131
mV
CHLIM = VREF
96
100
104
mV
CHLIM = FLOAT
61
65
69
mV
CHLIM = GND
26
30
34
mV
CSOP to CSON Input Full Scale Sense Voltage in Trickle CHLIM = 3.3V
Charge Mode
CHLIM = VREF
Trickle Charge Threshold Voltage
CHLIM Input Bias Current
6
8.9
12.0
mV
4.0
7.5
11.0
mV
CHLIM = FLOAT
2.5
5.6
9.0
mV
CHLIM = GND
1.5
3.7
7.0
mV
CSON Rising
3.0
3.1
3.2
V/CELL
Hysteresis
20
100
180
mV/CELL
CHLIM = GND or VREF
-25
25
µA
7
25
V
-2
2
mV
ADAPTER CURRENT SENSING AMPLIFIER
Input Common Mode Range
Input Offset Voltage
Guarantee by design
Input Bias Current at CSIP
0 < CSIP < DCIN
325
475
µA
Input Bias Current at CSIN
0 < CSIN < DCIN
1
10
µA
ADAPTER CURRENT LIMIT THRESHOLD
CSIP to CSIN Input Full Scale Sense Voltage
4
ACLIM = VREF
100
103
106
mV
ACLIM = Float
75
78
81
mV
ACLIM = GND
50
53
56
mV
ISL6253
Electrical Specifications
DCIN = CSIP = CSIN = 18V, CSOP = CSON = 12V, ACSET = DCSET = 1.5V, VREF = ACLIM = CHLIM,
VADJ = Floating, EN = VDD = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDD = 1µF IVDD = 0mA,
TA = -10°C to +100°C, TJ ≤ 125°C, unless otherwise noted. (Continued)
PARAMETER
ACLIM Input Bias Current
TEST CONDITIONS
MIN
ACLIM = GND, or VREF
-25
CELLS = VDD
20
TYP
MAX
UNITS
25
µA
40
µA/V
4
V
VOLTAGE REGULATION ERROR AMPLIFIER
Error Amplifier Trans-conductance from CSON to
VCOMP
VCOMP Voltage Range
30
0.3
CURRENT REGULATION ERROR AMPLIFIER
Charging Current Error Amplifier Trans conductance
µA/V
50
Adapter Current Error Amplifier Trans conductance
µA/V
50
ICOMP Voltage Range
0.3
4
V
0.5
V
VREF
+ 0.3
V
BATTERY CELL SELECTOR
CELLS Input Low Voltage
CELLS Input Float Voltage
CELLS = Float
CELLS Input High Current
VREF
-0.3
VREF
VDD0.5
V
MOSFET DRIVER
BGATE Pull-up Current
CSIP-BGATE = 3V
1
1.4
2
mA
BGATE Pull-down Current
CSIP-BGATE = 5V
3.25
5.0
7
mA
8.0
10
12
V
0
0.1
V
-200
0
200
mV
CSIP-BGATE Voltage High
CSIP-BGATE Voltage Low
DCIN-CSON Threshold for CSIP-BGATE Going High
DCIN = 12V, CSON Rising
DCIN-CSON Threshold for CSIP-BGATE Going Low
DCIN = 12V, CSON Falling
DCIN-CSON Threshold Hysteresis
0
300
500
mV
250
300
400
mV
SGATE Pull-up Current
CSIP-SGATE = 3V
1.0
1.4
2.0
mA
SGATE Pull-down Current
CSIP-SGATE = 2.5V
40
150
350
µA
5
8.5
15
V
0
0.1
V
CSIP-SGATE Voltage High
CSIP-SGATE Voltage Low
CSIP-CSIN Threshold for CSIP-SGATE Going High
3
8.5
15
mV
CSIP-CSIN Threshold Hysteresis
1
2
6
mV
0.8
V
13
mA
1
µA
13
mA
1
µA
2.085
V
LOGIC INTERFACE
EN Low Level Input Voltage
EN High Level Input Voltage
2.0
ACPRN Sink Current
ACPRN = 0.4V
3
ACPRN Leakage Current
ACPRN = 20V
-1
DCPRN Sink Current
DCPRN = 0.4V
3
DCPRN Leakage Current
DCPRN = 20V
-1
ICM Output Voltage
Iload = 0 to 100µA, CSIP-CSIN = 103mV
Thermal Shutdown Temperature
Rising
Thermal Shutdown Temperature Hysteresis
1.885
V
8
8
1.980
150
°C
20
°C
NOTE:
7. This is sum of currents in these pins (CSIP, CSIN, BOOT, UGATE, PHASE, CSOP and CSON) tied together to 18V. EN, ACSET, DCSET, VADJ,
CELLS, ACLIM, CHLIM, VDD, DCIN = 0.
5
ISL6253
Typical Operating Performance
0.01
0.1
VDCIN = 20V
VDD = 5.075V
EN = 0
0
VDD =5.075V
IVDD = 0mA
0
-0.1
VDD ACCURACY (%)
VDD LOAD REGULATION ACCURACY (%)
Circuit of Figure 18, VDCIN = 20V, 4S2P Li-Ion Battery, TA = 25°C, unless otherwise noted.
-0.2
-0.3
-0.4
-0.01
-0.02
-0.5
-0.03
-0.6
-0.7
0
5
10
15
20
LOAD CURRENT (mA)
25
-0.04
30
5
5.0
17.0
4.0
16.0
3.0
15.0
2.0
14.0
1.0
13.0
LOAD
CURRENT
5A/DIV
ADAPTER
CURRENT
5A/DIV
CHARGE
CURRENT
2A/DIV
BATTERY
VOLTAGE
2V/DIV
t: 1ms/DIV, CHLIM = ACLIM = VREF
LOAD STEP: 0A TO 4A
CHARGE CURRENT = 3A
AC ADAPTER CURRENT LIMIT = 5A
12.0
0.0
0
26
50
75
25
FIGURE 2. VDD LINE REGULATION
BATTERY VOLTAGE (V)
CHARGE CURRENT (A)
FIGURE 1. VDD LOAD REGULATION
10
15
20
INPUT VOLTAGE VDCIN (V)
99
TIME (MINUTE)
FIGURE 3. BATTERY CHARGE V-I CHARACTERISTICS
FIGURE 4. LOAD TRANSIENT RESPONSE
INDUCTOR
CURRENT
1A/DIV
5
VDCIN
10V/DIV
0
ICM ACCURACY (%)
EN=0
BATTERY
VOLTAGE
10V/DIV
CHARGE
CURRENT
1A/DIV
-5
-10
-15
t: 10ms/DIV
CHARGE CURRENT: 2.6A
INPUT VOLTAGE STEP: 18.0V TO 25V
LOAD CURRENT: 1A
FIGURE 5. LINE TRANSIENT RESPONSE
6
-20
0
1
2
3
AC ADAPTER CURRENT (A)
4
5
FIGURE 6. ICM ACCURACY vs AC ADAPTER CURRENT
ISL6253
Typical Operating Performance (Continued)
Circuit of Figure 18, VDCIN = 20V, 4S2P Li-Ion Battery, TA = 25°C, unless otherwise noted.
EN
5V/DIV
SYSTEM BUS
VOLTAGE
5V/DIV
INDUCTOR
CURRENT
2A/DIV
BGATE-CSIP
2V/DIV
BGATE-CSIP
2V/div
SGATE-CSIP
SGATE-CSIP
2V/DIV
2V/DIV
2V/div
INDUCTOR
CURRENT
2A/DIV
BATTERY
VOLTAGE
2V/DIV
t: 0.5ms/DIV
t: 0.5ms/DIV
FIGURE 7. CHARGE ENABLE AND SHUTDOWN
FIGURE 8. AC ADAPTER INSERTION
LOAD CURRENT = 2A
BGATE-CSIP
2V/DIV
INDUCTOR
CURRENT
2A/DIV
SYSTEM BUS
VOLTAGE
5V/DIV
BATTERY
VOLTAGE
10V/DIV
SGATE-CSIP
2V/DIV
ICOMP
INDUCTOR
CURRENT
2A/DIV
VCOMP
2V/DIV
t: 1ms/DIV
t: 5µs/DIV
FIGURE 9. AC ADAPTER REMOVAL
FIGURE 10. BATTERY INSERTION
BATTERY
VOLTAGE
2V/DIV
VPHASE
10V/DIV
VCSON
10V/DIV
INDUCTOR
CURRENT
1A/DIV
INDUCTOR
CURRENT
2A/DIV
CHARGE
CURRENT
1A/DIV
t: 20µs/DIV
FIGURE 11. BATTERY REMOVAL
7
t: 0.1ms/DIV
FIGURE 12. CHARGE MODE TRANSITION FROM TRICKLE
MODE TO CONSTANT CURRENT MODE
ISL6253
Typical Operating Performance (Continued)
Circuit of Figure 18, VDCIN = 20V, 4S2P Li-Ion Battery, TA = 25°C, unless otherwise noted.
VDCIN = 20V
VDCIN=20V
VCSON = 16.8V
VCSON=16.8V
PHASE
10V/DIV
PHASE
10V/DIV
INDUCTOR
CURRENT
0.5A/DIV
LGATE
LGATE
UGATE
UGATE
UGATE
LGATE
2V/DIV
CHARGE
CURRENT
0.2A/DIV
t: 100ns/DIV
1µs/DIV
FIGURE 13. SWITCHING WAVEFORMS AT CONSTANT
CHARGE CURRENT MODE
FIGURE 14. SWITCHING WAVEFORMS AT TRICKLE CHARGE
MODE
1
EFFICIENCY (%)
0.96
VCSON = 12.6V
3 CELLS
0.92
VCSON = 8.4V
2 CELLS
VCSON = 16.8V
4 CELLS
0.88
0.84
0.8
0.76
0
0.5
1
1.5
2
2.5
3
3.5
4
CHARGE CURRENT (A)
FIGURE 15. EFFICIENCY vs CHARGE CURRENT
Functional Pin Descriptions
BOOT
BGATE
Connect BOOT to a 0.1µF ceramic capacitor to the PHASE
pin and connect to the cathode of the bootstrap Schottky
diode.
UGATE is the high side MOSFET gate drive output.
BGATE power source select output. This pin drives an
external P-channel MOSFET used to switch the battery as
the system power source. When the voltage at the CSON
pin is higher than the AC adapter output voltage at DCIN,
BGATE is driven to low and selects the battery as the power
source.
SGATE
LGATE
SGATE is the AC adapter power source select output. The
SGATE pin drives an external P-MOSFET used to switch to
AC adapter as the system power source.
LGATE is the low side MOSFET gate drive output; swing
between PGND and VDDP.
UGATE
8
ISL6253
PHASE
PGND
The Phase connection pin connects to the high side
MOSFET source, output inductor, and low side MOSFET
drain.
PGND is the power ground. Connect PGND to the source of
the low side MOSFET for the low side MOSFET gate driver.
CSOP/CSON
VDD is an internal LDO output to supply the IC analog
circuit. Connect a 1µF ceramic capacitor to ground.
CSOP/CSON are the battery charging current sensing
positive/negative inputs. The differential voltage across
CSOP and CSON is used to sense the battery charging
current, and is compared with the charging current limit
threshold to regulate the charging current. The CSON pin is
also used as the battery feedback voltage to perform voltage
regulation.
CSIP/CSIN
CSIP/CSIN are the AC adapter current sensing positive/
negative inputs. The differential voltage across CSIP and
CSIN is used to sense the AC adapter current, and is
compared with the AC adapter current limit to regulate the
AC adapter current.
VDD
VDDP
VDDP is the supply voltage for the MOSFET gate driver.
Connect a 4.7Ω resistor to VDD and a 1µF ceramic capacitor
to power ground.
ICOMP
ICOMP is a current loop error amplifier output. Connect a
ceramic capacitor to ground.
VCOMP
VCOMP is a voltage loop amplifier output. Connect a
ceramic capacitor in series with a resistor to ground.
CELLS
GND
This pin is used to select the battery voltage. CELLS = VDD
for a 4S battery, CELLS = GND for a 3S battery pack, and
CELLS = Float for a 2S battery pack.
GND is an analog ground.
DCIN
The DCIN pin is the input of the internal 5V LDO. Connect it
to the AC adapter output. Connect DCIN to a 0.1µF ceramic
capacitor.
VADJ
ACSET is an AC adapter detection input. Connect a resistor
divider to an AC adapter.
VADJ adjusts battery regulation voltage. VADJ = VREF for
4.2V +5%/cell; VADJ = Floating for 4.2V/cell; VADJ = GND
for 4.2V -5%/cell. Connect to a resistor divider from VREF to
program the desired battery cell voltage between 4.2V -5%
and 4.2V +5%.
ACPRN
CHLIM
ACPRN is an AC adapter present open drain output.
ACPRN is active low when ACSET is higher than 1.26V; and
active high when ACSET is lower than 1.26V.
CHLIM is the battery charge current limit set pin.
CHLIM = VREF for 100mV, CHLIM = Floating for 65mV;
CHLIM = GND for 30mV. Connect 3.3V for 127mV. Connect
a resistor divider to program the charge current limit
threshold between 30mV and 127mV.
ACSET
DCSET
DCSET is a lower voltage adapter detection input (like
aircraft power 15V). This allows power to the system, but
power is not used to charge the battery.
DCPRN
DCPRN is a DC adapter present open drain output. DCPRN
is active low when DCSET is higher than 1.26V; and active
high when DCSET is lower than 1.26V.
EN
EN is the Charge Enable input. Connecting EN to high
enables the charge control function, connecting EN to low
disables charging functions.
ICM
ICM is the adapter current output. The output of this pin
produces a voltage proportional to the adapter current.
9
ACLIM
ACLIM is the adapter current limit set pin. ACLIM = VREF for
103mV, ACLIM = Floating for 78mV, and ACLIM = GND for
53mV. Connect a resistor divider from VREF to program the
adapter current limit threshold between 53mV and 103mV.
VREF
VREF is a reference output pin. Do not connect a decoupling
capacitor.
ISL6253
DCIN
SGATE
CSIP
CSIN
DCSET
DCPRN
+
VDDP
--
+
1.26V
CA1
ICM
DCIN
+
CSON
ACSET
+
ACPRN
-
BIAS
Bias
REGULATOR
Regulator
1.26V
BGATE
-
VDD
gm3
ADAPTER
CURRENT LIMIT SET
ACLIM
BOOT
+
MIN.
CURRENT
BUFFER
ICOMP
2.1V
UGATE
MIN
VOLTAGE
BUFFER
+
gm1
-
PHASE
VDDP
-
+
VCOMP
0.25V
LGATE
PWM
CA2
+
VOLTAGE
VADJ
PGND
SELECTOR
3.1V/CELL
gm2
UV
+
-
+
--
CELLS
VCA2
Charge
CHARGE
Current
CURRENT
Set SET
Reference
REFERENCE
GND
CHLIM
FIGURE 16. BLOCK DIAGRAM
10
CSON
+
CSOP
CA2
VDD
VREF
-
EN
ISL6253
INPUT: 7V TO 25V
Q3
C9
0.1µF
D3
R8
130K
1%
CSON
SGATE
DCIN
R9
10.2K
1%
ACSET
CSIP
CSIP
ISL6253
ISL6253
C3
0.1µF
R2
20mΩ
VDDP
C12
1µF
R3: 18Ω
VDD
R5
100K
DIGITAL
INPUT
C10
1µF
OUTPUT
BOOT
BOOT
ACPRN
UGATE
UGATE
CHLIM
PHASE
PHASE
EN
LGATE
LGATE
ICM
5A INPUT
C13
CURRENT LIMIT
0.1
0.1µF
C7
6.8nF
ACLIM
VCOMP
C6
10nF
R10, R11, R12
10K
FLOATING
4.2V/CELL
PGND
PGND
CSOP
CSOP
C4
1µF
VREF
ICOMP
SCL
SDL
A/D INPUT
GND
L
15µH
R1
25mΩ
25m
R4
2.2Ω
BAT+
CSON
CSON
4 CELLS
CELLS
CELLS
VDD
C11
22µF
BATTERY
PACK
GND
GND
VADJ
DCSET
DCSET
FIGURE 17. TYPICAL APPLICATION CIRCUIT 1
11
D1
OPTIONAL
Q2
A/D INPUT
AVDD/VREF
Q1
C5
0.1µF
R13: 100Ω
R6
10K
C2
10µF
C1:10µF
VDDP
D2
D/A OUTPUT
HOST
LOAD
CSIN
CSIN
R7
4.7
4.7Ω
VCC
SYSTEM
SCL
SDL
TEMP
BAT-
ISL6253
AC ADAPTER
Q5
R13
100K
1%
R8
130K
1%
R14
10.5K
1%
R9
10.2K
1%
C9
0.1µF CSON
VDD
DCIN
Q3
SGATE
SGATE
ACSET
CSIP
CSIP
DCSET
C8: 1µF
VCC
R5, R14
100K
R7
4.7Ω
C10: 1µF
ISL6253
ISL6253
VDDP
CSIN
CSIN
C3
0.1µF
R2
20mΩ
R3: 18Ω
VDD
ACPRN
DIGITAL
INPUT
DCPRN
CHLIM
EN
OUTPUT
R13: 100Ω
ICM
A/D INPUT
C12
0.1µF
5A INPUT
CURRENT LIMIT
HOST
ACLIM
VREF
C7
6.8nF
ICOMP
R10,R11
R12:10k
AVDD/VREF
VCOMP
R6
10k
C6
10nF
C2
10µF
BGATE
BGATE
VDDP
DIGITAL
INPUT
D/A OUTPUT
SYSTEM LOAD
C1:10µF
BOOT
BOOT
Q4
D2
UGATE
PHASE
Q1
C5
0.1µF
Q2
D1
Optional
LGATE
L
15µH
PGND
CSOP
C4
1µF
R4
2.2Ω
R1
25mΩ
BAT+
CSON
CELLS
3 CELLS
GND
VADJ
C11
22µF
BATTERY
PACK
FLOAT
4.2V/CELL
SCL
SDL
A/D INPUT
GND
SCL
SDL
TEMP
BAT-
FIGURE 18. ISL6253 CONTROLLED TYPICAL APPLICATION 2
12
ISL6253
Theory of Operation
Introduction
The ISL6253 includes all of the functions necessary to
charge 2 to 4 cell Li-Ion and Li-polymer batteries. A high
efficiency synchronous buck converter is used to control the
charging voltage and charging current up to 10 amps. The
ISL6253 has input current limiting and analog inputs for
setting the charge current and charge voltage; CHLIM inputs
are used to control charge current and VADJ inputs are used
to control charge voltage.
The ISL6253 safely conditions over-discharged battery cells
with a percentage of full charge current until the battery
voltage exceeds 3.1V × number of series connected cells.
When the battery voltage exceeds 3.1V × number of series
connected cells, the ISL6253 charges the battery with
constant charge current, set by CHLIM input, until the battery
voltage rises to a programmed charge voltage set by VADJ
input; then the charger begins to operate at constant voltage
charge mode. The charger drives an adapter isolation
p-channel MOSFET to efficiently switch in the adapter
supply.
ISL6253 is a complete power source selection controller for
single battery systems and also aircraft power applications.
ISL6253 drives a battery selector p-channel MOSFET to
efficiently select between a single battery and the adapter. It
controls the battery discharging MOSFET and switches to
the battery when the AC adapter is removed, or, switches to
the AC adapter when the AC adapter is inserted for a single
battery system. The EN input allows shutdown of the charger
from a micro-controller. The amount of adapter current is
reported on the ICM output. Figure 16 shows the IC
functional block diagram.
The synchronous buck converter uses external N-channel
MOSFETs to convert the input voltage to the required
charging current and charging voltage. Figure 17 shows the
ISL6253 typical application circuit 1 without power source
selection function. The typical application circuit 2 shown in
Figure 18 has automatic power source selection functionality
and supports aircraft power applications. The voltage at
CHLIM and the value of R1 sets the charging current. The
DC-DC converter generates the control signals to drive two
external N-channel MOSFETs to regulate the voltage and
current set by the ACLIM, CHLIM, VADJ and CELLS inputs.
The ISL6253 features a voltage regulation loop (VCOMP)
and two current regulation loops (ICOMP). The VCOMP
voltage regulation loop monitors CSON to ensure that its
voltage never exceeds the voltage set by VADJ. The ICOMP
current regulation loops regulate the battery charging current
delivered to the battery to ensure that it never exceeds the
charging current limit set by CHLIM; and the ICOMP current
regulation loops regulate the input current drawn from the
AC adapter to ensure that it never exceeds the input current
13
limit set by ACLIM, and to prevent a system crash and AC
adapter overload.
PWM Control
The ISL6253 employs a fixed frequency PWM current mode
control architecture with a feed forward function. The feedforward function maintains a constant modulator gain of 11
to achieve fast line regulation as the buck input voltage
changes. When the battery charge voltage approaches the
input voltage, the DC-DC converter operates in dropout
mode, where there is a timer to prevent the frequency from
dropping into the audible frequency range. It can achieve a
maximum duty cycle of up to 99.6%.
An adaptive gate drive scheme is used to control the dead
time between two switches. The dead time control circuit
monitors the LGATE output and prevents the upper side
MOSFET from turning on until LGATE is fully off, preventing
cross-conduction and shoot-through. In order for the dead
time circuit to work properly, there must be a low resistance,
low inductance path from the LGATE driver to MOSFET
gate, and from the source of MOSFET to PGND. The
external Schottky diode is between the VDDP pin and BOOT
pin to keep the bootstrap capacitor charged.
The PWM controller is disabled when EN = GND, but the rest
of the circuitry, including the AC or DC adapter detecting
circuit and AC adapter current monitoring circuits, is still alive.
Setting the Battery Regulation Voltage
The ISL6253 uses a high-accuracy trimmed band-gap
voltage reference to regulate the battery charging voltage.
The VADJ input adjusts the charger output voltage, and the
VADJ control voltage can vary from 0 to VREF (2.39V),
providing a 10% adjustment range (from 4.2V -5% to
4.2V +5%) on CSON regulation voltage. An overall voltage
accuracy of better than 0.5% is achieved.
The per-cell battery termination voltage is a function of the
battery chemistry. Consult the battery manufacturers to
determine this voltage.
Float VADJ to set the battery voltage VCSON = 4.2V ×
number of the cells,
• Connect VADJ to VREF to set 4.41V × number of cells,
• Connect VADJ to ground to set 3.99V × number of cells.
Note that other battery charge voltages can be set by
connecting a resistor divider from VREF to ground. The
resistor divider should be sized to draw no more than 100µA
from VREF; or connect a low impedance voltage source like
the D/A converter in the micro-controller. The programmed
battery voltage per cell can be determined by the following
equation:
V CELL = 0.175V VADJ + 3.99V
(EQ. 1)
ISL6253
Connect CELLS as shown in Table 1 to charge 2, 3, or 4 Li+
cells. When charging other cell chemistries, use CELLS to
select an output voltage range for the charger. The internal
error amplifier gm1 maintains voltage regulation. The voltage
error amplifier is compensated at VCOMP. The component
values shown in Figure 18 provide suitable performance for
most applications. Individual compensation of the voltage
regulation and current-regulation loops allows for optimal
compensation.
the input current by reducing the charging current, when the
input current exceeds the input current−limit set point.
System current normally fluctuates as portions of the system
are powered up or down. Without input current regulation,
the source must be able to supply the maximum system
current and the maximum charger input current
simultaneously. By using the input current limiter, the current
capability of the AC adapter can be lowered, reducing
system cost.
TABLE 1. CELL NUMBER PROGRAMMING
The ISL6253 limits the battery charge current when the input
current-limit threshold is exceeded, ensuring the battery
charger does not load down the AC adapter voltage. An
internal amplifier gm3 compares the voltage between CSIP
and CSIN to the input current limit threshold voltage set by
ACLIM. Connect ACLIM to REF, Float and GND for the fullscale input current limit threshold voltage of 103mV, 78mV,
and 53mV, respectively, or use a resistor divider from VREF to
ground to set the input current limit as the following equation:
CELLS
CELL NUMBER
VCC
4
GND
3
Float
2
Setting the Battery Charge Current Limit
The CHLIM input sets the maximum charging current. The
current set by the current sense-resistor connects between
CSOP and CSON. There are three default battery charge
current-sense threshold voltages: 127mV for CHLIM = 3.3V,
100mV for CHLIM = VREF, 65mV for Float, and 30mV for
ground. The full-scale differential voltage between CSOP
and CSON is 100mV for CHLIM = VREF, so the maximum
charging current is 4.0A for a 25mΩ sensing resistor. Other
battery charge current-sense threshold values can be set by
connecting a resistor divider from VREF or VDD to ground,
or by connecting a low impedance voltage source like a D/A
converter in the microcontroller. The charge current limit
threshold is given by:
1 0.07
I CHG = -------  ----------------- V CHLIM + 0.03 

R 1  VREF
(EQ. 2)
If the battery voltage is less than 3.0V/cell and the battery
charging voltage is a percentage of the charging current in
constant current charge mode, the trickle charge current limit
threshold is given by:
1
I TR ,CHG = ------- ( 0.00157 × V CHLIM + 0.0037 

R1
(EQ. 3)
1 0.05
I INPUT = -------  ----------------- V ACLIM + 0.053

R 2  VREF
(EQ. 4)
When choosing the current sense resistor, note that the
voltage drop across this resistor causes further power
dissipation, reducing efficiency. The AC adapter current
sense accuracy is very important. Use a 1% tolerance
current-sense resistor. The highest accuracy of ± 3% is
achieved with 103mV current-sense threshold voltage for
ACLIM = VREF, but it has the highest power dissipation. For
example, it has 400mW power dissipation for rated 4A AC
adapter, and a 1W sensing resistor may have to be used. ±
4% and ± 6% accuracy can be achieved with 78mV and
53mV current-sense threshold voltage for ACLIM = Floating
and ACLIM = GND, respectively.
A low pass filter is recommended to eliminate the switching
noise. Connect the resistor to the CSIN pin instead of the
CSIP pin because the CSIN pin has lower bias current and
less influence on the current-sense accuracy.
AC Adapter Detection
When choosing the current sensing resistor, note that the
voltage drop across the sensing resistor causes further
power dissipation, reducing efficiency. However, adjusting
CHLIM voltage to reduce the voltage across the current
sense resistor R1 will degrade accuracy due to the smaller
signal to the input of the current sense amplifier. There is a
trade-off between accuracy and power dissipation. A low
pass filter is recommended to eliminate switching noise.
Connect the resistor to the CSOP pin instead of the CSON
pin, as the CSOP pin has lower bias current and less
influence on current-sense accuracy.
Connect the AC adapter voltage through a resistor divider to
ACSET to detect when AC power is available, as shown in
Figure 17. ACPRN is an open-drain output and is high
impedance when ACSET is less than Vth,rise and active low
impedance when ACSET is above Vth,fall . Vth,rise and
Vth,fall are given by:
Setting the Input Current Limit
Where Ihys is the ACSET input bias current hysteresis and
VACSET = 1.235V (min), 1.26V (typ.) and 1.285V (max.). The
hysteresis is Ihys R8, where Ihys = 2µA (min.), 3.4µA (typ.)
and 4.8µA (max.)
The total input current from an AC adapter, or other DC
source, is a function of the system supply current and the
battery-charging current. The input current regulator limits
14
R

V th, rise =  ------8- + 1 • V ACSET
 R9

(EQ. 5)
R

V th, fall =  ------8- + 1 • V ACSET – I hys R 8
 R9

ISL6253
DC Adapter Detection
Connect the DC adapter voltage like aircraft power through a
resistor divider to DCSET to detect when DC power is
available, as shown in Figure 18. DCPRN is an open-drain
output and is high impedance when DCSET is less than
Vth,rise, and active low impedance when DCSET is above
Vth,fall. Vth,rise and Vth,fall are given by:
 R 13

V th, rise =  --------- + 1 • V DCSET
 R 14

(EQ. 6)
 R 13

V th, fall =  --------- + 1 • V DCSET – I hys R 13
 R 14

Short Circuit Protection and 0V Battery Charging
Where Ihys is the DCSET input bias current hysteresis and
VACSET = 1.235V (min), 1.26V (typ.) and 1.285V (max.). The
hysteresis is IhysR13, where Ihys = 2µA (min.), 3.4µA (typ.)
and 4.8µA (max.)
Current Measurement
Use ICM to monitor the input current being sensed across
CSIP and CSIN. The output voltage range is 0 to 2.5V. The
voltage of ICM is proportional to the voltage drop across
CSIP and CSIN, and is given by the following equation:
ICM = 19.22 • I INPUT • R 2
(EQ. 7)
where IINPUT is the DC current drawn from the AC adapter.
ICM has ±5% accuracy. Connect a low pass filter to ICM to
bypass the switching frequency noise.
LDO Regulator
VDD provides a 5.0V supply voltage from the internal LDO
regulator from DCIN and can deliver up to 30mA of current.
This 30mA current is only used to supply the analog and
logic circuits for the IC and power the gate drivers. The
MOSFET drivers are powered by VDDP, and VDDP
connects to VDD through an external low pass filter. Bypass
VDDP and VDD with a 1µF capacitor.
Supply Isolation
If the voltage across the adapter sense resistor R2 is
typically greater than 8.5mV, the p-channel MOSFET
controlled by SGATE is turned on reducing the power
dissipation. If the voltage across the adapter sense resistor
R2 is less than 2mV, SGATE turns off the p-channel
MOSFET isolating the adapter from the system bus.
Battery Power Source Selection and Aircraft
Power Application
The battery voltage is monitored by CSON. If the battery
voltage measured on CSON is less than the adapter voltage
measured on DCIN, then the p-channel MOSFET controlled
by BGATE turns off. If it is greater, then BGATE turns on the
battery discharge p-channel MOSFET to minimize the power
loss. In the meantime, it also disables charging function and
turns off the AC adapter isolation p-channel MOSFET
controlled by SGATE. If designing for airplane power,
15
DCSET is tied to a resistor divider sensing the adapter
voltage. When a user is plugged into the 15V airplane supply
and their battery is lower than 15V, the MOSFET driven by
BGATE (see Figure 18) is turned off and keeps the battery
from supplying the system bus. The comparator looking at
CSON and DCIN has 300mV of hysteresis to avoid
chattering. For aircraft power applications the ISL6253 is
only able to support 2S and 3S battery packs when disabling
the charging function based on DCPRN and ACPRN signals.
For 4S battery packs, DCSET = 0 and the DCPRN signal is
not available to support aircraft power applications.
Since the battery charger will regulate the charge current to
be trickle charge current, as long as the battery voltage is
below 3.0V/cell, it automatically has short circuit protection
and is able to provide the trickle charge current and “wakeup” an extremely discharged battery.
Over Temperature Protection
If the die temp exceeds 150°C, it stops charging. Once the
die temp drops below 130°C, charging will start up again.
Application Information
The following battery charger design refers to the typical
application circuit in Figure 17, where a typical battery
configuration of 4S2P is used. This section describes how to
select the external components including the inductor, input
and output capacitors, switching MOSFETs, and current
sensing resistors.
Inductor Selection
The inductor selection has trade-offs between cost, size and
efficiency. For example, the lower the inductance, the
smaller the size, but ripple current is higher. This also results
in higher ac losses in the magnetic core and the windings,
which decrease the system efficiency. On the other hand,
the higher inductance results in lower ripple current and
smaller output filter capacitors, but it has higher DCR (dc
resistance of the inductor) loss, and has slower transient
response. So, the practical inductor design is based on the
inductor ripple current being ±(15-20)% of the maximum
operating dc current at maximum input voltage. The required
inductance can be calculated from:
V BAT
V IN ,MAX – V BAT
L = --------------------------------------------- ---------------------------∆I L
V IN ,MAX f s
(EQ. 8)
where VIN,MAX, VBAT, and fs are the maximum input
voltage, battery voltage and switching frequency,
respectively. The inductor ripple current ∆IL is found from:
∆I L = 30% ⋅ I BAT ,MAX
(EQ. 9)
where the maximum peak-to-peak ripple current is 30% of
the maximum charge current used.
ISL6253
For VIN,MAX = 20V, VBAT = 16.8V, IBAT,MAX = 4A, and
fs = 300kHz, the calculated inductance is 12.8µH. Choosing
the closest standard value gives L = 15µH. Ferrite cores are
often the best choice since they are optimized at 300kHz to
600kHz operation with low core loss. The core must be large
enough not to saturate at the peak inductor current IPeak:
1
I Peak = I BAT ,MAX + --- ∆I L
2
(EQ. 10)
Output Capacitor Selection
The output capacitor in parallel with the battery is used to
absorb the high frequency switching ripple current and
smooth the output voltage. The RMS value of the output
ripple current Irms is given by:
V IN ,MAX
I RMS = ----------------------- D ( 1 – D )
12 L f s
(EQ. 11)
where the duty cycle D is the ratio of the output voltage
(battery voltage) over the input voltage for continuous
conduction mode, which is typical operation for a battery
charger. During the battery charge period, the output voltage
varies from its initial battery voltage to the rated battery
voltage; therefore, the duty cycle change can be in the range
of between 0.375 and 0.63 for the minimum battery voltage
of 7.5V (2.5V/Cell) and the maximum battery voltage of
12.8V. The maximum RMS value of the output ripple current
occurs at the duty cycle of 0.5 and is expressed as:
V IN ,MAX
I RMS = ----------------------4 12 Lf s
(EQ. 12)
For VIN,MAX = 20V, L = 15µH, and fs = 300kHz, the
maximum RMS current is 0.32A. A typical 10µF or 22µF
ceramic capacitor is a good choice to absorb this current,
and also has very small size. The tantalum capacitor has a
known failure mechanism when subjected to high surge
current.
EMI considerations usually make it desirable to minimize
ripple current in the battery leads. Beads may be added in
series with the battery pack to increase the battery
impedance at 300kHz switching frequency. Switching ripple
current splits itself between the battery and the output
capacitor, depending on the ESR of the output capacitor and
the battery impedance. If the ESR of the output capacitor is
20mΩ and the battery impedance is raised to 2Ω with a
bead, then only 1% of the ripple current will flow in the
battery.
MOSFET Selection
The notebook battery charger synchronous buck converter
has input voltage from the AC adapter output. The maximum
AC adapter output voltage does not exceed 25V; therefore, a
30V logic MOSFET should be used.
The high side MOSFET must be able to dissipate the
conduction losses plus the switching losses. For the battery
16
charger application, the input voltage of the synchronous
buck converter is equal to the AC adapter output voltage,
which is relatively constant. The maximum efficiency is
achieved by selecting a high side MOSFET that has the
conduction losses equal to the switching losses. Ensure that
the ISL6253 LGATE gate driver can supply sufficient gate
current to prevent it from conduction, which is due to the
injected current into the drain-to-source parasitic capacitor
(Miller capacitor Cgd), and caused by the voltage rising rate
at the phase node at the moment of the high-side MOSFET
turning on; otherwise, cross-conduction problems may
occur. Reasonably slowing the turn-on speed of the highside MOSFET, by connecting a resistor between the BOOT
pin and gate drive supply source, and the high sink current
capability of the low-side MOSFET gate driver, helps reduce
the possibility of cross-conduction.
For the high-side MOSFET, the worst-case conduction
losses occur at the minimum input voltage:
V OUT 2
P Q1 ,Conduction = ---------------- I BAT R DSON
V IN
(EQ. 13)
The optimum efficiency occurs when the switching losses
equal the conduction losses. However, it is difficult to
calculate the switching losses in the high-side MOSFET
since it must allow for difficult-to-quantify factors that
influence the turn-on and turn-off times. These factors
include: the MOSFET internal gate resistance, gate charge,
threshold voltage, stray inductance, and pull-up and pulldown resistance of the gate driver. The following switching
loss calculation provides a rough estimate:
(EQ. 14)
Q gd
Q gd
1
1
PQ1 ,Switching = --- V IN I LV f s ------------------------+ --- V IN I LP f s ---------------- + Q rr VIN f s
I g ,source 2
I g , sin k
2
where Qgd is drain-to-gate charge; Qrr is total reverse
recovery charge of the body-diode in low side MOSFET; ILV
is inductor valley current; ILP is Inductor peak current; and
Ig,sink and Ig,source are the peak gate-drive source/sink
current of Q1, respectively.
Achievement of low switching losses requires low drain-togate charge, Qgd. Generally, the lower the drain-to-gate
charge, the higher the on-resistance; therefore, there is a
trade-off between the on-resistance and drain-to-gate
charge. Good MOSFET selection is based on the Figure of
Merit (FOM), which is the product of the total gate charge
and on-resistance. Usually, the smaller the value of FOM,
the higher the efficiency for the same application.
For the low-side MOSFET, the worst-case power dissipation
occurs at minimum battery voltage and maximum input
voltage:
V OUT 2

P Q2 =  1 – --------------- I
R
V IN  BAT DSON

(EQ. 15)
ISL6253
Choose a low-side MOSFET that has the lowest possible
on-resistance, has a moderate-sized package like SO-8 and
is reasonably priced. The switching losses are not an issue
for the low side MOSFET because it operates at zerovoltage-switching.
Choose a Schottky diode in parallel with low-side MOSFET
Q2 with a forward voltage drop low enough to prevent the
low-side MOSFET Q2 body-diode from turning on during the
dead time. This also reduces the power loss in the high-side
MOSFET associated with the reverse recovery of the lowside MOSFET Q2 body diode.
As a general rule, select a diode with a DC current rating
equal to one-third of the load current. One option is to
choose a combined MOSFET and Schottky diode in a single
package. The integrated packages may work better in
practice because there is less stray inductance due to a
short connection. This Schottky diode is optional and may be
removed if efficiency loss can be tolerated. In addition,
ensure that the required total gate drive current for the
selected MOSFETs is less than 26mA. The total gate charge
for the high-side and low-side MOSFETs is limited by the
following equation:
I GATE
Q GATE ≤ ----------------fs
(EQ. 16)
where IGATE is the total gate drive current and should be
less than 26mA. Substituting IGATE = 26mA and fs = 300kHz
into the above equation yields a total gate charge which
should be less than 86nC; therefore, the ISL6253 easily
drives the battery charge current up to 10A.
Input Capacitor Selection
The input capacitor absorbs the ripple current from the
synchronous buck converter, which is given by:
V OUT ( V IN – V OUT )
I rms = I BAT ---------------------------------------------------V IN
(EQ. 17)
This RMS ripple current must be smaller than the rated RMS
current in the capacitor data sheet. Non-tantalum
chemistries (ceramic, aluminum, or OSCON) are preferred
due to their resistance to power-up surge currents when the
AC adapter is plugged into the battery charger. For
Notebook battery charger applications, a ceramic capacitor
or a polymer capacitor from Sanyo is recommended due to
its small size and reasonable cost.
Table 2 shows the component lists for the typical application
circuit in Figure 18.
TABLE 2. COMPONENT LIST
PARTS
C1, C2
PART NUMBERS AND MANUFACTURER
10µF/25V ceramic capacitor,
TDK, C4532X7R1E106M
C3, C5, C9, C12 0.1µF/50V ceramic capacitor
C4, C8, C10
1µF/10V ceramic capacitor,
Taiyo Yuden LMK212BJ105MG
C6
10nF ceramic capacitor
C7
6.8nF ceramic capacitor
C11
10µF or 22µF/25V/10mΩ ceramic capacitor
TDK, C5750X7R1E226M
D1
30V/3A Schottky Diode, EC31QS03L, Nihon
(optional)
D2, D3
L
100mA/30V Schottky Diode, Central
Semiconductor
15µH/4.5A/20mΩ, Sumida, CDRH127-150
Q1
30V/14mΩ, IRF7811AV, International Rectifier
Q2
30V/30mΩ, FDS6612A, Fairchild
Q3
-30V/9.5mΩ, Si4413DY, Siliconix
R1
25mΩ, ±1%, LRC-LR2010-01-R025-F, IRC
R2
20mΩ, ±1%, LRC-LR2010-01-R020-F, IRC
R3
18Ω, ±5%, (0805)
R4
2.2Ω, ±5%, (0805)
R5
100kΩ, ±5%, (0805)
R6
10k, ±5%, (0805)
R7
4.7Ω, ±5%, (0805)
R8
130kΩ, ±1%, (0805)
R9
10.2kΩ, ±1%, (0805)
R10, R11, R12
10kΩ, ±5%, (0805)
R13
100Ω, ±5%, (0805)
R14
100kΩ, ±5%, (0805)
Loop Compensation Design
ISL6253 uses constant frequency current mode control
architecture to achieve fast loop transient response.
Accurate current sensing resistors in series with the output
inductor is used to regulate the charge current, and the
sensed current signal is injected into the voltage loop to
achieve current mode control to simplify the loop
compensation design. The inductor is not considered as a
state variable for current mode control, and the system
becomes a single order system. It is much easier to design a
type II compensator to stabilize the voltage loop than voltage
mode control.
Figure 19 shows the small signal model of the synchronous
buck regulator.
17
ISL6253
PWM Comparator Gain Fm
The PWM comparator gain Fm for peak current mode
control is given by:
1
dˆ
F m = ----------------- = ----------------V PWM
v̂ comp
(EQ. 18)
where VPWM is the peak-peak voltage of the PWM ramp
signal.
Current Sampling Transfer Function He(S)
In current loop, the current signal is sampled every switching
cycle. It has the following transfer function:
2
(EQ. 19)
S
S
H e ( S ) = ------- + --------------- + 1
2 ω Q
n
n
ωn
If Ti(S)>>1, then the above equation can be simplified as
follows:
S
1 + -----------V FB R o + R L
ω esr A v ( S )
1
L v ( S ) = ----------- --------------------- ---------------------- ----------------, ω p ≈ --------------S He ( S )
V o R TV
Ro Co
1 + ------ωp
Where RTV is the trans-resistance due to the current
information fed into the voltage loop. From the above
equation, it is shown that the system is a single order
system, which has a single pole located at ω p before the
half switching frequency. Therefore, a simple type II
compensator can be easily used to stabilize the system.
Figure 20 shows the type II compensator, and its transfer
function is expressed as follows:
2
where Qn and ωn are given by Q n = – ---, ω n = πf s ,
π
respectively.
^
iin
Power Stage Transfer Functions
v^in
+
Transfer function F1(S) from control to output voltage is:
S
1 + -----------ω esr
v̂ o
F 1 ( S ) = ------ = V in --------------------------------------2
dˆ
S
S
------- + --------------- + 1
2 ω Q
o p
ωo
^ 1:D
ILd
^
iL
^
Vo
L
^
Vind
Rc
+
RT
Ro
Co
(EQ. 20)
Ti(S)
^
d
K
Fm
C
1
1
Where ω esr = --------------, Q ≈ R o ------o-, ω o = --------------L
Rc Co p
LC o
Transfer function F2(S) from control to inductor current is,
S
1 + -----(EQ. 21)
ˆi
ωz
V in
1
o -------------------- --------------------------------------- where ω z ≈ --------------F 2 ( S ) = ---- =
Ro Co
dˆ R o + R L S 2
S
------- + --------------- + 1
2 ω Q
o p
ωo
Current loop gain Ti(S) is expressed as the following
equation:
T i ( S ) = R T F m F 2 ( S )H e ( S )
(EQ. 25)
He(S)
+
Tv(S)
V^comp
-Av(S)
FIGURE 19. SMALL SIGNAL MODEL OF SYNCHRONOUS
BUCK REGULATOR
Vo
(EQ. 22)
VFB
where RT is the trans-resistance in current loop. RT is
usually equal to the product of the current sensing resistance
of the current amplifier. For ISL6253, RT = 24R1.
The voltage gain with open current loop is:
VREF
+
gm
VCOMP
R1
C2
C1
(EQ. 23)
T v ( S ) = KF m F 1 ( S )A v ( S )
V
FB
where K = ----------- , VFB is the feedback voltage of the voltage
Vo
FIGURE 20. TYPE II COMPENSATOR
error amplifier.
The Voltage loop gain with current loop closed is given by:
Tv ( S )
L v ( S ) = ----------------------1 + Ti ( S )
(EQ. 24)
18
ISL6253
Figure 20 shows the type II compensator and its transfer
function is expressed as follows:
S
1 + ---------ω cz
gm
v̂ comp
A v ( S ) = ----------------- = --------------------- -----------------------------C1 + C2 
S
v̂ FB
S 1 + ----------

ω 
(EQ. 26)
cp
PCB Layout Considerations
Power and Signal Layer Placement on the PCB
As a general rule, power layers should be close together,
either on the top or bottom of the board, with signal layers on
the opposite side of the board. For example, layer
arrangement on a 4 layer board is shown below:
1
1
2
where ω cz = --------------- , ω cp = ----------------------
C +C
Layer 1: Small signal external components
R1 C1 C2
Layer 2: Signal Ground
Compensator design goal:
Layer 3: Power Ground
• High DC gain
Layer 4: Bottom Layer: Power MOSFET, Inductors and
other Power traces
R1 C1
1 1
• Loop bandwidth fc:  --- – ------ f s
5 30
Separate the power voltage and current flowing path from
the control and logic level signal path. The controller IC will
stay on the signal layer, which is isolated by the signal
ground to the power signal traces.
• Gain margin: >10dB
• Phase margin: 40°
The compensator design procedure is as follows:
1
1. Put compensator zero at ω cz = ( 1 – 3 ) -------------R C
Component Placement
o o
2. Put one compensator pole at zero frequency to achieve
high DC gain, and put another compensator pole at either
esr zero frequency or half switching frequency, whichever
is lower.
The loop gain Tv(S) at cross over frequency of fc has unity
gain. Therefore, the compensator resistance R1 is
determined by:
2πf c V o C o R T
R 1 = ----------------------------------g m V FB
(EQ. 27)
where gm is the trans-conductance of the voltage error
amplifier. Compensator capacitor C1 is then given by:
C1
1
C 1 = -----------------, C 2 = ----------------------------------------2πR 1 C 1 f esr – 1
R 1 ω cz
(EQ. 28)
Example: Vin = 20V, Vo = 16.8V, Io = 4A, fs = 300kHz,
Co = 22µF/10mΩ, L = 15µH, gm = 250µs, RT = 0.15Ω
(Rcs = 25mΩ, Ac = 6), VFB = 2.1V, VPWM = VIN/11,
fc = 15kHz, then compensator resistance R1 = 10kΩ.
Put the compensator zero at 1.7kHz, and put the
compensator pole at esr zero which is 725kHz. The
compensator capacitors are:
C1 = 10nF, C2 = 22pF
Such small C2 may not be necessary since it does not affect
the phase and gain at such high frequency.
19
The power MOSFET should be close to the IC so that the
gate drive signal, the LGATE, UGATE, PHASE, and BOOT
traces can be short.
Place the components in such a way that the area under the
IC has fewer noise traces with high dv/dt and di/dt, such as
gate signals and phase node signals.
SIGNAL GROUND AND POWER GROUND CONNECTION
At minimum, a reasonably large area of copper, which will
shield other noise couplings through the IC, should be used
as signal ground beneath the IC. The best tie-point between
the signal ground and the power ground is at the negative
side of the output capacitor on each side, where there is little
noise; a noisy trace beneath the IC is not recommended.
GND AND VDD PIN
At least one high quality ceramic decoupling cap should be
used to cross the GND and VDD pins. The decoupling cap
can be put close to the IC.
LGATE PIN
This is the gate drive signal for the bottom MOSFET of the
buck converter. The signal going through this trace has both
high dv/dt and high di/dt, and the peak charging and
discharging current is very high. These two traces should be
short, wide, and away from other traces. There should be no
other traces in parallel with these traces on any layer.
PGND PIN
The PGND pin should be laid out to the negative side of the
relevant output cap with separate traces. The negative side
of the output capacitor must be close to the source node of
the bottom MOSFET.
ISL6253
PHASE PIN
DCIN PIN
This trace should be short, and positioned away from other
weak signal traces. This node has a very high dv/dt with a
voltage swing from the input voltage to ground. No trace
should be in parallel with it. This trace is also the return path
for UGATE. Connect this pin to the high-side MOSFET
source.
This pin connects to AC adapter output voltage, and should
be less noise sensitive.
UGATE PIN
This pin has a square shape waveform with high dv/dt. It
provides the gate drive current to charge and discharges the
top MOSFET with high di/dt. This trace should be wide,
short, and away from other traces similar to the LGATE.
BOOT PIN
The BOOT pins di/dt is as high as the UGATE; therefore, this
trace should be as short as possible.
CSOP, CSON PINS
The current sense resistor connects to the CSON and the
CSOP pins through a low pass filter. The CSON pin is also
used as the battery voltage feedback. The traces should be
away from the high dv/dt and di/di pins like the PHASE and
BOOT pins. In general, the current sense resistor should be
close to the IC. Other layout arrangements should be
adjusted accordingly.
EN PIN
This pin stays high at enable mode and low at idle mode,
and is relatively robust. Enable signals should refer to the
signal ground.
20
Copper Size for the Phase Node
The capacitance of PHASE should be kept very low to
minimize ringing. It would be best to limit the size of the
PHASE node copper in strict accordance with the current
and thermal management of the application.
Identify the Power and Signal Ground
The input and output capacitors of the converters, and the
source terminal of the bottom switching MOSFET PGND
should connect to the power ground. The other components
should connect to signal ground. Signal and power ground
are tied together at one point.
Clamping Capacitor for Switching MOSFET
It is recommended that ceramic caps be used closely
connected to the drain of the high-side MOSFET and the
source of the low-side MOSFET. This capacitor reduces the
noise and the power loss of the MOSFET.
ISL6253
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
2X
L28.5x5
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VHHD-1 ISSUE C)
0.15 C A
MILLIMETERS
D
A
9
D/2
D1
D1/2
2X
N
6
INDEX
AREA
0.15 C B
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
9
0.30
5,8
A3
1
2
3
E1/2
b
E/2
E1
E
9
0.15 C B
2X
0.15 C A
5.00 BSC
-
4.75 BSC
9
E2
A2
A
0.08 C
9
4X P
-
4.75 BSC
2.95
3.10
9
3.25
7,8
-
0.50 BSC
-
k
0.25
-
-
L
0.50
0.60
0.75
8
L1
-
-
0.15
10
N
28
2
7
3
8
Ne
8
7
P
-
-
0.60
θ
-
-
12
7
NX k
D2
2 N
4X P
1
(DATUM A)
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
8
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
9
CORNER
OPTION 4X
(Nd-1)Xe
REF.
9
2. N is the number of terminals.
E2/2
N e
9
NOTES:
(Ne-1)Xe
REF.
E2
7
NX L
3
Rev. 0 02/03
2
3
6
INDEX
AREA
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
BOTTOM VIEW
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
A1
NX b
5
C
L
7,8
Nd
D2
8
3.25
0.10 M C A B
5
NX b
(DATUM B)
A1
A3
SIDE VIEW
3.10
5.00 BSC
e
/ / 0.10 C
C
SEATING PLANE
2.95
E1
0
4X
9
D
E
B
TOP VIEW
0.23
D1
D2
2X
0.20 REF
0.18
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
SECTION "C-C"
C
L
L1
10
L
L1
e
10
L
e
C C
TERMINAL TIP
FOR ODD TERMINAL/SIDE
FOR EVEN TERMINAL/SIDE
21
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
ISL6253
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
M28.15
N
INDEX
AREA
H
0.25(0.010) M
E
2
SYMBOL
3
0.25
0.010
SEATING PLANE
-A-
INCHES
GAUGE
PLANE
-B1
28 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
B M
A
D
L
h x 45°
-C-
α
e
A2
A1
B
C
0.10(0.004)
0.17(0.007) M
C A M
B S
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2
of Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
MIN
MAX
MILLIMETERS
MIN
MAX
NOTES
A
0.053
0.069
1.35
1.75
-
A1
0.004
0.010
0.10
0.25
-
A2
-
0.061
-
1.54
-
B
0.008
0.012
0.20
0.30
9
C
0.007
0.010
0.18
0.25
-
D
0.386
0.394
9.81
10.00
3
E
0.150
0.157
3.81
3.98
4
e
0.025 BSC
0.635 BSC
-
H
0.228
0.244
5.80
6.19
-
h
0.0099
0.0196
0.26
0.49
5
L
0.016
0.050
0.41
1.27
6
N
α
28
0°
28
8°
0°
7
8°
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
Rev. 1 6/04
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch)
per side.
5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of “B”
dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
22
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