DATASHEET

ISL6260C
®
Data Sheet
June 21, 2010
Multiphase PWM Regulator for IMVP-6+
Mobile CPUs
The ISL6260C is a multiple phase PWM buck regulator for
miroprocessor core power supply. The multiple phase
implementation results in better system performance, superior
thermal management, lower component cost, reduced power
dissipation, and smaller implementation area. The ISL6260C
multiphase controller together with ISL6208 external gate
drivers provide a complete solution to power Intel's mobile
microprocessors. The PWM modulator of ISL6260C is based
on Intersil's Robust Ripple Regulator technology (R3).
Compared with the traditional multiphase buck regulator, the R3
modulator commands variable switching frequency during load
transients, which achieves faster transient response. With the
same modulator, the switching frequency is reduced at light
load conditions resulting higher operation efficiency.
FN9259.3
Features
• Precision Multiphase Core Voltage Regulation
- 0.5% System Accuracy Over Temperature
- Enhanced Load Line Accuracy
• Microprocessor Voltage Identification Input
- 7-Bit VID Input
- 0.300V to 1.500V in 12.5mV Steps
- Supports VID Changes On-The-Fly
• Multiple Current Sensing Approaches Supported
- Lossless DCR Current Sensing
- Precision Resistive Current Sensing
• Supports PSI# and Narrow VDC for Enhanced Battery Life
(EBL) Initiatives
• Superior Noise Immunity and Transient Response
Intel Mobile Voltage Positioning (IMVP) reduces power
dissipation for Intel Pentium processors. The ISL6260C is
designed to be completely compliant with IMVP-6+
specifications. ISL6260C responds to PSI# signal by adding or
dropping PWM2 and adjusting overcurrent protection
accordingly. To reduce audible noise, the DPRSLPVR signal
can be used to reduce output voltage slew rates when entering
and exiting Deeper Sleep State according to Intel specification.
• Power Monitor and Thermal Monitor
The ISL6260C has several other key features. ISL6260C
reports output power through a power monitor pin. Current
sense can be achieved by using either inductor DCR or
discrete precision resistor. In the case of DCR current
sensing, a single NTC thermistor is used to thermally
compensate the inductor DCR variation with temperature. A
unity gain, differential amplifier is available for remote
voltage sensing. This allows the voltage on the CPU die to
be accurately regulated to meet Intel IMVP-6+
specifications.
• IMVP-6+ Compliant
• Differential Remote Voltage Sensing
• High Efficiency Across Entire Load Range
• Programmable 1, 2 or 3 Power Channels
• Excellent Dynamic Current Balance between Channels
• Small Footprint 40 Ld 6x6 QFN Package
• Pb-Free (RoHS Compliant)
Applications
• Mobile Laptop Computers
Ordering Information
PART NUMBER
(Note)
PART MARKING
TEMP. RANGE
(°C)
PACKAGE
(Pb-Free)
PKG.
DWG. #
ISL6260CCRZ
ISL6260 CCRZ
-10 to +100
40 Ld 6x6 QFN
L40.6x6
ISL6260CCRZ-T*
ISL6260 CCRZ
-10 to +100
40 Ld 6x6 QFN Tape and Reel
L40.6x6
ISL6260CIRZ
ISL6260 CIRZ
-40 to +100
40 Ld 6x6 QFN
L40.6x6
ISL6260CIRZ-T*
ISL6260 CIRZ
-40 to +100
40 Ld 6x6 QFN Tape and Reel
L40.6x6
*Please refer to TB347 for details on reel specifications.
NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin
plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are
MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2006, 2010. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6260C
Pinout
OCSET
ISL6260C
(40 LD QFN)
TOP VIEW
PGOOD
3V3
CLK_EN#
DPRSTP#
DPRSLPVR
VR_ON
VID6
VID5
VID4
VID3
Overcurrent set input. A resistor from this pin to VO sets
DROOP voltage limit for OC trip. A 10µA current source is
connected internally to this pin.
40
39
38
37
36
35
34
33
32
31
VW
A resistor from this pin to COMP programs the switching
frequency. (7kΩ gives approximately 300kHz). VW pin
sources current.
PSI# 1
30 VID2
PMON 2
29 VID1
RBIAS 3
28 VID0
VR_TT# 4
27 PWM1
NTC 5
26 PWM2
GND PAD
(BOTTOM)
SOFT 6
25 PWM3
COMP
This pin is the output of the error amplifier.
FB
This pin is the inverting input of error amplifier.
VDIFF
This pin is the output of the differential amplifier.
11
12
13
14
15
16
17
18
19
20
VDD
RTN
VSS
21 ISEN3
VIN
FB 10
VSUM
Remote core voltage sense input. Connect to
microprocessor die.
VO
22 ISEN2
DFB
COMP 9
DROOP
23 ISEN1
RTN
VW 8
VSEN
24 FCCM
VDIFF
OCSET 7
VSEN
Remote voltage sensing return. Connect to ground at
microprocessor die.
DROOP
Output of droop amplifier. Output = VO + DROOP.
Functional Pin Description
DFB
PSI#
Inverting input to droop amplifier.
Low load current indicator input. When asserted low,
indicates a reduced load-current condition. For ISL6260C,
when PSI# is asserted low, PWM2 will be disabled.
VO
An input to the IC that reports the local output voltage.
VSUM
PMON
An analog output. PMON sends out an analog signal
proportional to the product of VCCSENSE voltage and the
droop voltage.
This pin is connected to the current summation junction.
VIN
Battery supply voltage, used for feed forward.
RBIAS
VSS
147k Resistor to VSS sets internal current reference.
Signal ground; Connect to local controller ground.
VR_TT#
VDD
Thermal overload output indicator.
5V bias power.
NTC
ISEN3
Thermistor input to VR_TT# circuit.
Individual current sensing for channel 3.
SOFT
ISEN2
A capacitor from this pin to Vss sets the maximum slew rate
of the output voltage. It affects both soft start and VID
transitioning slew rate. Soft pin is the non-inverting input of
the error amplifier.
Individual current sensing for channel 2.
ISEN1
Individual current sensing for channel 1.
FCCM
Forced Continuous Conduction Mode (FCCM) enable pin to
MOSFET drivers. It will disable diode emulation.
2
FN9259.3
June 21, 2010
ISL6260C
PWM3
VR_ON
PWM output for channel 3. When PWM3 is pulled to 5V
VDD, PWM3 will be disabled and allow other channels to
operate.
Voltage Regulator enable input. A high level logic signal on
this pin enables the regulator.
PWM2
PWM output for channel 1.
Deeper Sleep Enable signal. At steady state, a high level
logic signal on this pin indicates that the micro-processor is
in Deeper Sleep Mode. Between active and sleep mode
transition, high logic level on this pin programs slow C4 entry
and exit; low logic level on this pin programs large charging
or discharging soft pin current, and therefore fast output
voltage transition slew rate.
VID0, VID1, VID2, VID3, VID4, VID5, VID6
DPRSTP#
VID input with VID0 = LSB and VID6 = MSB.
Deeper Sleep Enable signal. A low level logic signal on this
pin indicates that the micro-processor is in Deeper Sleep
Mode.
PWM output for channel 2. For ISL6260C, PSI# low will
make this output tri-state. When PWM2 is pulled to 5V VDD,
PWM2 will be disabled and allow other channels to operate.
PWM1
CLK_EN#
Digital output to enable System PLL Clock; Goes active after
13 switching cycles after Vcore is within 10% of Boot
Voltage.
DPRSLPVR
PGOOD
Power Good open-drain output. Will be pulled up externally
by a 680Ω resistor to VCCP or 1.9kΩ to 3.3V.
3V3
3.3V supply voltage for CLK_EN# logic, such an
implementation will improve power consumption from 3.3V
compared to open drain circuit other wise.
3
FN9259.3
June 21, 2010
ISL6260C
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to +7V
Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +25V
Open Drain Outputs, PGOOD, VR_TT# . . . . . . . . . . . . -0.3 to +7V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to (VDD + 0.3V)
Thermal Resistance (Notes 1, 2)
θJA (°C/W) θJC (°C/W)
QFN Package. . . . . . . . . . . . . . . . . . . .
30
5.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile. . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Operating Conditions
Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +100°C
Supply Voltage Range (Typical). . . . . . . . . . . . . . . . . . . . . +5V ±5%
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. θJA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Operating Conditions: VDD = 5V, TA = -40°C to +100°C, unless otherwise noted. Parameters with MIN and/or
MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by
characterization and are not production tested.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
3.6
INPUT POWER SUPPLY
+5V Supply Current
IVDD
4.2
mA
VR_ON = 0V
1
µA
I3V3
No load on CLK_EN#
1
µA
Battery Supply Current
IVIN
VR_ON = 0V
VIN Input Resistance
RVIN
VR_ON = 3.3V
Power-On-Reset Threshold
PORr
VDD rising
PORf
VDD falling
3.95
4.15
V
VDD falling, TA = -10°C to +100°C
4.00
4.15
V
No load; closed loop, active mode range
VID = 0.75V - 1.50V
-0.8
+0.8
%
No load; closed loop, active mode range
VID = 0.75V - 1.50V, TA = -10°C to +100°C
-0.5
+0.5
%
VID = 0.5V - 0.7375V
-10
+10
mV
VID = 0.5V - 0.7375V, TA = -10°C to +100°C
-8
+8
mV
VID = 0.3 - 0.4875V
-18
+18
mV
VID = 0.3 - 0.4875V, TA = -10°C to +100°C
-15
+15
mV
1.224
V
+3.3V Supply Current
VR_ON = 3.3V
1
900
4.35
µA
kΩ
4.5
V
SYSTEM AND REFERENCES
System Accuracy
%Error
(VCC_CORE)
VBOOT
1.176
1.200
Maximum Output Voltage
VCC_CORE(max) VID = [0000000]
1.500
V
Minimum Output Voltage
VCC_CORE(min) VID = [1100000]
0.300
V
0.0
V
VID Off State
VID = [1111111]
RBIAS Voltage
RBIAS = 147kΩ
1.45
1.47
Rfset = 7kΩ, 3 channel operation, VCOMP = 2V
285
300
See Equation 4 RFSET selection
200
1.49
V
CHANNEL FREQUENCY
Nominal Channel Frequency
fSW(nom)
Adjustment Range
315
kHz
500
kHz
+0.3
mV
AMPLIFIERS
Droop Amplifier Offset
-0.3
Error Amp DC Gain
Av0
Error Amp Gain-Bandwidth Product
4
GBW
(Note 3)
90
dB
CL= 20pF (Note 3)
18
MHz
FN9259.3
June 21, 2010
ISL6260C
Electrical Specifications
Operating Conditions: VDD = 5V, TA = -40°C to +100°C, unless otherwise noted. Parameters with MIN and/or
MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by
characterization and are not production tested. (Continued)
PARAMETER
SYMBOL
FB Input Current
TEST CONDITIONS
MIN
IIN(FB)
TYP
MAX
UNITS
10
150
nA
2
mV
ISEN
Imbalance Voltage
Maximum of ISENs - Minimum of ISENs
Input Bias Current
20
nA
SOFT CURRENT
Soft-start Current
ISS
SOFT Geyserville Current
IGV
|SOFT-VDAC| >100mV
-47
-42
-37
µA
±180
±205
±230
µA
IC4
DPRSLPVR = 3.3V
-47
-42
-37
µA
SOFT Deeper Sleep Exit Current
IC4EA
DPRSLPVR = 3.3V
37
42
47
µA
SOFT Deeper Sleep Exit Current
IC4EB
DPRSLPVR = 0V
180
205
230
µA
0.26
0.4
V
SOFT Deeper Sleep Entry Current
POWER GOOD AND PROTECTION MONITORS
PGOOD Low Voltage
VOL
IPGOOD= 4mA
PGOOD Leakage Current
IOH
PGOOD = 3.3V
-1
1
µA
PGOOD Delay
tpgd
CLK_ENABLE# LOW to PGOOD HIGH
6.3
7.6
8.9
ms
Overvoltage Threshold
OVH
VO rising above setpoint for >1ms
160
200
240
mV
Severe Overvoltage Threshold
OVHS
1.675
1.7
1.725
V
OCSET Reference Current
I(RBIAS) = 10µA
9.8
10
10.2
µA
OC Threshold Offset
DROOP rising above OCSET for >150µs
-2
4
mV
-235
mV
1.0
V
Current Imbalance Threshold
VO rising for >2µs
One ISEN above another ISEN for >1.2ms
Undervoltage Threshold
(VDIFF/SOFT)
UVf
VO falling below setpoint for >1.2ms
9
-355
-295
mV
LOGIC THRESHOLDS
VR_ON and DPRSLPVR Input Low
VIL(3.3V)
VR_ON and DPRSLPVR Input High
VIH(3.3V)
VID0-VID6, PSI#, DPRSTP# Input
Low
VIL(1.0V)
VID0-VID6, PSI#, DPRSTP# Input
High
VIH(1.0V)
2.3
V
0.3
0.7
V
V
PWM
PWM (PWM1-PWM3) Output Low
VOL(5.0V)
Sinking 5mA
FCCM Output Low
VOL_FCCM
Sinking 3mA
PWM (PWM1-PWM3) and FCCM
Output High
VOH(5.0V)
Sourcing 5mA
3.5
PWM = 2.5V
-1
PWM Tri-State Leakage
1.0
V
1.0
V
V
1
µA
THERMAL MONITOR
NTC Source Current
NTC = 1.3V
Over-Temperature Threshold
V (NTC) falling
VR_TT# Low Output Resistance
RTT
I = 20mA
CLK_EN# High Output Voltage
VOH
3V3 = 3.3V, I = -4mA
CLK_EN# Low Output Voltage
VOL
I = 4mA
53
60
67
µA
1.18
1.2
1.22
V
6.5
9
Ω
CLK_EN# OUTPUT LEVELS
2.9
3.1
V
0.26
0.4
V
POWER MONITOR
PMON Output Voltage
Vpmon
5
VSEN = 1.2V, Droop-Vo = 80mV
1.638
1.68
1.722
V
VSEN = 1.0V, Droop-Vo = 20mV
0.308
0.35
0.392
V
FN9259.3
June 21, 2010
ISL6260C
Electrical Specifications
Operating Conditions: VDD = 5V, TA = -40°C to +100°C, unless otherwise noted. Parameters with MIN and/or
MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by
characterization and are not production tested. (Continued)
PARAMETER
SYMBOL
PMON Maximum Voltage
TEST CONDITIONS
Vpmonmax
PMON Sourcing Current
VSEN = 1.0V, Droop-Vo = 50mV
PMON Sinking Current
VSEN = 1.0V, Droop-Vo = 50mV
Maximum Current Sinking Capability
See Figure 36
PMON Impedance
When PMON is within its sourcing/sinking
current range (Note 3)
MIN
TYP
2.8
3
MAX
V
2.0
mA
2.0
Vpmon/
250
UNITS
mA
Vpmon/
180
7
Vpmon/
130
A
Ω
3. Limits established by characterization and are not production tested.
6
FN9259.3
June 21, 2010
ISL6260C
Typical Operating Performance 3 Phase, DCR Sense, (1) 7821, (2) 7832 per phase, 300kHz, 0.5µH
100
1.46
VIN = 8.0V
1.44
VIN = 19.0V
1.42
80
VIN = 12.6V
VOUT (V)
EFFICIENCY (%)
90
VIN = 19.0V
70
1.40
VIN = 8.0V
VIN = 12.6V
1.38
1.36
60
1.34
50
1
1.32
0
100
10
10
20
IOUT (A)
100
1.43
90
1.42
80
VOUT (V)
EFFICIENCY (%)
50
1.44
VIN = 8.0V
VIN = 19.0V
VIN = 12.6V
40
FIGURE 2. ACTIVE MODE LOAD LINE, 3 PHASE, CCM,
PSI# = HIGH VID = 1.435V
FIGURE 1. ACTIVE MODE EFFICIENCY, 3 PHASE, CCM,
PSI# = HIGH, VID = 1.4375V
70
30
IOUT (A)
VIN = 12.6V
1.41
1.40
VIN = 8.0V
VIN = 19.0V
1.39
1.38
60
1.37
50
0.1
1.0
IOUT (A)
FIGURE 3. DEEPER SLEEP MODE EFFICIENCY, 3 PHASE,
DCM OPERATION, PSI# = LOW, VID = 1.4375V
100
VIN = 8.0V
1.36
10
0.76
VIN = 12.6V
70
0.73
0.72
0.71
VIN = 19.0V
0.69
10
FIGURE 5. DEEPER SLEEP MODE EFFICIENCY, 3 PHASE,
DCM OPERATION, PSI# = LOW, VID = 0.75V
7
30
0.70
VIN = 19.0V
1.0
IOUT (A)
20
VIN = 8.0V
0.74
80
50
0.1
IOUT (A)
VIN = 12.6V
0.75
VOUT (V)
EFFICIENCY (%)
10
FIGURE 4. DEEPER SLEEP MODE LOAD LINE, 3 PHASE,
CCM, PSI# = LOW VID = 1.435V
90
60
0
0.68
0
20
10
30
IOUT (A)
FIGURE 6. DEEPER SLEEP MODE LOAD LINE, 3 PHASE,
DCM OPERATION, PSI# = LOW, VID = 0.75V
FN9259.3
June 21, 2010
ISL6260C
Typical Operating Performance 3 Phase, DCR Sense, (1) 7821, (2) 7832 per phase, 300kHz, 0.5µH (Continued)
100
100
VIN = 8.0V
90
EFFICIENCY (%)
EFFICIENCY (%)
90
80
VIN = 19.0V
VIN = 12.6V
70
60
50
0
10
IOUT (A)
70
VIN = 19.0V
50
0.1
100
1.0
IOUT (A)
10
FIGURE 8. DEEPER SLEEP MODE EFFICIENCY, 2 PHASE,
DCM OPERATION, PSI# = LOW, VID = 1.4375V
1.44
100
90
VIN = 12.6V
VIN = 8.0V
VIN = 12.6V
1.42
1.40
80
70
VOUT (V)
EFFICIENCY (%)
VIN = 12.6V
60
FIGURE 7. ACTIVE MODE EFFICIENCY, 2 PHASE, CCM,
PSI# = HIGH, VID = 1.4375V
VIN = 19.0V
1.38
1.34
50
0.1
1.0
IOUT (A)
1.32
10
VIN = 19.0V
0
20
10
40
30
50
IOUT (A)
FIGURE 9. DEEPER SLEEP MODE EFFICIENCY, 2 PHASE,
DCM OPERATION, PSI# = LOW, VID = 0.75V
FIGURE 10. ACTIVE MODE LOAD LINE, 2 PHASE, CCM,
PSI# = HIGH, VID = 1.435V
0.76
1.44
VIN = 12.6V
1.43
0.75
VIN = 8.0V
1.42
VOUT (V)
1.40
1.39
VIN = 19.0V
1.38
VIN = 12.6V
0.74
1.41
VIN = 8.0V
0.73
0.72
0.71
0.70
VIN = 19.0V
0.69
1.37
1.36
VIN = 8.0V
1.36
60
VOUT (V)
VIN = 8.0V
80
0.68
0
10
20
30
IOUT (A)
FIGURE 11. DEEPER SLEEP MODE LOAD LINE, 2 PHASE,
DCM OPERATION, PSI# = LOW, VID = 1.4375V
8
0
10
20
30
IOUT (A)
FIGURE 12. DEEPER SLEEP MODE LOAD LINE, 2 PHASE, DCM
OPERATION, PSI# = LOW, VID = 0.75V
FN9259.3
June 21, 2010
ISL6260C
Typical Operating Performance
VSOFT (Green)
VOUT
VOUT (Brown)
PGOOD
CLK_EN#
VR_ON
VR_ON
FIGURE 13. SOFT-START WAVEFORM 0V TO 1.2V (BOOT
VOLTAGE) AND CLK_EN# TIMING
VIN
FIGURE 14. SOFT-START WAVEFORM SHOWING PGOOD
VOUT
VOUT
FIGURE 15. 12V-18V INPUT LINE TRANSIENT RESPONSE
FIGURE 16. SOFT-START INRUSH CURRENT, VIN = 8V
FIGURE 17. 3 PHASE CURRENT BALANCE, FULL LOAD = 50A
FIGURE 18. 2 PHASE CURRENT BALANCE, FULL LOAD = 50A
9
FN9259.3
June 21, 2010
ISL6260C
Typical Operating Performance (Continued)
VOUT
COMP PIN
FIGURE 19. TRANSIENT LOAD RESPONSE, 40A LOAD STEP
@ 200A/µs, 3 PHASE
FIGURE 20. TRANSIENT LOAD 3 PHASE OPERATION CURRENT BALANCE
FIGURE 21. TRANSIENT LOAD 3 PHASE OPERATION, ZOOM
OF RISING EDGE CURRENT BALANCE
FIGURE 22. TRANSIENT LOAD 3 PHASE OPERATION, ZOOM
OF FALLING EDGE CURRENT BALANCE
VID MSB
VID MSB
VOUT
FIGURE 23. IVID MSB BIT CHANGE FROM 1.4375V TO 0.65V
SHOWING 9mV/µs SLEW RATE, DPRSLPVR = 0,
DPRSTP# = 1
10
VOUT
FIGURE 24. SLEW RATE ENTERING C4, VID MSB BIT
CHANGE FROM 1.4375V TO 0.65V SHOWING
2mV/µs SLEW RATE, DPRSLPVR = 1, DPRSTP# = 0
FN9259.3
June 21, 2010
ISL6260C
Typical Operating Performance (Continued)
VOUT
VOUT @ 1.7V
PWM
DPRSTP# AND PSI#
VOUT @ 0.85V
DPRSLPVR AND MSB
FIGURE 25. C4 ENTRY AND EXIT SLEW RATES WITH
DPRSLPVR AND DPRSTP#
FIGURE 26. 1.7V OVP SHOWING OUTPUT PULLED LOW TO
0.85V AND PWM TRI_STATE
PWM
PWM
VOUT
IPHASE
PGOOD
VOUT
FIGURE 27. UNDERVOLTAGE RESPONSE SHOWING PWM
TRI-STATE, VOUT < VID - 300mV
PGOOD
FIGURE 28. OCP - RESPONSE
PWM
PSI#
CLK_EN#
IPHASE
VOUT
VOUT
PGOOD
FIGURE 29. WOCP - SHORT CIRCUIT PROTECTION
11
PHASE 2
FIGURE 30. ISL6260C, PHASE ADDING AND DROPPING IN
ACTIVE MODE, LOAD CURRENT = 15A
FN9259.3
June 21, 2010
ISL6260C
Typical Operating Performance (Continued)
PHASE 3 CURRENT
PSI#
CLK_EN#
PHASE 1 CURRENT
VOUT
PHASE 2
CURRENT
PHASE 2
PHASE 2
FIGURE 31. ISL6260C PHASE ADDING AND DROPPING IN
DEEPER SLEEP MODE, LOAD CURRENT = 4.35A
FIGURE 32. ISL6260C, INDUCTOR CURRENT WAVEFORM
WITH PHASE ADDING AND DROPPING IN DCM
OR DEEPER SLEEP MODE
PHASE 3
CURRENT
PHASE 1 CURRENT
PHASE 2 CURRENT
PHASE 1 CURRENT
PGOOD
PHASE 2 CURRENT
FIGURE 33. ISL6260C, INDUCTOR CURRENT WAVEFORM
WITH PHASE ADDING AND DROPPING IN CCM
OR ACTIVE MODE
FIGURE 34. ISL6260C, OVERCURRENT DUE TO PHASE
DROPPING
1.8
PMON (V)
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0.0
0.0
0.8
19V, 1.15V, 40A
VID = 1.15V, IOUT = 15A
0.7
7Ω
0.6
PMON (V)
1.6
19V, 1.15V, 30A
19V, 1.15V, 20A
0.5
VID = 1.15V, IOUT = 10A
0.4
180Ω
0.3
VID = 1.15V, IOUT = 5A
0.2
19V, 1.15V, 10A
0.1
19V, 1.15V, 5A
1.0
2.0
3.0
4.0
5.0
6.0
CURRENT SOURCING (mA)
7.0
FIGURE 35. POWER MONITOR CURRENT SOURCING
CAPABILITY
12
0.00
0.0
VID = 1.15V, IOUT = 2.5A
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
CURRENT SINKING (mA)
FIGURE 36. POWER MONITOR CURRENT SINKING
CAPABILITY
FN9259.3
June 21, 2010
ISL6260C
Typical Operating Performance (Continued)
25%
60.0
VIN = 19V
POWER (W)
15%
10%
5%
0%
0.0
50.0
VID = 1.15V
20%
40.0
PMON
30.0
20.0
MEASURED OUTPUT POWER
10.0
10.0
20.0
30.0
40.0
50.0
OUTPUT CURRENT (A)
FIGURE 37. POWER MONITOR ACCURACY
13
0.0
0.0
10.0
20.0
30.0
40.0
50.0
CURRENT (A)
FIGURE 38. POWER MONITOR vs OUTPUT CURRENT
FN9259.3
June 21, 2010
ISL6260C
Simplified Application Circuit for DCR Current Sensing
the regulator to operate in Diode Emulation for improved
light load efficiency. As shown in the circuit diagram, the
FCCM pin is connected to ISL6260C, which programs the
CCM or DCM mode.
Figure 39 shows a simplified application circuit for the
ISL6260C converter with inductor DCR current sensing. The
ISL6208 MOSFET gate driver has a force-continuousconduction-mode (FCCM) input, that when disabled, allows
V+5 VIN V+3.3
VIN
VDD
VIN
3V3
V+5
RBIAS
VCC
NTC
VR_TT#
PWM1
PWM
ISEN1
VR_TT#
BOOT
LO
UGATE
ISL6208
PHASE
SOFT
7
VID<0:6>
VIDs
DPRSTP#
RL CL
FCCM
LGATE
GND
ISEN1
DPRSTP#
ISL6260C
DPRSLPVR
DPRSLPVR
PSI#
VIN
PSI#
CLK_ENABLE#
VCC
PWM
PWM2
CLK_EN#
ISEN2
VR_ON
VR_ON
IMVP-6+_PWRGD
VO
V+5
PMON
PWR MONITOR
CO
BOOT
LO
UGATE
ISL6208
PHASE
PGOOD
RL CL
FCCM
LGATE
GND
VSEN
REMOTE SENSE
AT CPU CORE
VO'
VSUM
ISEN2
RTN
Ri
VSUM
FCCM
C3
R3
V+5
VCC
FB
C1
VO'
VIN
VDIFF
BOOT
R1
PWM
PWM3
ISEN3
COMP
C2
LO
UGATE
ISL6208
RFSET
VSUM
VSUM
VW
PHASE
FCCM
LGATE
GND
OCSET
GND
DFB
DROOP
VO
RL CL
ISEN3
VSUM
RN
VO'
CCS
VO'
FIGURE 39. TYPICAL APPLICATION CIRCUIT FOR DCR SENSING
14
FN9259.3
June 21, 2010
ISL6260C
Simplified Application Circuit for Resistive Current Sensing
stability margin of the channel current balance loop. No NTC
thermistor is needed and the droop circuit is simplified.
Figure 40 shows a simplified application circuit for the
ISL6260C converter with external resistor current sensing. A
capacitor is added in parallel with RL in order to improve the
V+5 V+3.3
VIN
VIN
VDD
3V3
VIN
V+5
RBIAS
VCC
NTC
PWM1
ISEN1
VR_TT#
VR_TT#
PWM
BOOT
LO
UGATE
RSEN
ISL6208
SOFT
7
VID<0:6>
PHASE
VIDs
VSUM
FCCM
LGATE
GND
DPRSTP#
CL
VO'
ISL6260C
DPRSLPVR
VIN
PSI#
PWR MONITOR
VCC
PWM
PWM2
CLK_EN#
ISEN2
VR_ON
VR_ON
IMVP-6+_PWRGD
VO
V+5
PMON
CLK_ENABLE#
CO
BOOT
LO
UGATE
RSEN
ISL6208
PHASE
PGOOD
VSUM
FCCM
LGATE
GND
VSEN
REMOTE SENSE
AT CPU CORE
ISEN2
RTN
FCCM
C1
R1
CL
VIN
VDIFF
R3
RL
VO'
Ri
C3
RL
ISEN1
V+5
VCC
FB
BOOT
PWM3
PWM
ISEN3
COMP
C2
UGATE
LO
RSEN
ISL6208
RFSET
PHASE
VW
VSUM
VSUM
FCCM
LGATE
GND
OCSET
GND
DFB DROOP
VSUM
ISEN3
RL
CL
VO
VO'
VO'
FIGURE 40. TYPICAL APPLICATION CIRCUIT FOR DISCRETE RESISTOR CURRENT SENSING
15
FN9259.3
June 21, 2010
Functional Block Diagram
RBIAS
PMON
3V3 CLK_EN# VIN
PGOOD
VO
ISEN1 ISEN2 ISEN3
VDD
VIN
VID0
POWER
VID1
PROTECTION
VID2
54µA
6µA
CURRENT
BALANCE
LOGIC
MONITOR
16
DAC
VID3
Dacout
VID4
IBAL
-
VID6
MODE
CONTROL
FCCM
+
NTC
FLT
OC
VDIFF
VID5
FAST_OC OR
WAY-OC
1.20V
OC
MULTIPLIER
VO
VIN
VO
1.24V
VR_TT#
+
2X
VR_ON
FLT
MODULATOR
MODE
SOFT
CONTROL
PWM1
DPRSLPVR
VO VSEN
OC
DPRSTP#
VIN
VO
NUMBER OF
PHASES
10µA
GAIN SELECT)
OCSET
FLT
MODULATOR
-
PWM2
OC
+
VSUM
DFB
OC
+
DROOP
-
DROOP
+
1
-
FLT
PWM3
E/A
+
NUMBER OF PHASES
+
1
-
+
CHANNEL
CLOCK
MODE
CONTROL
VIN
VO VSEN RTN
VO
MODULATOR
+
VO
VSEN
VIN
VDIFF
SOFT FB
COMP
VO
VW
FN9259.3
June 21, 2010
FIGURE 41. SIMPLIFIED BLOCK DIAGRAM
GND
SELECT
ISL6260C
PSI#
IBAL
CLK_EN#
GOOD
ISL6260C
Theory of Operation
VDD
Operational Description
The ISL6260C is a multiphase regulators implementing
Intel® IMVP-6+ protocol. It can be programmed for one-,
two- or three-channel operation for microprocessor core
applications up to 70A. With ISL6208 gate driver capable of
diode emulation, the ISL6260C provides optimum efficiency
in both heavy and light conditions.
ISL6260C uses Intersil patented R3 (Robust Ripple
Regulator™) modulator. The R3 modulator combines the
best features of fixed frequency PWM and hysteretic PWM
while eliminating many of their shortcomings. The ISL6260C
modulator internally synthesizes analog signals inside the IC
emulating the inductor ripple currents and use hysteretic
comparators on those signals to determine switching pulse
widths. Operating on these large-amplitude, noise-free
synthesized signals allows the ISL6260C to achieve lower
output ripple and lower phase jitter than conventional
hysteretic and fixed PWM mode controllers. Unlike
conventional hysteretic converters, the ISL6260C has an
error amplifier that allows the controller to maintain a 0.5%
output voltage accuracy. At heavy load conditions, the
ISL6260 is switching at a relatively constant switching
frequency similar to fixed frequency PWM controller. At light
load conditions, the ISL6260C is switching at a frequency
proportional to load current similar to hysteretic mode
controller.
ISL6260C disables PWM2 when PSI# is asserted low. And
the power monitor pin provides an analog signal
representing the output power of the converter.
10mV/µs
2mV/µs
VR_ON
120µs
VBOOT
90%
VID COMMANDED
VOLTAGE
SOFT & VO
13 SWITCHING CYCLES
CLK_EN#
~7ms
IMVP-6+ PGOOD
FIGURE 42. SOFT-START WAVEFORMS USING A 20nF SOFT
CAPACITOR
Static Operation
A) Voltage Regulation at Zero Load Current
After the start sequence, the output voltage will be regulated
to the value set by the VID inputs per Table 1. The entire VID
Table is presented in the Intel IMVP-6+™ specification. The
ISL6260C will control the no-load output voltage to an
accuracy of ±0.5% over the range of 0.75V to 1.5V.
TABLE 1. TRUNCATED VID TABLE FOR INTEL IMVP-6+™
SPECIFICATION
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VOUT
0
0
0
0
0
0
0
1.500V
0
0
0
0
0
0
1
1.4875
Start-up Timing
0
0
0
0
1
0
1
1.4375
With the controller's +5V VDD voltage above the POR
threshold, the start-up sequence begins when VR_ON
exceeds the 3.3V logic HIGH threshold. Approximately
120µs later SOFT and VOUT start ramping up to the boot
voltage of 1.2V. During this interval, the SOFT capacitor is
charged with approximately 40µA. Therefore, if the SOFT
capacitor is selected to be 20nF, the SOFT ramp will be at
about 2mV/µs for a soft-start time of 600µs. Once VOUT
(VDIFF) is within 10% of the boot voltage for 13 PWM cycles
(43µs for frequency = 300kHz), then CLK_EN# is pulled
LOW and the SOFT capacitor is charged up with
approximately 200µA. Therefore, VOUT slews at +10mV/µs
to the voltage set by the VID pins. Approximately 7ms later,
PGOOD is asserted HIGH. A typical start-up timing is shown
in Figure 42. Similar results occur if VR_ON is tied to VDD,
with the soft-start sequence starting 120µs after VDD
crosses the POR threshold.
0
0
0
0
1
1
1
1.4125
0
0
0
1
0
0
0
1.4000
0
0
1
0
0
0
1
1.2875
0
0
1
1
0
0
0
1.2000
0
0
1
1
1
0
0
1.1500
0
1
0
1
0
0
0
1.0000
0
1
0
1
0
1
1
0.9625
0
1
1
1
1
0
0
0.7500
1
0
0
0
1
0
0
0.6500
1
0
1
0
0
0
0
0.5000
1
1
0
0
0
0
0
0.300
1
1
0
0
0
0
1
Off
1
1
0
0
0
1
0
Off
17
...
Off
1
1
1
1
1
1
0
Off
1
1
1
1
1
1
1
Off
FN9259.3
June 21, 2010
ISL6260C
A differential amplifier allows voltage sensing for precise
voltage regulation at the microprocessor die. The inputs to
the amplifier are the VSEN and RTN pins.
B) Load Line or Droop Accomplishment
As the load current increases from zero, the output voltage
will drop from the VID table value by an amount proportional
to load current to achieve the IMVP-6+ load line. The
ISL6260C provides for current to be sensed using resistors
in series with the channel inductors as shown in the
application circuit of Figure 40 or using the intrinsic series
resistance of the inductors as shown in the application circuit
of Figure 39. In both cases, signals representing the inductor
currents are summed at VSUM which is the non-inverting
input to the DROOP amplifier shown in the block diagram of
Figure 41. The voltage at the DROOP pin minus the output
voltage at VO pin is the total load current multiplied by a gain
factor. This value is used as an input to the differential
amplifier to achieve the IMVP-6+ load line as well as the
input to the overcurrent circuit.
When using inductor DCR current sensing, a single NTC
element is used to compensate the positive temperature
coefficient of the copper winding thus sustaining the load-line
accuracy with reduced cost.
C) Phase Current Balance
In addition to the total current which is used for DROOP and
OCP, the individual channel average currents are also
monitored by the phase node voltage. Channel current
differences are sensed by comparing ISEN1, ISEN2, and
ISEN3 voltage. The IBAL circuit will adjust the channel
pulse-widths up or down relative to the other channels to
cause the voltages presented to the ISEN pins to be equal.
D) Enable and Disable Phases
The ISL6260C controller can be configured for three-, twoor single-channel operation. To disable channel two and/or
channel three, its PWM output pin should be tied to +5V and
the ISEN pins should be grounded. In three-channel
operation, the three channel PWM's are phase shifted by
120°, and in two-channel operation they are phase shifted by
180°.
E) Switching Frequency in CCM/DCM mode
The switching frequency is adjusted by the resistor between
the error amplifier output and the VW pin. When ISL6260C is
in continuous conduction mode (CCM), the switching
frequency may not be as constant as that of a fixed
frequency PWM controllers. However, the switching
frequency variation will be kept small to maintain the output
voltage ripple within SPEC. In general, the switching
frequency will be very close to the set value at high input
voltage and heavy load conditions.
When DPRSLPVR is high and DPRSTP# is low, the FCCM
pin will become low, and discontinuous conduction mode
18
(DCM) operation will be allowed in the ISL6208 gate drive. In
DCM, ISL6208 turns off the lower FET after its channel
current across zero. As load is further reduced, channel
switching frequency will drop, providing optimized efficiency
at light loading. FCCM logic low is the signal to enable, or to
allow the DCM operation. Only if the inductor current is really
cross zero, does the true DCM occur.
V o ut
10 m V /us
-2 m V /us
2 m V /u s
D P R S TP #
D P R S LP V R
M S B of V ID
FIGURE 43. DEEPER SLEEP TRANSITION SHOWING
DPRSLPVR’s EFFECT ON EXIT SLEW RATE
Dynamic Operation
Refer to Figure 43. The ISL6260C responds to changes in
VID command voltage by slewing to new voltages with a
dV/dt set by the SOFT capacitor and by the state of
DPRSLPVR. With CSOFT = 20nF and DPRSLPVR HIGH,
the output voltage will move at ±2mV/µs for large changes in
voltage. For DPRSLPVR LOW, the large signal dV/dt will be
±10mV/µs. As the output approaches the VID command
voltage, the dV/dt rate moderates to prevent overshoot.
During Geyserville III transitions where there is one LSB VID
step each 5µs, the controller will follow the VID command
with its dV/dt rate of ±2.5mV/µs.
Keeping DPRSLPVR HIGH during VID transitions will result
in reduced dV/dt slew rate and lesser audio noise. For
fastest recovery from Deeper Sleep to Active mode,
DPRSLPVR LOW achieves higher dV/dt as required by
IMVP-6+ DPRSTP# and DPRSLPVR logic SPEC.
Intersil's R3 intrinsically has voltage-feed-forward. The
output voltage is insensitive to a fast slew input voltage
change. Refer to Figure 15 in the “Typical Operating
Performance” on page 9 for Input Transient Performance.
The hysteresis window voltage is constructed with a resistor on
the Vw pin to the error amplifier outputs. The synthesized
inductor current ripple signal compares with the window voltage
and generates PWM signal. At load current step up, the
switching frequency is increased resulting in a faster response
than conventional fixed frequency PWM controllers. As all the
phases shares the same hysteretic window voltage, it also
ensures excellent dynamic current balance between phases.
The individual average phase voltages are monitored and
controlled to achieve steady state current balance among the
phases with current balance loop.
FN9259.3
June 21, 2010
ISL6260C
Modes of Operation Programmed by Logic Signals
The operational modes of ISL6260C are programmed by the
control signals of DPRSLPVR, DPRSTP#, and PSI#.
ISL6260C responds PSI# signal by adding or dropping
PWM2 and adjusting the overcurrent protection level
accordingly. For example, if the ISL6260C is initially used as
three phase controller, the PSI# signal will add or drop
PWM2 and leave PWM1 and PWM3 always in operation.
Meanwhile, after PWM2 is dropped, the phase shift between
the PWM1 and PWM3 is adjusted from 120° to 180° and the
overcurrent and the way-overcurrent protection level will be
adjusted to 2/3 of the initial value. If the ISL6260C is initially
used as two phase operation, it is suggested that PWM1 and
PWM2 pair, not PWM1 and PWM3 pair, should be used such
that the PSI# signal will enable or disable PWM2 with PWM1
in operation always. The overcurrent and way-overcurrent
protection level in two-to-one phase mode operation will be
adjusted as two-to-one as well.
The DCM mode operation is independent of PSI# for
ISL6260C. It responds to the DPRSLPVR and DPRSTP#.
Table 2 shows the operation modes of ISL6260C with
combinations of control logic.
When PSI# is de-asserted low, ISEN2 pin is connected to
the ISEN pins of the operational phases internally to keep
proper current balance and minimize the inductor current
overshoot and undershoot when the disabled phase is
enabled again.
TABLE 2. ISL6260C MODE OF OPERATIONS
DPRSLPVR DPRSTP# PSI#
IMVP-6+
Logic
Other
Logic
MODE OF
OPERATION
CPU
MODE
0
1
1
N phase CCM
Active
0
1
0
N-1 phase CCM Active
1
0
1
N phase DCM
1
0
0
N-1 phase DCM Deeper
sleep
0
0
1
N phase CCM
0
0
0
N-1 phase CCM
1
1
1
N phase CCM
1
1
0
N-1phase CCM
Deeper
sleep
Protection
The ISL6260C provides overcurrent, overvoltage, and
undervoltage protection. Overcurrent protection is related to
the voltage droop which is determined by the load line
requirement. After the load-line is set, the OCSET resistor can
be selected to detect overcurrent at any level of droop voltage.
For overcurrent less that 2.5x the OCSET level, the overload
condition must exist for 120µs in order to trip the OC fault
latch. This is shown in Figure 28.
19
For overload exceeding 2.5 times the OCSET level, the
PWM outputs will immediately shut off and PGOOD will go
low to maximize protection due to hard short circuit. This
protection was referred to as way-overcurrent or fast over
current, for short-circuit protections.
In addition, excessive phase unbalance due to gate driver
failure will be detected and will shut down the controller. The
phase unbalance is detected by the voltage on the ISEN pin.
If the ISEN pin voltage difference is greater than 9mV for 1ms,
the controller will latch off.
Undervoltage protection is independent of the overcurrent
limit. If the output voltage is less than the VID set value by
300mV or more, a fault will latch after 1ms in that condition.
The PWM outputs will turn off and PGOOD will go low. This
is shown in Figure 27. Note that most practical core voltage
regulators will have the overcurrent set to trip before the
-300mV undervoltage limit.
There are two levels of overvoltage protection with different
response. The first level of overvoltage protection is referred
to as PGOOD overvoltage protection. Basically, for output
voltage exceeding the set value by +200mV for 1ms, a fault
will be declared with PGOOD latched low.
All of the above faults have the same action taken: PGOOD
is latched low and the upper and lower power FETs are
turned off so that inductor current will decay through the FET
body diodes. This condition can be reset by bringing VR_ON
low or by bringing VDD below POR threshold. When these
inputs are returned to their high operating levels, a soft-start
will occur.
The second level of overvoltage protection behaves
differently. If the output exceeds 1.7V, an OV fault is
immediately declared, PGOOD is latched low and the
low-side FETs are turned on. The low-side FETs will remain
on until the output voltage is pulled down below 0.85V at
which time all FETs are turned off. If the output again rises
above 1.7V, the process is repeated. This affords the
maximum amount of protection against a shorted high-side
FET while preventing output ringing below ground. The 1.7V
OVP can not be reset with VR_ON, but requires that VDD be
lowered to reset. The 1.7V OV detector is active at all times
when the controller is enabled including after one of the
other faults occurs. This ensures the processor is protected
against high-side FET leakage while the FETs are
commanded off.
FN9259.3
June 21, 2010
ISL6260C
TABLE 3. SUMMARY OF THE FAULT PROTECTION AND RESET OPERATIONS OF ISL6260C
FAULT DURATION
PRIOR TO
PROTECTION
PROTECTION
ACTIONS
FAULT RESET
Overcurrent
120µs
PWMs tri-state, PGOOD latched low
VR_ON toggle or VDD toggle
Way-Overcurren (2.5X OC)
<2µs
PWMs tri-state, PGOOD latched low
VR_ON toggle or VDD toggle
Overvoltage 1.7V
Immediately
Low side MOSFET on until Vcore <0.85V, then PWM
tri-state, PGOOD latched low.
VDD toggle
Overvoltage +200mV
1ms
PWMs tri-state, PGOOD latched low
VR_ON toggle or VDD toggle
Undervoltage -300mV
1ms
PWMs tri-state, PGOOD latched low
VR_ON toggle or VDD toggle
Phase Current Unbalance
1ms
PWMs tri-state, PGOOD latched low
VR_ON toggle or VDD toggle
Over Temperature
Immediately
VR_TT# goes low
N/A
The ISL6260C has a thermal throttling feature. If the voltage
on the NTC pin goes below the 1.18V OT threshold, the
VR_TT# pin is pulled low indicating the need for thermal
throttling to the system oversight processor. No other action
is taken within the ISL6260C in response to NTC pin voltage.
Fault protection is summarized in Table 3.
Power Monitor
The power monitor signal is an analog output. Its magnitude
is proportional to the product of VCCSENSE and the voltage
difference between VDROOP and VO, which is the
programmed load line impedance (2.1mΩ) multiplied by load
current. The output voltage of the PMON pin is given by:
voltage is commanded to rise, and out of the SOFT capacitor
when the output voltage is commanded to fall.
The two slew rates are determined by the currents into the
SOFT pin. As can be seen in Figure 44, the SOFT pin has a
capacitance to ground. Also, the SOFT pin is the input to the
error amplifier and is, therefore, the commanded system
voltage. Depending on the state of the system, i.e. Start-Up
or Active mode, and the state of the DPRSLPVR pin, one of
the two currents shown in Figure 44 will be used to charge or
discharge this capacitor, thereby controlling the slew rate of
the commanded voltage. These currents can be found under
the Soft Current section of the “Electrical Specifications”
table on page 4.
VPMON = VCCSENSE*(VDROOP-VO)*17.5 (Volt)
ISL6260C
The power consumed by the CPU can be calculated by:
Pcpu = VPMON/(17.5*0.0021) (Watt)
where the 0.0021 is the load line impedance.The power
monitor load regulation is about 7Ω. Basically, within its
sourcing/sinking current capability range, when the power
monitor loading changes 1mA, the output of the power
monitor will change 7mV. The 7Ω impedance is associated
with the layout and packaging resistance of PMON pin inside
the IC. Compared to the load resistance on the power
monitor pin in practical applications, 7Ω output impedance
contributes no significance of error.
Component Selection and Application
Soft-Start and Mode Change Slew Rates
The ISL6260C uses 2 slew rates for various modes of
operation. The first is a slow slew rate, used to reduce inrush
current on start-up. It is also used to reduce audible noise
when entering or exiting Deeper Sleep Mode. A faster slew
rate is used to exit out of Deeper Sleep and to increase
system performance by achieving active mode regulation
more quickly. Note that the SOFT cap current is bidirectional
and is flowing into the SOFT capacitor when the output
20
ISS
I2
ERROR
AMPLIFIER
+
SOFT
CSOFT
+
VREF
FIGURE 44. SOFT PIN CURRENT SOURCES FOR FAST AND
SLOW SLEW RATES
The first current, labeled ISS, is given in the Specification
Table as 42µA. This current is used during Soft-Start. The
second current, I2 sums with ISS to get the large current
labeled IGV in the “Electrical Specifications” table on page 4.
This total current is typically 205µA with a minimum of
180µA.
The IMVP-6+™ specification reveals the critical timing
associated with regulating the output voltage. The symbol,
Slewrate, as given in the IMVP-6+™ specification, will
FN9259.3
June 21, 2010
ISL6260C
determine the choice of the SOFT capacitor, CSOFT, by
Equation 1:
I GV
C SOFT = -----------------------------------SLEWRATE
(EQ. 1)
Using a SLEWRATE of 10mV/µs, and the typical IGV value,
given in the Table Electrical Specifications on page 4 of
205µA, CSOFT is:
205μA
C SOFT = ------------------ = 0.0205μF
10mV
---------------1μs
(EQ. 2)
A choice of 0.015µF would guarantee a SLEWRATE of
10mV/µs is met for minimum IGV value, given in the
“Electrical Specifications” table on page 4.
Now this choice of CSOFT will then control the start-up
slewrate as well. One should expect the output voltage to
slew to the Boot value of 1.2V at a rate given by Equation 3:
I SS
42μA
dV
mV
- = ----------------------- = 2.8 --------------- = -----------------0.015μF
C SOFT
dt
μs
(EQ. 3)
Generally, when output voltage is approaching its steady
state, its dv/dt will slow down to prevent overshoot. In order
to compensate the slow-down effect, faster initial dv/dt slew
rates can be used with small soft capacitors such as 10nF to
achieve the desired overall dv/dt in the allocated time
interval.
Selecting RBIAS
To properly bias the ISL6260C, a reference current is
established by placing a 147kΩ, 1% tolerance resistor from
the RBIAS pin to ground. This will provide a highly accurate,
10μA current source from which OCSET reference current
can be derived.
Care should be taken in layout that the resistor is placed
very close to the RBIAS pin and that a good quality signal
ground is connected to the opposite side of the RBIAS
resistor. Do not connect any other components to this pin.
Capacitance on this pin would create instabilities and should
be avoided.
Start-up Operation - CLK_EN# and PGOOD
The ISL6260C provides a 3.3V logic output pin for
CLK_EN#. The 3V3 pin allows for a system 3.3V source to
be connected to separated circuitry inside the ISL6260C,
solely devoted to the CLK_EN# function. The output is a
3.3V CMOS signal with 4mA of source and sinking
capability. This implementation removes the need for an
external pull-up resistor on this pin, and due to the normal
level of this signal being a low, removes the leakage path
from the 3.3V supply to ground through the pull-up resistor.
This reduces 3.3V supply current, that would occur under
normal operation with a pull-up resistor, and prolongs battery
life. The 3.3V supply should be decoupled to digital ground,
not to analog ground for noise immunity.
21
As mentioned in the “Theory of Operation” on page 17 of this
datasheet, CLK_EN# is logic level high at start-up. When the
output voltage reaches 90% of Boot voltage, a counter is
enabled, it counts 13 switching cycles, about 43µs for
300kHz operation, then CLK_EN# goes low. This in turn
triggers an internal timer for the IMVP-6+_PWRGD signal.
This timer allows IMVP-6+_PWRGD to go high
approximately 7ms after CLK_EN# goes low.
Static Mode of Operation - Processor Die Sensing
Die sensing allows the Voltage Regulator to compensate for
various resistive drops in the power path and insure that the
voltage seen at the CPU die is the correct level independent
of load current.
The VSEN and RTN pins of the ISL626C are connected to
Kelvin sense leads at the die of the processor through the
processor socket. These signal names are VCC_SENSE and
VCC_SENSE respectively. This allows the Voltage Regulator
to tightly control the processor voltage at the die,
independent of layout inconsistencies and drops. This Kelvin
sense technique provides for extremely tight load line
regulation.
These traces should be laid out as noise sensitive traces.
For optimum load line regulation performance, the traces
connecting these two pins to the Kelvin sense leads of the
processor must be laid out in parallel and away from rapidly
rising voltage nodes (switching nodes) and other noisy
traces. To achieve optimum performance, place common
mode and differential mode RC filters to analog ground on
VSEN and RTN as shown in Figure 46. However, the filter
resistors should be in order of 10Ω so that they do not
interact with the 50kΩ input resistance of the differential
amplifier.
Due to the fact that the voltage feedback to the switching
regulator is sensed at the processor die, there exists the
potential of an overvoltage due to an open circuit feedback
signal, should the regulator be operated without the
processor installed. Due to this fact, we recommend the use
of the Ropn1 and Ropn2 connected to VOUT and ground as
shown in Figure 46. These resistors will provide voltage
feedback in the event that the system is powered up without
a processor installed. These resistors are typically 100Ω.
Setting the Switching Frequency - FSET
The R3 modulator scheme is not a fixed frequency PWM
architecture. The switching frequency can increase during
the application of a load to improve transient performance.
However, it also varies slightly due changes in input and
output voltage and output current, but this variation is
normally less than 10% in continuous conduction mode.
Refer to Figure 39. A resistor connected between the VW
and COMP pins of the ISL6260C adjusts the switching
window, and therefore adjusts the switching frequency. The
Rfset resistor that sets up the switching frequency of the
FN9259.3
June 21, 2010
ISL6260C
converter operating in CCM can be determined using the
following relationship, where Rfset is in kΩ and the switching
period is in μs.
(EQ. 4)
Rfset ( kΩ ) = ( Period ( μs ) – 0.29 ) × 2.33
In discontinuous conduction mode, (DCM), the ISL6260C
runs in period stretching mode. It should be noted that the
switching frequency in the Electrical Specification Table is
tested with the error amplifier output or Comp pin voltage at
2V. When Comp pin voltage is lower, the switching
frequency will not be at the tested value but can still maintain
the output voltage ripple within spec.
Voltage Regulator Thermal Throttling
lntel™ IMVP-6+™ technology supports thermal throttling of
the processor to prevent catastrophic thermal damage to the
voltage regulator. The ISL6260C feature a thermal monitor
which senses the voltage change across an externally
placed negative temperature coefficient (NTC) thermistor,
see Figure 45. Proper selection and placement of the NTC
thermistor allows for detection of a designated temperature
rise by the system.
Figure 45 shows the thermal throttling feature with
hysteresis. At low temperature, SW1 is on and SW2
connects to the 1.20V side. The total current going from NTC
pin is 60µA. The voltage on NTC pin is higher than threshold
voltage of 1.20V and the comparator output is low. VR_TT#
is pulling up high by the external resistor.
54µA
6µA
+
1.24V
Therefore, proper NTC thermistor has to be chosen such
that 2.96k resistor change will be corresponding to required
temperature hysteresis. Regular external resistor may need
to be in series with NTC resistors to meet the threshold
voltage values.
The following is an example.
For Panasonic NTC thermistor with B = 4700, its resistance
will drop to 0.03322 of its nominal at +105°C, and drop to
0.03956 of its nominal at +100°C. If the requirement for the
temperature hysteresis is (+105°C to +100°C), the required
resistance of NTC will be:
2.96kΩ
------------------------------------------------------ = 467kΩ
( 0.03956 – 0.03322 )
(EQ. 6)
Therefore a larger value thermistor, such as 470k NTC
should be used.
At 105°C, 470k NTC resistance becomes
(0.03322*470k) = 15.6k. With 60µA on NTC pin, the voltage
is only (15.6k*60µA) = 0.937V. This value is much lower than
the threshold voltage of 1.20V. Therefore, a resistor is
needed to be in series with the NTC. The required resistance
can be calculated by:
(EQ. 7)
4.42k is a standard resistor value. Therefore, the NTC
branch should have a 470k NTC and 4.42k resistor in series.
The part number for the NTC thermistor is ERTJ0EV474J. It
is a 0402 package. NTC thermistor will be placed in the hot
spot of the board.
-
Rs
(EQ. 5)
VR_TT#
SW1
RNTC
1.24V 1.20V
---------------- – ---------------- = 2.96k
54μA 60μA
1.20V
---------------- – 15.6kΩ = 4.4kΩ
60μA
NTC
+
VNTC
-
CPU operation and decrease the power consumption. When
the temperature goes down, the NTC thermistor voltage will
eventually go up. If NTC voltage increases to 1.24V, the
comparator will then be able to flip back. The external
resistance difference in these two conditions is:
SW2
1.20V
INTERNAL TO
ISL6260C
FIGURE 45. CIRCUITRY ASSOCIATED WITH THE THERMAL
THROTTLING FEATURE OF THE ISL6260C
When temperature increases, the NTC thermistor resistance
on NTC pin decreases. The voltage on NTC pin decreases
to a level lower than 1.20V. The comparator changes polarity
and turns SW1 off and throws SW2 to 1.24V. This pulls
VR_TT# low and sends the signal to start thermal throttle.
There is a 6µA current reduction on NTC pin and 40mV
voltage increase on threshold voltage of the comparator in
this state. The VR_TT# signal will be used to change the
22
FN9259.3
June 21, 2010
ISL6260C
IS E N 1
IS E N 2
10uA
OCSET
+
VO'
V D IF F
VSUM
VSUM
VO'
0 .0 1 u F
R opn1
VCC_SENSE
VSS_SENSE
Ropn2
R L2
VO'
DCR
+ V d c r2 -
RO2
C L2
VO'
L3
R L3
IS E N 3
10
0 .2 2 u F
RS3
-
RO1
L2
Ip h a s e 3
Rntc
VSEN
RS2
IS E N 1
IS E N 2
Cn
+
1 RTN
Rseries
DROOP
+
1 -
Rdrp2
+
D FB
C L1
R L1
Ip h a s e 2
VSUM
Rpar
+
DROOP
-
In te rn a l to
IS L 6 2 6 0
Σ
V d cr1
DCR
VSUM
VO'
VSUM
+
+
RS1
R OCSET
Rdrp1
OC
L1
IS E N 3
IS E N 2
IS E N 1
Ip h a s e 1
IS E N 3
+
V d cr3
Vout
-
DCR
C L3
RO3
C b u lk
VO'
to V o u t
ESR
T o P ro c e s s o r
S o c k e t K e lvin
C o n n e c tio n s
FIGURE 46. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DCR SENSING
Static Mode of Operation - Static Droop using DCR
Sensing
As previously mentioned, the ISL6260C has an internal
differential amplifier which provides for extremely accurate
voltage regulation at the die of the processor. The load line
regulation is also very accurate, and the process of selecting
the components for the appropriate load line droop is
explained here.
For DCR sensing, the process of compensation for DCR
resistance variation to achieve the desired load line droop
has several steps and is somewhat iterative.
In Figure 46 we show a 3 phase solution using DCR
sensing. There are two resistors around the inductor of each
phase. These are labeled RS and RO. These resistors are
used to sense the DC voltage drop across each inductor.
Each inductor will have a certain level of DC current flowing
through it, this current when multiplied by the DCR of the
inductor creates a small DC level of voltage. When this
voltage is summed with the other channels DC voltages, the
total DC load current can be derived.
RO is typically 5Ω to 10Ω. This resistor is used to tie the
outputs of all channels together and thus create a summed
average of the local CORE voltage output. RS is determined
through an understanding of both the DC and transient load
currents. This value will be covered in the next section.
However, it is important to keep in mind that the output of
each of these RS resistors are tied together to create the
VSUM voltage node. With both the outputs of RO and RS
tied together, the simplified model for the droop circuit can
be derived. This is presented in Figure 47.
phase and one RS resistor can replace the RS resistors of
each phase. The total DCR drop due to load current can be
replaced by a DC source, the value of which is given by
Equation 8.
I OUT × DCR
Vdcr EQV = ---------------------------------N
(EQ. 8)
where N is the number of channels designed for active
operation. Another simplification was done by reducing the
NTC network comprised of Rntc, Rseries and Rparallel,
given in Figure 46, to a single resistor given as Rn as shown
in Figure 47.
The first step in droop load line compensation is to adjust
Rn, ROEQV and RSEQV such that sufficient droop voltage
exists even at light loads between the VSUM and VO’ nodes.
We recognize that these components form a voltage divider.
As a rule of thumb we start with the voltage drop across the
Rn network, VN, to be 0.57 x Vdcr. This ratio provides for a
fairly reasonable amount of light load signal from which to
arrive at droop.
First we calculate the equivalent NTC network resistance,
Rn. Typical values that provide good performance are,
Rseries = 3.57k_1%, Rpar = 4.53k_1% and Rntc = 10kΩ
NTC, ERT-J1VR103J from Panasonic. Rn is then given by
Equation 9.
( Rseries + Rntc ) × Rpar
Rn = -------------------------------------------------------------------- = 3.4kΩ
Rseries + Rntc + Rpar
(EQ. 9)
In our second step we calculate the series resistance from
each phase to the Vsum node, labeled RS1, RS2 and RS3
in Figure 46.
Figure 47 shows the simplified model of the droop circuitry.
Essentially one resistor can replace the RO resistors of each
23
FN9259.3
June 21, 2010
ISL6260C
10uA
OCSET
+
VSUM
+
DROOP
-
+
+
1 +
1 -
RTN
VDIFF
Vdcr EQV = Iout ×
DROOP
+
VSEN
Cn
+
VSUM
RS
N
DFB
Rdrp1
Internal to
ISL6260
Σ
RS EQV =
Rdrp2
OC
VO'
VN
Rn =
DCR
N
(Rntc + Rseries ) × Rpar
(Rntc + Rseries ) + Rpar
VO'
RO EQV =
RO
N
FIGURE 47. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DCR SENSING
We do this using the assumption that we desire
approximately a 0.57 gain from the DCR voltage, Vdcr, to the
Rn network. We call this gain, G1.
(EQ. 10)
G1 = 0.57
After simplification, then RSEQV is given by Equation 11:
1
RS EQV = ⎛ -------- – 1⎞ Rn = 2.56kΩ
⎝ G1
⎠
(EQ. 11)
The individual resistors from each phase to the VSUM node,
labeled RS1, RS2 and RS3 in Figure 46, are then given by
Equation 12, where N is 3, for the number of channels in
active operation.
RS = N × RS EQV = 7.69kΩ
(EQ. 12)
Choosing RS = 7.68k_1% is a good choice. Once we know
the attenuation of the RS and RN network, we can then
determine the Droop amplifier Gain required to achieve the
load line. Setting Rdrp1 = 1k_1%, then Rdrp2 is can be
found using Equation 13.
N × Rdroop
Rdrp2 = ⎛ -------------------------------- – 1⎞ × Rdrp1
⎝ DCR × G1
⎠
(EQ. 13)
Setting N = 3 for 3 channel operation, Droop Impedance
(Rdroop) = 0.0021 (V/A) as per the Intel IMVP-6+
specification, DCR = 0.0012Ω typical, Rdrp1 = 1kΩ and the
attenuation gain (G1) = 0.57, Rdrp2 is then:
3 × 0.0021
Rdrp2 = ⎛ ------------------------------------ – 1⎞ × 1K = 8.21kΩ
⎝ 0.0012 × 0.57
⎠
24
(EQ. 14)
Rdrp2 is selected to be a 8.25k_1% resistor. Note, we
choose to ignore the RO resistors because they do not add
significant error.
These values are extremely sensitive to layout and coupling
factor of the NTC to the inductor. As only one NTC is
required in this application, this NTC should be placed as
close to the Channel 1 inductor as possible. And very
importantly, the PCB traces sensing the inductor voltage
should be go directly to the inductor pads.
Once the board has been laid out, some adjustments may
be required to adjust the full load droop voltage. This can be
accomplished by allowing the system to achieve thermal
equilibrium at full load, and then adjusting Rdrp2 to obtain
the appropriate load line slope.
To see whether the NTC has compensated the temperature
change of the DCR, the user can apply full load current and
wait for the thermal steady state and see how much the
output voltage will deviate from the initial voltage reading. A
good NTC thermistor compensation can limit the output
voltage drift to 2mV. If the output voltage is decreasing with
temperature increase, that ratio between the NTC thermistor
value and the rest of the resistor divider network has to be
increased. Users should use the ISL6260C evaluation board
component values and follow the evaluation board layout of
NTC as much as possible to minimize engineering time.
The 2.1mV/A load line should be adjusted by Rdrp2 based on
maximum current steps, not based on small current steps.
Basically, if the max current is 40A, the required droop voltage
is 84mV with 2.1mΩ load line impedance. The user should
have 40A load current on the converter and look for 84mV
droop. If the droop voltage is less than 84mV, for example,
80mV. The new value will be calculated by Equation 15:
FN9259.3
June 21, 2010
ISL6260C
Rdrp 2 _ new
=
84 mV
( Rdrp 1 + Rdrp 2 ) − Rdrp 1
80 mV
(EQ. 15)
For the best accuracy, the equivalent resistance on the DFB
and VSUM pins should be identical so that the bias current
of the droop amplifier does not cause an offset voltage. In
the example above, the resistance on the DFB pin is Rdrp1
in parallel with Rdrop2, that is, 1k in parallel with 8.21k or
890Ω. The resistance on the VSUM pin is Rn in parallel with
RSEQV or 3.4k in parallel with 2.56k or 1460Ω. The
mismatch in the effective resistances is 1460 - 890 = 570Ω.
To reduce the mismatch, multiply both Rdrp1 and Rdrp2 by
the appropriate factor. The appropriate factor in the example
is 1460/890 = 1.64.
Dynamic Mode of Operation - Dynamic Droop
using DCR Sensing
Droop is very important for load transient performance. If the
system is not compensated correctly, the output voltage
could sag excessively upon load application and potentially
create a system failure. The output voltage could also take a
long period of time to settle to its final value.
The L/DCR time constant of the inductor must be matched to
the Rn*Cn time constant as shown in Equation 16:
⎛ Rn × RS EQV⎞
L
-⎟ × Cn
------------- = ⎜ ---------------------------------DCR
⎝ Rn + RS EQV⎠
(EQ. 16)
Solving for Cn we now have Equation 17:
L
------------DCR
Cn = ----------------------------------------⎛ Rn × RS EQV⎞
⎜ -----------------------------------⎟
⎝ Rn + RS EQV⎠
(EQ. 17)
Note, RO was neglected. As long as the inductor time
constant matches the droop circuit RC time constants as
given above, the transient performance will be optimum. The
selection of Cn may require a slight adjustment to correct for
layout inconsistencies and component tolerance. For the
example of L = 0.5µH, Cn is calculated in Equation 18.
0.5μH
-----------------0.0012
Cn = ------------------------------------------------- = 28.5nF
3.4kΩ × 2.56kΩ⎞
⎛ -----------------------------------------⎝ 3.4kΩ + 2.56kΩ⎠
(EQ. 18)
The value of this capacitor is selected to be 27nF. As the
inductors tend to have 20% to 30% tolerances, this cap
generally will be tuned on the board by examining the
transient voltage. If the output voltage transient has an initial
dip, lower than the voltage required by the load line, and is
slowly increasing back to the steady state, the cap should be
increased and vice versa. It is better to have the cap value a
little bigger to cover the tolerance of the inductor to prevent
the output voltage from going lower than the spec. This cap
needs to be a high grade cap like X7R with low tolerance.
There is another consideration in order to achieve better
time constant match mentioned above. The NPO/COG
25
(class-I) capacitors have only 5% tolerance and a very good
thermal characteristics. But those caps are only available in
small capacitance values. In order to use such capacitors,
the resistors and thermistors surrounding the droop voltage
sensing and droop amplifier has to be resized up to 10x to
reduce the capacitance by 10x. But attention has to be paid
in balancing the impedance of droop amplifier in this case.
Dynamic Mode of Operation - Compensation
Parameters
Considering the voltage regulator as a black box with a
voltage source controlled by VID and a series impedance, in
order to achieve the 2.1mV/A load line, the series
impedance inside the black box needs to be 2.1mΩ. The
compensation design has to ensure the output impedance of
the converter be lower than 2.1mΩ. There is a mathematical
calculation file available to the user. The power stage
parameters such as L and Cs are needed as the input to
calculate the compensation component values. Attention
has be paid to the input resistor to the FB pin. Too high of a
resistor will cause an error to the output voltage regulation
because of bias current flowing in the FB pin. It is better to
keep this resistor below 3k when using this file.
Static Mode of Operation - Current Balance using
DCR or Discrete Resistor Current Sensing
Current Balance is achieved in the ISL6260C through the
matching of the voltages present on the ISEN pins. The
ISL6260C adjusts the duty cycles of each phase to maintain
equal potentials on the ISEN pins. RL and CL around each
inductor, or around each discrete current resistor, are used
to create a rather large time constant such that the ISEN
voltages have minimal ripple voltage and represent the DC
current flowing through each channel’s inductor. For
optimum performance, RL is chosen to be 10kΩ and CL is
selected to be 0.22µF. When discrete resistor sensing is
used, a capacitor of 10nF should be placed in parallel with
RL to properly compensate the current balance circuit.
ISL6260C uses RC filter to sense the average voltage on
phase node and forces the average voltage on the phase
node to be equal for current balance. Even though the
ISL6260C forces the ISEN voltages to be almost equal, the
inductor currents will not be exactly the same. Take DCR
current sensing as example, two errors have to be added to
find the total current imbalance. 1) Mismatch of DCR: If the
DCR has a 5% tolerance, then the resistors could mismatch
by 10% worst case. If each phase is carrying 20A then the
phase currents mismatch by 20A*10% = 2A. 2) Mismatch of
phase voltages/offset voltage of ISEN pins. The phase
voltages are within 2mV of each other by current balance
circuit. The error current that results is given by 2mV/DCR. If
DCR = 1mΩ then the error is 2A.
In the above example, the two errors add to 4A. For a two
phase DC/DC, the currents would be 22A in one phase and
18A in the other phase. In the above analysis, the current
FN9259.3
June 21, 2010
ISL6260C
balance can be calculated with 2A/20A = 10%. This is the
worst case calculation, for example, the actual tolerance of
two 10% DCRs is 10%*√(2) = 7%.
There are provisions to correct the current imbalance due to
layout or to purposely divert current to certain phase for better
thermal management. Customer can put a resistor in parallel
with the current sensing capacitor on the phase of interest in
order to purposely increase the current in that phase. It is
highly recommended to use symmetrical layout in order to
achieve natural current balance.
In the case the PC board trace resistance from the inductor
to the microprocessor are not the same on all three phases,
the current will not be balanced. On the phases that have too
much trace resistance a resistor can be added in parallel
with the ISEN capacitor that will correct for the poor layout.
not required. Secondly, there is no time constant matching
required, therefore, the Cn component is not needed to
match the L/DCR time constant, but this component does
indeed provide noise immunity, especially to noise voltage
caused by the ESL of the current sensing resistors. A 47pF
capacitor can be used for such purposes.
The Rs values in the previous section, Rs = 7.68k_1% are
sufficient for this approach.
Now, the input to the Droop amplifier is the Vrsense voltage.
This voltage is given by Equation 21:
Rsense
Vrsense = ---------------------- × I OUT
N
(EQ. 21)
The gain of the Droop amplifier, G2, must be adjusted equal
to the load line impedance. We use Equation 22:
An estimate of the value of the resistor is as shown in
Equation 19:
Rdroop
G2 = ---------------------- × N
Rsense
Risen∗ [ 2∗ Rdcr – Rtrace – Rmin ) ]
Rtweak = -----------------------------------------------------------------------------------------------[ 2 ( ( Rtrace ) – Rmin ) ]
Assuming N = 3, Rdroop = 0.0021(V/A) as per the Intel
IMVP-6+ specification, Rsense = 0.001Ω, we obtain
G2 = 6.3.
(EQ. 19)
(EQ. 22)
where Risen is the resistance from the phase node to the
ISEN pin; usually 10kΩ. Rdcr is the DCR resistance of the
inductor. Rtrace is the trace resistance from the inductor to
the microprocessor on the phase that needs to be tweaked.
It should be measured with a good microΩ meter. Rmin is
the trace resistance from the inductor to the microproccessor
on the phase with the least resistance.
The values of Rdrp1 and Rdrp2 are selected to satisfy two
requirements. First, the ratio of Rdrp2 and Rdrp1 determine
the gain G2 = (Rdrp2/Rdrp1)+1. Second, the parallel
combination of Rdrp1 and Rdrp2 should equal the parallel
combination of the Rs resistors. Combining these
requirements gives:
For example, if the PC board trace on one phase is 0.5mΩ
and on another trace is 0.3mΩ; and if the DCR is 1.2mΩ;
then the tweaking resistor is as shown in Equation 20:
Rdrp2 = (G2-1) * Rdrp1
10kΩ∗ [ 2∗ 1.2 – ( 0.5 – 0.3 ) ]
Rtweak = ------------------------------------------------------------------------ = 55kΩ
[ 2∗ ( 0.5 – 0.3 ) ]
(EQ. 20)
For extremely unsymmetrical layout causing phase current
unbalance, ISL6260C applications schematics can be
modified to correct the problem.
Droop using Discrete Resistor Sensing - Static/
Dynamic Mode of Operation
When choosing current sense resistor, not only the tolerance
of the resistance is important, but also the TCR. And its
combined tolerance at a wide temperature range should be
calculated.
Figure 48 shows the equivalent circuit of a discrete current
sense approach. Figure 40 shows the simplified schematic
of this approach.
For discrete resistor current sensing circuit, the droop circuit
parameters can be solved the same way as the DCR
sensing approach with a few slight modifications.
First, there is no NTC required for thermal compensation,
therefore, the Rn resistor network in the previous section is
26
Rdrp1 = G2/(G2-1) * Rs/N
In the example above, Rs = 7.68k, N = 3, and G2 = 6.3 so
Rdrp 3k and Rdrp2 is 15.8kΩ.
These values are extremely sensitive to layout. Once the
board has been laid out, some tweaking may be required to
adjust the full load Droop. This is fairly easy and can be
accomplished by allowing the system to achieve thermal
equilibrium at full load, and then adjusting Rdrp2 to obtain
the desired Droop value.
Power Monitor
The power monitor signal tracks the inductor current. Due to
the dynamic operation of the CPU, the inductor current is
pulsating and the power monitor signal needs to be filtered.
If the RC filter is followed by an A/D converter, the input
impedance of the A/D converter needs to be much larger
than the resistor used for the RC filter. Otherwise, the input
impedance of the A/D converter and the RC filter resistor will
construct a resistor divider causing the A/D converter
reading incorrect information. It is desirable to choose a
small RC filter resistor in order to reduce the resistor divider
effect. The ISL6260C comes with a very strong current
sinking capability, users can use kΩ resistors for the RC
filter. Some A/D converters might have 100kΩ input
impedance, 1kΩ resistor will cause 1% error. As shown in
FN9259.3
June 21, 2010
10uA
OCSET
+
VSUM
+
DROOP
-
VDIFF
+
+
1 -
DROOP
+
1 -
RTN
RS
N
DFB
+
VSEN
VO'
Cn
+
RS EQV =
VSUM
Rdrp1
Internal to
ISL6260
Σ
Roc
Rdrp2
OC
Voc
-
+
ISL6260C
Vrsense EQV = Iout ×
Rsense
N
VN
VO'
ROEQV =
RO
N
FIGURE 48. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DISCRETE RESISTOR SENSING
Figure 36, when the CPU is at 2.5A load, PMON can still
sink 0.6mA current. This allows the RC filter capacitor to
discharge when the CPU is at low current, thus providing
correct average power information on the capacitor.
Fault Protection - Overcurrent Fault Setting
As previously described, the overcurrent protection of the
ISL6260C is related to the Droop voltage. Previously we
have calculated that the Droop Voltage = ILoad * Rdroop,
where Rdroop is the load line slope specified as 0.0021
(V/A) in the Intel IMVP-6+ specification. Knowing this
relationship, the overcurrent protection threshold can be set
up as a voltage droop level. Knowing this voltage droop
level, one can program in the appropriate drop across the
Roc resistor. This voltage drop will be referred to as Voc.
Once the droop voltage is greater than Voc, the PWM drives
will turn off and PGOOD will go low.
The selection of Roc is given below in Equation 23.
Assuming we desire an overcurrent trip level, Ioc, of 55A,
and knowing from the Intel Specification that the load line
slope, Rdroop is 0.0021 (V/A), we can then calculate for Roc
as shown in Equation 23.
Ioc × Rdroop
55 × 0.0021
Roc = ------------------------------------- = ------------------------------- = 11.5kΩ
10μA
10x10 – 6
(EQ. 23)
Note, if the droop load line slope is not -0.0021 (V/A) in the
application, the overcurrent setpoint will differ from
predicted.
A capacitor may be added in parallel with Roc to improve
noise rejection but the Roc*capacitor time constant cannot
exceed 20µs. Do not remove Roc if overcurrent protection is
not desired. The maximum Roc is 30k.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
27
FN9259.3
June 21, 2010
ISL6260C
Package Outline Drawing
L40.6x6
40 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 3, 10/06
4X 4.5
6.00
36X 0.50
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
40
31
30
1
6.00
4 . 10 ± 0 . 15
21
10
0.15
(4X)
11
20
TOP VIEW
0.10 M C A B
40X 0 . 4 ± 0 . 1
4 0 . 23 +0 . 07 / -0 . 05
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 90 ± 0 . 1
(
C
BASE PLANE
( 5 . 8 TYP )
SEATING PLANE
0.08 C
SIDE VIEW
4 . 10 )
( 36X 0 . 5 )
C
0 . 2 REF
5
( 40X 0 . 23 )
0 . 00 MIN.
0 . 05 MAX.
( 40X 0 . 6 )
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
28
FN9259.3
June 21, 2010
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