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ISL6532
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ACPI Regulator/Controller for
Dual Channel DDR Memory Systems
The ISL6532 provides a complete ACPI compliant power
solution for up to 4 DIMM dual channel DDR/DDR2 memory
systems. Included are both a synchronous buck controller
and integrated LDO to supply VDDQ with high current during
S0/S1 states and standby current during S3 state. During
Run mode, a fully integrated sink-source regulator generates
an accurate (VDDQ/2) high current VTT voltage without the
need for a negative supply. A buffered version of the VDDQ/2
reference is provided as VREF.
September 12, 2013
FN9112.4
Features
• Generates 2 Regulated Voltages
- Synchronous Buck PWM Controller with Standby LDO
- 3A Integrated Sink/Source Linear Regulator with
Accurate VDDQ/2 Divider Reference.
- Glitch-free Transitions During State Changes
• ACPI Compliant Sleep State Control
• Integrated VREF Buffer
• PWM Controller Drives Low Cost N-Channel MOSFETs
• 250kHz Constant Frequency Operation
The switching PWM controller drives two N-Channel
MOSFETs in a synchronous-rectified buck converter
topology. The synchronous buck converter uses voltagemode control with fast transient response. Both the switching
regulator and integrated standby LDO provide a maximum
static regulation tolerance of ±2% over line, load, and
temperature ranges. The output is user-adjustable by means
of external resistors down to 0.8V.
• Tight Output Voltage Regulation
- Both Outputs: ±2% Over Temperature
Switching the memory core output between the PWM
regulator and the standby LDO during state transitions is
accomplished smoothly via the internal ACPI control
circuitry. The NCH signal provides synchronized switching of
a backfeed blocking switch during the transitions eliminating
the need to route 5V Dual to the memory supply.
• Over Current Protection and Under/Over-Voltage
Monitoring of Both Outputs
An integrated soft-start feature brings VDDQ into regulation in
a controlled manner when returning to S0/S1 state from
S4/S5 or mechanical off states. During S0 the PGOOD signal
indicates that all supplies are within spec and operational.
• 5V or 3.3V Down Conversion
• Fully-Adjustable Outputs with Wide Voltage Range: Down
to 0.8V supports DDR and DDR2 Specifications
• Simple Single-Loop Voltage-Mode PWM Control Design
• Fast PWM Converter Transient Response
• Integrated Thermal Shutdown Protection
• QFN Package Option
- QFN Compliant to JEDEC PUB95 MO-220 QFN - Quad
Flat No Leads - Product Outline
- QFN Near Chip Scale Package Footprint; Improves
PCB Efficiency, Thinner in Profile
• Pb-free available
Each output is monitored for under and over-voltage events.
Current limiting is included on the VTT and VDDQ standby
regulators. Thermal shutdown is integrated.
Applications
Pinout
• Graphics cards - GPU and memory supplies
ISL6532 (QFN) TOP VIEW
5VSBY
1
GND
2
• Single and Dual Channel DDR Memory Power Systems in
ACPI compliant PCs
UGATE
LGATE
P12V
S5#
S3#
• ASIC power supplies
• Embedded processor and I/O supplies
20
19
18
17
16
• DSP supplies
15 NCH
Ordering Information
14 PGOOD
GND
21
VTT
3
VTT
4
12 COMP
VDDQ
5
11
13 GND
FB
PART NUMBER
TEMP. RANGE
(oC)
PACKAGE
PKG.
DWG. #
ISL6532CR
0 to 70
20 Ld 6x6 QFN
L20.6x6
ISL6532CRZ
(See Note)
0 to 70
20 Ld 6x6 QFN
(Pb-free)
L20.6x6
9
10
VREF_OUT
VREF_IN
VTTSNS
8
P5VSBY
7
VDDQ
*Add “-T” suffix to part number for tape and reel packaging.
6
1
NOTE: Intersil Pb-free products employ special Pb-free material sets; molding
compounds/die attach materials and 100% matte tin plate termination finish, which
is compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that meet or
exceed the Pb-free requirements of IPC/JEDEC J Std-020B.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2002-2004, 2013. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
Block Diagram
P3V3SBY
SLP_S3#
VDDQ S3
REGULATOR
SLP_S5#
5VSBY
VOLTAGE
REFERENCE
0.800V
NCH
0.680V (-15%)
VDDQ(2)
2
0.920V (+15%)
5V
VTTSNS
POR
VTT
REG
VTT(2)
S3
S0
DISABLE
12V
POR
PWM ENABLE
S0/S3
P12V
SOFT-START
RU
PWM
EA1
VREF_IN
COMP
OSCILLATOR
{
PWM
LOGIC
UGATE
250kHz
UV/OV
RL
LGATE
UV/OV
VREF_OUT
PGOOD
FB
COMP
GND
ISL6532
{
SLEEP,
SOFT-START,
PGOOD,
AND FAULT
LOGIC
ISL6532
Simplified Power System Diagram
12V
5VSBY
5V
SLEEP
STATE
LOGIC
SLP_S3
SLP_S5
Q1
VDDQ
PWM
CONTROLLER
+
Q2
5VSBY/3V3SBY
STANDBY
LDO
ISL6532
VREF
VTT
REGULATOR
VTT
+
Typical Application - 5V or 3.3V Input
5VSBY
+12V
+3.3V
+5V OR +3.3V
P12V
P5VSBY
5VSBY
CBP
RNCH
PGOOD
VDDQ
S3#
SLP_S3
NCH
S5#
SLP_S5
VREF_OUT
VREF
+
CIN
VREF_IN
UGATE
+
Q1
CSS
ISL6532
VTT
VTT
VDDQ
VTT
VDDQ
CVTT
VTTSNS
FB
COMP
GND
2.5V
+
LGATE
+
3
VDDQ
LOUT
Q2
CVDDQ
ISL6532
Typical Application - Input From 5V Dual
5VSBY
+12V
+3.3V
5V DUAL
P12V
P5VSBY
5VSBY
CBP
RNCH
PGOOD
VDDQ
S3#
SLP_S3
NCH
S5#
SLP_S5
VREF_OUT
VREF
+
CIN
VREF_IN
UGATE
Q1
CSS
ISL6532
VTT
VTT
VDDQ
VTT
VDDQ
CVTT
VTTSNS
FB
COMP
GND
2.5V
+
LGATE
+
4
VDDQ
LOUT
Q2
CVDDQ
ISL6532
Absolute Maximum Ratings
Thermal Information
5VSBY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +7V
P12V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +14V
UGATE, LGATE, NCH . . . . . . . . . . . . . . GND - 0.3V to P12V + 0.3V
All other Pins . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5VSBY + 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Level 2
Thermal Resistance (Typical, Notes 1, 2)
θJA (oC/W) θJC (oC/W)
QFN Package . . . . . . . . . . . . . . . . . . .
32
5
Maximum Junction Temperature (Plastic Package) . . . . . . . 150oC
Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC
Recommended Operating Conditions
Supply Voltage on 5VSBY . . . . . . . . . . . . . . . . . . . . . . . . +5V ±10%
Supply Voltage on P12V . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10%
Supply Voltage on 3V3SBY . . . . . . . . . . . . . . . . . . . . . +3.3V ±10%
Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . 0oC to 70oC
Junction Temperature Range . . . . . . . . . . . . . . . . . . 0oC to 125oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams and Typical Application Schematics
Electrical Specifications
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
5VSBY SUPPLY CURRENT
Nominal Supply Current
ICC_S0
S3# & S5# HIGH, UGATE/LGATE Open
3.00
5.25
7.25
mA
ICC_S3
S3# LOW, S5# HIGH, UGATE/LGATE
Open
3.50
-
4.75
mA
ICC_S5
S5# LOW, S3# Don’t Care,
UGATE/LGATE Open
300
-
800
μA
Rising 5VSBY POR Threshold
4.00
-
4.35
V
Falling 5VSBY POR Threshold
3.60
-
3.95
V
Rising P12V POR Threshold
10.0
-
10.5
V
Falling P12V POR Threshold
8.80
-
9.75
V
POWER-ON RESET
OSCILLATOR AND SOFT-START
PWM Frequency
fOSC
220
250
280
kHz
Ramp Amplitude
ΔVOSC
-
1.5
-
V
Error Amp Reset Time
tRESET
S5# LOW to S5# HIGH
6.5
-
9.5
ms
tSS
S5# LOW to S5# HIGH
6.5
-
9.5
ms
-
0.800
-
V
-2.0
-
+2.0
%
-
80
-
dB
GBWP
15
-
-
MHz
SR
-
6
-
V/μs
S3# Transition Level
VS3
-
1.5
-
V
S5# Transition Level
VS5
-
1.5
-
V
VDDQ Soft-Start Interval
REFERENCE VOLTAGE
Reference Voltage
VREF
System Accuracy
PWM CONTROLLER ERROR AMPLIFIER
DC Gain
Guaranteed By Design
Gain-Bandwidth Product
Slew Rate
STATE LOGIC
5
ISL6532
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams and Typical Application Schematics (Continued)
Electrical Specifications
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
PWM CONTROLLER GATE DRIVERS
UGATE and LGATE Source
IGATE
-
-0.8
-
A
UGATE and LGATE Sink
IGATE
-
0.8
-
A
-
-
6
mA
9.0
9.5
10
V
P5VSBY = 5.0V
-
-
650
mA
P5VSBY = 3.3V
-
-
550
mA
NCH BACKFEED CONTROL
NCH Current Sink
INCH
NCH Trip Level
VNCH
NCH = 0.8V
VDDQ STANDBY LDO
Output Drive Current
VTT REGULATOR
Upper Divider Impedance
RU
-
2.5
-
kΩ
Lower Divider Impedance
RL
-
2.5
-
kΩ
IVREF_OUT
-
-
2
mA
-3
-
3
A
VREF_OUT Buffer Source Current
Maximum VTT Load Current
IVTT_MAX
Periodic load applied with 30% duty cycle
and 10ms period using ISL6532EVAL1
evaluation board (see Application Note
AN1055)
PGOOD
PGOOD Rising Threshold
VVTTSNS/VVDDQ S3# & S5# HIGH
-
57.5
-
%
PGOOD Falling Threshold
VVTTSNS/VVDDQ S3# & S5# HIGH
-
45.0
-
%
PROTECTION
VDDQ OV Level
VFB/VREF
S3# & S5# HIGH
-
115
-
%
VDDQ UV Level
VFB/VREF
S3# & S5# HIGH
-
85
-
%
By Design
-
140
-
°C
Thermal Shutdown Limit
TSD
Functional Pin Description
GND (Pin 2, 13, 21)
5VSBY (Pin 1)
The GND terminals of the ISL6532 provide the return path
for the VTT LDO, Standby LDO and switching MOSFET gate
drivers. High ground currents are conducted directly through
the exposed paddle of the QFN package which must be
electrically connected to the ground plane through a path as
low in inductance as possible.
5VSBY is the bias supply of the ISL6532. It is typically
connected to the 5V standby rail of an ATX power supply.
During S4/S5 sleep states the ISL6532 enters a reduced
power mode and draws less than 1mA (ICC5) from the
5VSBY supply. This pin should be locally bypassed using a
0.1μF capacitor.
P12V (Pin 18)
P12V provides the gate drive current to the switching
MOSFETs of the PWM power stage. The VTT regulation
circuit is also powered by P12V. P12V is only required
during S0/S1/S2 operation. P12V is typically connected to
the +12V rail of an ATX power supply.
P5VSBY (Pin 8)
This pin provides the VDDQ output power during the S3
sleep state. The regulator is capable of providing standby
VDDQ power from either a 5V or 3.3V source.
6
UGATE (Pin 20)
UGATE drives the upper (control) FET of the VDDQ
synchronous buck switching regulator. UGATE is driven
between GND and P12V.
LGATE (Pin 19)
LGATE drives the lower (synchronous) FET of the VDDQ
synchronous buck switching regulator. LGATE is driven
between GND and P12V.
ISL6532
FB (Pin 11) and COMP (Pin 12)
NCH (Pin 15)
The VDDQ switching regulator employs a single voltage
control loop. FB is the negative input to the voltage loop
error amplifier. The positive input of the error amplifier is
connected to a precision 0.8V reference and the output of
the error amplifier is connected to the COMP pin. The VDDQ
output voltage is set by an external resistor divider
connected to FB. With a properly selected divider, VDDQ can
be set to any voltage between the power rail (reduced by
converter losses) and the 0.8V reference. Loop
compensation is achieved by connecting an AC network
across COMP and FB.
NCH is an open-drain output that controls the MOSFET
blocking backfeed from VDDQ to the input rail during sleep
states. A 2kΩ or larger resistor is to be tied between the 12V
rail and the NCH pin. Until the voltage on the NCH pin
reaches the NCH trip level, the PWM is disabled.
The FB pin is also monitored for under and over-voltage
events.
VDDQ (Pins 5, 6)
The VDDQ pins should be connected externally together to
the regulated VDDQ output. During S0/S1 states, the VDDQ
pins serve as inputs to the VTT regulator and to the VTT
Reference precision divider. During S3 (Suspend to RAM)
state, the VDDQ pins serve as an output from the integrated
standby LDO.
If NCH is not actively utilized, it still must be tied to the 12V
rail through a resistor. For systems using 5V dual as the
input to the switching regulator, a time constant, in the form
of a capacitor, can be added to the NCH pad to delay start of
the PWM switcher until the 5V dual has switched from
5VSBY to 5VATX.
PGOOD (Power Good) (Pin 14)
Power Good is an open-drain logic output that changes to a
logic low if the VTT regulator is out of regulation in S0/S1/S2
state. PGOOD will always be low in any state other than
S0/S1/S2.
S5# (Pin 17)
This pin accepts the SLP_S5# sleep state signal.
S3# (Pin 16)
This pin accepts the SLP_S3# sleep state signal.
VTT (Pins 3, 4)
The VTT pins should be connected together. During S0/S1
states, the VTT pins serve as the outputs of the VTT linear
regulator. During any sleep state, the VTT regulator is
disabled.
VTTSNS (Pin 7)
VTTSNS is used as the feedback for control of the VTT linear
regulator. Connect this pin to the VTT output at the physical
point of desired regulation.
VREF_OUT (Pin 9)
VREF_OUT is a buffered version of VTT and also acts as the
reference voltage for the VTT linear regulator. It is
recommended that a minimum capacitance of 0.1μF be
connected between VDDQ and VREF_OUT and also
between VREF_OUT and GND for proper operation.
VREF_IN (Pin 10)
A capacitor, CSS, connected between VREF_IN and ground
is required. This capacitor and the parallel combination of
the Upper and Lower Divider Impedance (RU||RL), sets the
time constant for the start up ramp when transitioning from
S3 to S0/S1/S2.
The minimum value for CSS can be found through the
following equation:
C VTTOUT ⋅ V DDQ
C SS > -----------------------------------------------10 ⋅ 2A ⋅ R U || R L
Functional Description
Overview
The ISL6532 provides complete control, drive, protection
and ACPI compliance for a regulator powering DDR memory
systems. It is primarily designed for computer applications
powered from an ATX power supply. A 250kHz Synchronous
Buck Regulator with a precision 0.8V reference provides the
proper Core voltage to the system memory of the computer.
An internal LDO regulator with the ability to both sink and
source current and an externally available buffered
reference that tracks the VDDQ output by 50% provides the
VTT termination voltage.
ACPI compliance is realized through the SLP_S3 and
SLP_S5 sleep signals and through monitoring of the 12V
ATX bus.
Initialization
The ISL6532 automatically initializes upon receipt of input
power. Special sequencing of the input supplies is not
necessary. The Power-On Reset (POR) function continually
monitors the input bias supply voltages. The POR monitors
the bias voltage at the 5VSBY and P12V pins. The POR
function initiates soft-start operation after the bias supply
voltages exceed their POR thresholds.
ACPI State Transitions
The calculated capacitance, CSS, will charge the output
capacitor bank on the VTT rail in a controlled manner without
reaching the current limit of the VTT LDO.
7
Cold Start (S5/S4 to S0 Transition)
At the onset of a mechanical start, the ISL6532 receives it’s
bias voltage from the 5V Standby bus (5VSBY). As soon as
the SLP_S3 and SLP_S5 signals have transitioned HIGH,
ISL6532
the ISL6532 starts an internal counter. Following a cold start
or any subsequent S5 state, state transitions are ignored
until the system enters S0/S1. None of the regulators will
begin the soft start procedure until the 5V Standby bus has
exceeded POR, the 12V bus has exceeded POR and VNCH
has exceeded the trip level.
Once all of these conditions are met, the PWM error
amplifier will first be reset by internally shorting the COMP
pin to the FB pin. This reset lasts for 2048 clock cycles which
is typically 8.2ms (one clock cycle = 1/fOSC). The digital soft
start sequence will then begin.
The PWM error amplifier reference input is clamped to a level
proportional to the soft-start voltage. As the soft-start voltage
slews up, the PWM comparator generates PHASE pulses of
increasing width that charge the output capacitor(s). The
internal VTT LDO will also soft start through the reference that
tracks the output of the PWM regulator. The soft start lasts for
2048 clock cycles, which is typically 8.2ms. This method
provides a rapid and controlled output voltage rise.
Figure 1 shows the soft start sequence for a typical cold
start. Due to the soft start capacitance, CSS, on the
S3
S5
12VATX 2V/DIV
5VSBY
1V/DIV
VDDQ
500mV/DIV
VTT
500mV/DIV
PGOOD
5V/DIV
12V POR
The VDDQ rail will be supported in the S3 state through the
standby VDDQ LDO. When S3 transitions LOW, the Standby
regulator is immediately enabled. The switching regulator is
disabled synchronous to the switching waveform. The shut
off time will range between 4 and 8µs. The standby LDO is
capable of supporting up to 650mA of load with P5VSBY tied
to the 5V Standby Rail. The standby LDO may receive input
from either the 3.3V Standby rail or the 5V Standby rail
through the P5VSBY pin. It is recommended that the 5V
Standby rail be used as the current delivery capability of the
LDO is greater.
Sleep to Active (S3 to S0 Transition)
When SLP_S3 transitions from LOW to HIGH with SLP_S5
held HIGH and after the 12V rail exceeds POR, the ISL6532
will enable the VDDQ switching regulator, disable the VDDQ
standby regulator, enable the VTT LDO and force the NCH
pin to a high impedance state turning on the blocking
MOSFET. The internal short between the VTT reference and
the VTT rail is released. Upon release of the short, the
capacitor on VREF_IN is then charged up through the
internal resistor divider network. The VTT output will follow
this capacitor charge-up, acting as the S3 to S0 transition
soft start for the VTT rail. The PGOOD comparator is
enabled only after 2048 clock cycles, or typically 8.2ms,
have passed following the S3 transition to a HIGH state.
S3
S5
2048 CLOCK
CYCLES
SOFT START ENDS
SOFT START
PGOOD COMPARATOR
INITIATES
ENABLED
FIGURE 1. TYPICAL COLD START
VREF_IN pin, the S5 to S0 transition profile of the VTT rail
will have a more rounded features at the start and end of the
soft start whereas the VDDQ profile has distinct starting and
ending points to the ramp up.
By directly monitoring 12VATX and the SLP_S3 and
SLP_S5 signals, the ISL6532 can achieve PGOOD status
significantly faster than other devices that depend on the
Latched_Backfeed_Cut signal for timing.
Active to Sleep (S0 to S3 Transition)
When SLP_S3 goes LOW with SLP_S5 still HIGH, the
ISL6532 will disable the VTT linear regulator. The VDDQ
standby regulator will be enabled and the VDDQ switching
regulator will be disabled. NCH is pulled low to disable the
8
12VATX 2V/DIV
VDDQ
VTT FLOATING
2048 CLOCK
CYCLES
backfeed blocking MOSFET. PGOOD will also transition
LOW. When VTT is disabled, the internal reference for the
VTT regulator is internally shorted to the VTT rail. This allows
the VTT rail to float. When floating, the voltage on the VTT
rail will depend on the leakage characteristics of the memory
and MCH I/O pins. It is important to note that the VTT rail
may not bleed down to 0V.
500mV/DIV
VTT
500mV/DIV
PGOOD
5V/DIV
2048 CLOCK
CYCLES
12V POR
PGOOD COMPARATOR
ENABLED
FIGURE 2. TYPICAL S3 to S0 STATE TRANSITION
Figure 2 illustrates a typical state transition from S3 to S0. It
should be noted that the soft start profile of the VTT LDO
output will vary according to the value of the capacitor on the
VREF_IN pin.
ISL6532
Active to Shutdown (S0 to S4/S5 Transition)
Shoot-Through Protection
When the system transitions from active, S0, state to
shutdown, S4/S5, state, the ISL6532 IC disables all
regulators and forces the PGOOD pin and the NCH pin
LOW.
A shoot-through condition occurs when both the upper and
lower MOSFETs are turned on simultaneously, effectively
shorting the input voltage to ground. To protect from a shootthrough condition, the ISL6532 incorporates specialized
circuitry which insures that complementary MOSFETs are
not ON simultaneously.
Over/Under Voltage Protection.
Both the internal VTT LDO and the VDDQ regulator are
protected from faults through internal Over/Under voltage
detection circuitry. If either rail falls below 85% of the targeted
voltage, then an undervoltage event is tripped. An under
voltage will disable all regulators for a period of 3 soft-start
cycles, after which a normal soft-start is initiated. If the output
remains under 85% of target, the regulators will continue to be
disabled and soft-started in a hiccup mode until the fault is
cleared. See Figure 3.
The adaptive shoot-through protection utilized by the VDDQ
regulator looks at the lower gate drive pin, LGATE, and the
upper gate drive pin, UGATE, to determine whether a
MOSFET is ON or OFF. If the voltage from UGATE or from
LGATE to GND is less than 0.8V, then the respective
MOSFET is defined as being OFF and the other MOSFET is
allowed to be turned ON. This method allows the VDDQ
regulator to both source and sink current.
Since the voltage of the MOSFET gates are being measured
to determine the state of the MOSFET, the designer is
encouraged to consider the repercussions of introducing
external components between the gate drivers and their
respective MOSFET gates before actually implementing
such measures. Doing so may interfere with the shootthrough protection.
VDDQ
VTT
Application Guidelines
500mV/DIV
Layout Considerations
Layout is very important in high frequency switching
converter design. With power devices switching efficiently at
250kHz, the resulting current transitions from one device to
another cause voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit,
and lead to device over-voltage stress. Careful component
layout and printed circuit board design minimizes these
voltage spikes.
INTERNAL DELAY
DELAY INTERVAL
T1
T0
T2
TIME
FIGURE 3. VTT/VDDQ LDO UNDER VOLTAGE PROTECTION
RESPONSES
If either rail exceeds 115% of the targeted voltage, then all
outputs are immediately disabled. The ISL6532 will not reenable the outputs until either the bias voltage is toggled in
order to initiate a POR or the SLP_S5 signal is forced LOW
and then back to HIGH.
Thermal Protection (S0/S3 State)
If the ISL6532 IC junction temperature reaches a nominal
temperature of 140oC, all regulators will be disabled. The
ISL6532 will not re-enable the outputs until the junction
temperature drops below 110oC and either the bias voltage
is toggled in order to initiate a POR or the SLP_S5 signal is
forced LOW and then back to HIGH.
9
As an example, consider the turn-off transition of the upper
MOSFET. Prior to turn-off, the MOSFET is carrying the full
load current. During turn-off, current stops flowing in the
MOSFET and is picked up by the lower MOSFET. Any
parasitic inductance in the switched current path generates a
large voltage spike during the switching interval. Careful
component selection, tight layout of the critical components,
and short, wide traces minimizes the magnitude of voltage
spikes.
There are two sets of critical components in the ISL6532
switching converter. The switching components are the most
critical because they switch large amounts of energy, and
therefore tend to generate large amounts of noise. Next are
the small signal components which connect to sensitive
nodes or supply critical bypass current and signal coupling.
A multi-layer printed circuit board is recommended. Figure 4
shows the connections of the critical components in the
converter. Note that capacitors CIN and COUT could each
ISL6532
represent numerous physical capacitors. Dedicate one solid
layer, usually a middle layer of the PC board, for a ground
plane and make all critical component ground connections
with vias to this layer. Dedicate another solid layer as a
power plane and break this plane into smaller islands of
common voltage levels. Keep the metal runs from the
PHASE terminals to the output inductor short. The power
plane should support the input power and output power
nodes. Use copper filled polygons on the top and bottom
circuit layers for the phase nodes. Use the remaining printed
circuit layers for small signal wiring. The wiring traces from
the GATE pins to the MOSFET gates should be kept short
and wide enough to easily handle the 1A of drive current.
12VATX
The switching components should be placed close to the
ISL6532 first. Minimize the length of the connections
between the input capacitors, CIN, and the power switches
by placing them nearby. Position both the ceramic and bulk
input capacitors as close to the upper MOSFET drain as
possible. Position the output inductor and output capacitors
between the upper and lower MOSFETs and the load.
The critical small signal components include any bypass
capacitors, feedback components, and compensation
components. Place the PWM converter compensation
components close to the FB and COMP pins. The feedback
resistors should be located as close as possible to the FB
pin with vias tied straight to the ground plane as required.
Feedback Compensation - PWM Buck Converter
CBP
P12V
Figure 5 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The
error amplifier output (VE/A) is compared with the oscillator
(OSC) triangular wave to provide a pulse-width modulated
(PWM) wave with an amplitude of VIN at the PHASE node.
The PWM wave is smoothed by the output filter (LO and CO).
VIN_DDR
GND
ISL6532
NCH
5VSBY
P5VSBY
5VSBY
GND
CIN
CBP
Q1
LGATE
COMP
Q2
COUT1
PWM
COMPARATOR
VDDQ
LOAD
UGATE
DRIVER
+
R1
FB
C3 R3
VDDQ
VTT(2)
VDDQ
CO
ESR
(PARASITIC)
VE/A
ZIN
-
+
REFERENCE
ERROR
AMP
VDDQ(2)
DETAILED COMPENSATION COMPONENTS
VTT
ZFB
C1
LOAD
COUT2
PHASE
ZFB
C1
R4
LO
-
ΔVOSC
C2
R2
VIN
DRIVER
OSC
LOUT
C2
VDDQ
ZIN
C3
R2
R3
GND PAD
R1
COMP
KEY
-
ISLAND ON POWER PLANE LAYER
+
ISLAND ON CIRCUIT PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 4. PRINTED CIRCUIT BOARD POWER PLANES
AND ISLANDS
In order to dissipate heat generated by the internal VTT
LDO, the ground pad, pin 21, should be connected to the
internal ground plane through at least four vias. This allows
the heat to move away from the IC and also ties the pad to
the ground plane through a low impedance path.
10
FB
R4
ISL6532
REFERENCE
R ⎞
⎛
V DDQ = 0.8 × ⎜ 1 + ------1-⎟
R 4⎠
⎝
FIGURE 5. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN AND OUTPUT
VOLTAGE SELECTION
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR . The DC Gain of
ISL6532
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage ΔVOSC .
100
1
F LC = ------------------------------------------2π x L O x C O
60
1
F ESR = -------------------------------------------2π x ESR x C O
The compensation network consists of the error amplifier
(internal to the ISL6532) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R1 , R2 ,
R3 , C1 , C2 , and C3) in Figure 5. Use these guidelines for
locating the poles and zeros of the compensation network:
1. Pick Gain (R2/R1) for desired converter bandwidth.
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).
3. Place 2ND Zero at Filter’s Double Pole.
4. Place 1ST Pole at the ESR Zero.
5. Place 2ND Pole at Half the Switching Frequency.
6. Check Gain against Error Amplifier’s Open-Loop Gain.
7. Estimate Phase Margin - Repeat if Necessary.
Compensation Break Frequency Equations
1
F Z1 = -----------------------------------2π x R 2 x C 2
1
F P1 = --------------------------------------------------------⎛ C 1 x C 2⎞
2π x R 2 x ⎜ ----------------------⎟
⎝ C1 + C2 ⎠
1
F Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3
1
F P2 = -----------------------------------2π x R 3 x C 3
Figure 6 shows an asymptotic plot of the DC-DC converter’s
gain vs. frequency. The actual Modulator Gain has a high
gain peak due to the high Q factor of the output filter and is
not shown in Figure 6. Using the above guidelines should
give a Compensation Gain similar to the curve plotted. The
open loop error amplifier gain bounds the compensation
gain. Check the compensation gain at FP2 with the
capabilities of the error amplifier. The Closed Loop Gain is
constructed on the graph of Figure 6 by adding the
Modulator Gain (in dB) to the Compensation Gain (in dB).
This is equivalent to multiplying the modulator transfer
function to the compensation transfer function and plotting
the gain.
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW)
overall loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
11
GAIN (dB)
Modulator Break Frequency Equations
80
40
20
FZ1 FZ2
FP1
FP2
OPEN LOOP
ERROR AMP GAIN
20LOG
(R2/R1)
20LOG
(VIN/ΔVOSC)
0
COMPENSATION
GAIN
MODULATOR
GAIN
-20
CLOSED LOOP
GAIN
-40
FLC
-60
10
100
1K
FESR
10K
100K
1M
10M
FREQUENCY (Hz)
FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Output Voltage Selection
The output voltage of the VDDQ PWM converter can be
programmed to any level between VIN and the internal
reference, 0.8V. An external resistor divider is used to scale
the output voltage relative to the reference voltage and feed
it back to the inverting input of the error amplifier, see
Figure 6.
However, since the value of R1 affects the values of the rest
of the compensation components, it is advisable to keep its
value less than 5kW. Depending on the value chosen for R1,
R4 can be calculated based on the following equation:
R1 × 0.8V
R4 = ----------------------------------V DDQ – 0.8V
If the output voltage desired is 0.8V, simply route VDDQ back
to the FB pin through R1, but do not populate R4.
The output voltage for the internal VTT linear regulator is set
internal to the ISL6532 to track the VDDQ voltage by 50%.
There is no need for external programming resistors.
Component Selection Guidelines
Output Capacitor Selection - PWM Buck Converter
An output capacitor is required to filter the inductor current
and supply the load transient current. The filtering
requirements are a function of the switching frequency and
the ripple current. The load transient requirements are a
function of the slew rate (di/dt) and the magnitude of the
transient load current. These requirements are generally met
with a mix of capacitors and careful layout.
DDR memory systems are capable of producing transient
load rates above 1A/ns. High frequency capacitors initially
supply the transient and slow the current load rate seen by the
bulk capacitors. The bulk filter capacitor values are generally
determined by the ESR (Effective Series Resistance) and
voltage rating requirements rather than actual capacitance
requirements.
ISL6532
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
Output Capacitor Selection - LDO Regulator
The output capacitors used in LDO regulators are used to
provide dynamic load current. The amount of capacitance
and type of capacitor should be chosen with this criteria in
mind.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
ΔI =
VIN - VOUT
Fs x L
x
VOUT
VIN
ΔVOUT = ΔI x ESR
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6532 will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
12
equations give the approximate response time interval for
application and removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
VOUT
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Input Capacitor Selection - PWM Buck Converter
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time the upper MOSFET
turns on. Place the small ceramic capacitors physically close
to the MOSFETs, between the drain of upper MOSFET and
the source of lower MOSFET.
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. Their voltage rating should be
at least 1.25 times greater than the maximum input voltage,
while a voltage rating of 1.5 times is a conservative
guideline. For worst cases, the RMS current rating
requirement for the input capacitor of a buck regulator is
approximately 1/2 the DC output load current.
The maximum RMS current required by the regulator may be
closely approximated through the following equation:
I RMS
MAX
=
V OUT ⎛
V IN – V OUT V OUT 2
2
1
-------------- × I OUT
+ ------ × ⎛ ----------------------------- × --------------⎞ ⎞
⎝
V IN
V IN ⎠ ⎠
12 ⎝ L × f sw
MAX
For a through hole design, several electrolytic capacitors
may be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge-current at
power-up. Some capacitor series available from reputable
manufacturers are surge current tested.
MOSFET Selection - PWM Buck Converter
The ISL6532 requires 2 N-Channel power MOSFETs for
switching power and a third MOSFET to block backfeed from
VDDQ to the Input in S3 Mode. These should be selected
based upon rDS(ON) , gate supply requirements, and thermal
management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed
between the two MOSFETs according to duty factor. The
switching losses seen when sourcing current will be different
ISL6532
from the switching losses seen when sinking current. When
sourcing current, the upper MOSFET realizes most of the
switching losses. The lower switch realizes most of the
switching losses when the converter is sinking current (see
the equations below). These equations assume linear
voltage-current transitions and do not adequately model
power loss due the reverse-recovery of the upper and lower
MOSFET’s body diode. The gate-charge losses are
dissipated in part by the ISL6532 and do not significantly heat
the MOSFETs. However, large gate-charge increases the
switching interval, tSW which increases the MOSFET
switching losses. Ensure that both MOSFETs are within their
maximum junction temperature at high ambient temperature
by calculating the temperature rise according to package
thermal-resistance specifications. A separate heatsink may be
necessary depending upon MOSFET power, package type,
ambient temperature and air flow.
5VSBY
PUPPER = Io2 x rDS(ON) x D
2
1
P LOWER = Io × r DS ( ON ) × ( 1 – D ) + --- ⋅ Io × V IN × t SW × f s
2
Where: D is the duty cycle = VOUT / VIN ,
tSW is the combined switch ON and OFF time, and
fs is the switching frequency.
ISL6532 Application Circuit
Figure 7 shows an application circuit utilizing the ISL6532.
Detailed information on the circuit, including a complete Billof-Materials and circuit board description, can be found in
Application Note AN1055.
SLP_S5#
S3
SLP_S3#
P12V
S5
P5VSBY
PGOOD
VCC5
R1
4.99kΩ
Q5
L1
2.1μH
NCH
C4,5
1μF
C26
0.1μF
VREF_OUT
C27
0.1μF
UGATE
C19
0.47μF
2.5V
ISL6532
Q2,4
VDDQ
VDDQ
VTT
VTT
C21
220μF
R4
1.74kΩ
FB
COMP
GND
VTTSNS
GND
+
C15
1000pF
C14
6.8nF
R3
19.1kΩ
C13
56nF
R6
825Ω
FIGURE 7. DDR SDRAM AND AGP VOLTAGE REGULATOR USING THE ISL6532
13
C1-3
2200μF
VDDQ
L2
2.1μH
LGATE
C20 +
220μF
+
Q1,3
VREF_IN
VDDQ
1.25V
Approximate Losses while Sinking current
C16
1μF
5VSBY
PGOOD
VTT
PLOWER = Io2 x rDS(ON) x (1 - D)
C17,18
1μF
R2
10.0kΩ
VREF
2
1
P UPPER = Io × r DS ( ON ) × D + --- ⋅ Io × V IN × t SW × f s
2
VCC12
+3.3V
VDDQ
Approximate Losses while Sourcing current
R5
22.6Ω
+
C6-8
1800μF
C9-12
22μF
ISL6532
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L20.6x6
20 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VJJB ISSUE C)
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
A3
b
0.28
D
0.33
9
0.40
5, 8
6.00 BSC
D1
D2
9
0.20 REF
-
5.75 BSC
3.55
3.70
9
3.85
7, 8
E
6.00 BSC
-
E1
5.75 BSC
9
E2
3.55
e
3.70
3.85
7, 8
0.80 BSC
-
k
0.25
-
-
-
L
0.35
0.60
0.75
8
L1
-
-
0.15
10
N
20
2
Nd
5
3
Ne
5
3
P
-
-
0.60
9
θ
-
-
12
9
Rev. 1 10/02
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
14
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