DATASHEET

Data Sheet
Mobile Microprocessor CORE Voltage
Regulator Multi-Phase Buck PWM
Controller
The ISL6223 multi-phase PWM control IC together with its
companion gate drivers, the HIP6601, HIP6602 or HIP6603
and Intersil MOSFETs provides a precision voltage
regulation system for advanced mobile microprocessors. A
single-stage regulator that directly converts a battery voltage
to a microprocessor core voltage can be designed when high
voltage drivers are employed. Multiphase power conversion
is a marked departure from earlier single phase converter
configurations previously employed to satisfy the ever
increasing current demands of modern microprocessors.
Multi-phase converters, by distributing the power and load
current, result in smaller and lower cost transistors with
fewer input and output capacitors. These reductions accrue
from the higher effective conversion frequency with higher
frequency ripple current due to the phase interleaving
process of this topology. For example, a two phase converter
operating at 350kHz will have a ripple frequency of 700kHz.
Moreover, greater converter bandwidth of this design results
in faster response to load transients.
Outstanding features of this controller IC include
programmable VID codes from the microprocessor that
range from 0.925V to 2.00V with a system accuracy of 1%.
Pull up currents on these VID pins eliminates the need for
external pull up resistors. In addition “droop” compensation,
used to reduce the overshoot or undershoot of the CORE
voltage, is easily programmed with a single resistor.
Another feature of this controller IC is the PGOOD monitor
circuit which is held low until the CORE voltage increases,
during its Soft-Start sequence, to 0.9V. Overvoltage, the
CORE voltage going above 2.35V, results in the converter
shutting down and turning the lower MOSFETs ON to clamp
and protect the microprocessor. Under voltage is also
detected and results in PGOOD low if the CORE voltage
falls below 0.9V. Overcurrent protection reduces the
regulator current to less than 25% of the programmed trip
value. An external capacitor connected to the DACOUT pin
and ground slows down the transition of the DAC output to
avoid triggering the overcurrent protection when the VID
code changes. These features provide monitoring and
protection for the microprocessor and power system.
FN9013.3
Features
• Mobile VID Compatible Multi-Phase Power Conversion
• Precision Channel Current Sharing
- Loss Less Current Sampling - Uses rDS(ON)
• Precision CORE Voltage Regulation
- 1% System Accuracy Over Temperature
• Microprocessor Voltage Identification Input
- 5-Bit VID Input
- 1.30V to 2.00V in 50mV Steps
- 0.925V to 1.275V in 25mV Steps
- Programmable “Droop” Voltage
• Fast Transient Recovery Time
• Over Current Protection
• High Ripple Frequency (Channel Frequency Times
Number of Channels, Two) . . . . . . . . . . .100kHz to 3MHz
• Pb-free available (RoHS compliant)
Ordering Information
PART
NUMBER
ISL6223CA
PART
MARKING
ISL 6223CA
ISL6223CAZA ISL6 223CAZ
(Note)
TEMP.
(°C)
PACKAGE
PKG.
DWG. #
0 to +70
20 Ld SSOP
M20.15
0 to +70
20 Ld SSOP
(Pb-free)
M20.15
*Add “-T” suffix for tape and reel. Please refer to TB347 for details on
reel specifications.
NOTE: These Intersil Pb-free plastic packaged products employ
special Pb-free material sets, molding compounds/die attach
materials, and 100% matte tin plate plus anneal (e3 termination
finish, which is RoHS compliant and compatible with both SnPb and
Pb-free soldering operations). Intersil Pb-free products are MSL
classified at Pb-free peak reflow temperatures that meet or exceed
the Pb-free requirements of IPC/JEDEC J STD-020.
Pinout
ISL6223 (SSOP)
TOP VIEW
VID4 1
20 VCC
VID3 2
19 PGOOD
VID2 3
18 NC
VID1 4
17 NC
VID0 5
16 ISEN1
COMP 6
15 PWM1
FB 7
14 PWM2
FS/DIS 8
13 ISEN2
GND 9
VSEN 10
1
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12 NC
11 DACOUT
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2003, 2008. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6223
Block Diagram
VCC
PGOOD
POWER-ON
RESET (POR)
UV
VSEN
0.9V
+
-
OV
LATCH
CLOCK AND
SAWTOOTH
GENERATOR
S
+
2.35V
THREE
STATE
OVP
FS/DIS
-
+
SOFTSTART
AND FAULT
LOGIC

+
PWM
-
-
PWM1
COMP
DACOUT
VID0
+
VID1
VID2
PWM

+
-
D/A
+
VID3
-
PWM2
E/A
-
VID4
CURRENT
CORRECTION
FB
I_TOT
OC
+
-

I_TRIP
+
ISEN1
+
ISEN2
GND
2
FN9013.3
July 9, 2008
ISL6223
Simplified Power System Diagram
VSEN
PWM 1
The DAC output. Connect a capacitor to this pin slows down
the transition of the DAC output that is also the reference
voltage to the error amplifier.
SYNCHRONOUS
RECTIFIED BUCK
CHANNEL
ISL6223
MICROPROCESSOR
PWM 2
VID
DACOUT (Pin 11)
SYNCHRONOUS
RECTIFIED BUCK
CHANNEL
NC (Pin 12, Pin 17, Pin 18)
No connection.
ISEN2 (Pin 13) and ISEN1 (Pin 16)
Current sense inputs from the individual converter channel’s
phase nodes.
PWM2 (Pin 14) and PWM1 (Pin 15)
Functional Pin Descriptions
PWM outputs for each driven channel in use. Connect these
pins to the PWM input of a HIP6601/2/3 driver.
VID4 1
20 VCC
PGOOD (Pin 19)
VID3 2
19 PGOOD
VID2 3
18 NC
VID1 4
17 NC
Power good. This pin provides an open-drain logic-high
signal when the microprocessor CORE voltage (VSEN pin)
is within specified limits and Soft-Start has timed out.
VID0 5
16 ISEN1
VCC (Pin 20)
COMP 6
15 PWM1
Bias supply. Connect this pin to a 5V supply.
FB 7
14 PWM2
FS/DIS 8
13 ISEN2
GND 9
12 NC
VSEN 10
11 DACOUT
VID4 (Pin 1), VID3 (Pin 2), VID2 (Pin 3), VID1 (Pin 4)
and VID0 (Pin 5)
Voltage Identification inputs from microprocessor. These pins
respond to TTL and 3.3V logic signals. The ISL6223 decodes
VID bits to establish the output voltage. See Table 1.
COMP (Pin 6)
Output of the internal error amplifier. Connect this pin to the
external feedback and compensation network.
FB (Pin 7)
Inverting input of the internal error amplifier.
FS/DIS (Pin 8)
Channel frequency, FSW, select and disable. A resistor from
this pin to ground sets the switching frequency of the
converter. Pulling this pin to ground disables the converter
and three states the PWM outputs. See Figure 11.
GND (Pin 9)
Bias and reference ground. All signals are referenced to this
pin.
VSEN (Pin 10)
Power good monitor input. Connect to the microprocessorCORE voltage.
3
FN9013.3
July 9, 2008
ISL6223
Typical Application - Two Phase Converter Using HIP6601 Gate Drivers
+12V
PVCC
VIN = +5V
BOOT
UGATE
+5V
PWM
COMP
FB
VCC
PHASE
DRIVER
HIP6601
VSEN
LGATE
GND
VCC
PGOOD
VID3
VID2
VID1
MAIN
CONTROL
ISL6223
PVCC
FS/DIS
GND
VIN = +5V
BOOT
UGATE
ISEN2
VID0
+VCORE
+12V
PWM2
VID4
VCC
PWM
PWM1
ISEN1
DACOUT
PHASE
DRIVER
HIP6601
LGATE
GND
Typical Application - Two Phase Converter Using a HIP6602 Gate Driver
+5V
BOOT1
+12V
FB
VSEN
COMP
VCC
VID4
ISEN1
PWM1
VID3
VID1
UGATE1
VCC
PGOOD
VID2
VIN = +12V
PHASE1
PWM1
MAIN
CONTROL
ISL6223
PWM2
VID0
FS/DIS
GND
DUAL
DRIVER
HIP6602
PVCC
BOOT2
+5V
VIN +12V
PWM2
UGATE2
ISEN2
PHASE2
DACOUT
LGATE2
GND
4
+VCORE
LGATE1
PGND
FN9013.3
July 9, 2008
ISL6223
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+7V
Input, Output, or I/O Voltage . . . . . . . . . . GND - 0.3V to VCC + 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 1
Thermal Resistance (Typical, Note 1)
JA (°C/W)
SSOP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . .
120
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile. . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V 5%
Ambient Temperature. . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTE:
1. JA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications
Operating Conditions: VCC = 5V, TA = 0°C to 70°C, Unless Otherwise Specified
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
-
10
15
mA
INPUT SUPPLY POWER
Input Supply Current
RT = 100k, Active and Disabled Maximum Limit
POR (Power-On Reset) Threshold
VCC Rising
4.25
4.38
4.5
V
VCC Falling
3.75
3.88
4.00
V
REFERENCE AND DAC
System Accuracy
Percent System Deviation from Programmed VID Codes
-1
-
1
%
DAC (VID0 - VID4) Input Low Voltage
DAC Programming Input Low Threshold Voltage
-
-
0.8
V
DAC (VID0 - VID4) Input High Voltage
DAC Programming Input High Threshold Voltage
2.0
-
-
V
VID Pull-Up
VIDx = 0V
5
12
30
µA
CHANNEL GENERATOR
Frequency, FSW
RT = 100k, 1%
245
275
305
kHz
Adjustment Range
See Figure 11
0.05
-
1.5
MHz
Disable Voltage
Maximum Voltage at FS/DIS to Disable Controller. IFS/DIS = 1mA
-
-
1.0
V
DC Gain
RL = 10k to GND
-
72
-
dB
Gain-Bandwidth Product
CL = 100pF, RL = 10k to GND
-
18
-
MHz
Slew Rate
CL = 100pF, Load = 400µA
-
5.3
-
V/µs
Maximum Output Voltage
RL = 10k to GND, Load = 400µA
3.6
4.1
-
V
Minimum Output Voltage
RL = 10k to GND, Load = -400µA
-
0.16
0.5
V
Full Scale Input Current
-
50
-
µA
Overcurrent Trip Level
60
75
90
µA
VSEN Rising
-
0.90
-
V
VSEN Falling
-
0.88
-
V
IPGOOD = 4mA
-
0.18
0.4
V
2.28
2.35
2.45
V
-
1.7
-
V
ERROR AMPLIFIER
ISEN
POWER GOOD MONITOR
Undervoltage Threshold
PGOOD Low Output Voltage
PROTECTION
Overvoltage Threshold
VSEN Rising
VSEN Falling After Overvoltage
5
FN9013.3
July 9, 2008
ISL6223
RIN
FB
VIN
ISL6223
ERROR
AMPLIFIER
COMPARATOR
CORRECTION
-
+
+
-
PWM
CIRCUIT
+

Q1
L01
PWM1
HIP6601
IL1
-
Q2
PHASE
PROGRAMMABLE
REFERENCE
DAC
+

CURRENT
SENSING
I AVERAGE
CURRENT
AVERAGING
VCORE
+

+
RISEN1
ISEN1
CURRENT
ISEN2
SENSING

CORRECTION
COUT
RISEN2
RLOAD
VIN
PHASE
COMPARATOR
+
-
Q3
PWM
CIRCUIT
L02
PWM2
HIP6601
IL2
Q4
FIGURE 1. SIMPLIFIED BLOCK DIAGRAM OF THE ISL6223 VOLTAGE AND CURRENT CONTROL LOOPS FOR A TWO POWER
CHANNEL REGULATOR
Operation
Figure 1 shows a simplified diagram of the voltage regulation
and current control loops. Both voltage and current feedback
are used to precisely regulate voltage and tightly control
output currents, IL1 and IL2 , of the two power channels. The
voltage loop comprises the Error Amplifier, Comparators,
gate drivers and output MOSFETs. The Error Amplifier is
essentially connected as a voltage follower that has an input,
the Programmable Reference DAC and an output that is the
CORE voltage.
Voltage Loop
Feedback from the CORE voltage is applied via resistor RIN
to the inverting input of the Error Amplifier. This signal can
drive the Error Amplifier output either high or low, depending
upon the CORE voltage. Low CORE voltage makes the
amplifier output move towards a higher output voltage level.
Amplifier output voltage is applied to the positive inputs of
the Comparators via the Correction summing networks. Outof-phase sawtooth signals are applied to the two
Comparators inverting inputs. Increasing Error Amplifier
voltage results in increased Comparator output duty cycle.
6
This increased duty cycle signal is passed through the PWM
CIRCUIT with no phase reversal and on to the HIP6601,
again with no phase reversal for gate drive to the upper
MOSFETs, Q1 and Q3. Increased duty cycle or ON time for
the MOSFET transistors results in increased output voltage
to compensate for the low output voltage sensed.
Current Loop
The current control loop works in a similar fashion to the
voltage control loop, but with current control information
applied individually to each channel’s Comparator. The
information used for this control is the voltage that is
developed across rDS(ON) of each lower MOSFET, Q2 and
Q4, when they are conducting. A single resistor converts and
scales the voltage across the MOSFETs to a current that is
applied to the Current Sensing circuit within the ISL6223.
Output from these sensing circuits is applied to the current
averaging circuit. Each PWM channel receives the
difference current signal from the summing circuit that
compares the average sensed current to the individual
channel current. When a power channel’s current is greater
than the average current, the signal applied via the summing
Correction circuit to the Comparator, reduces the output
FN9013.3
July 9, 2008
ISL6223
pulse width of the Comparator to compensate for the
detected “above average” current in that channel.
state condition that makes these outputs essentially open.
This state results in no gate drive to the output MOSFETs.
Droop Compensation
Once the VCC voltage reaches 4.375V (+125mV), a voltage
level to insure proper internal function, the PWM outputs are
enabled and the Soft-Start sequence is initiated. If for any
reason, the VCC voltage drops below 3.875V (+125mV). The
POR circuit shuts the converter down and again three states
the PWM outputs.
In addition to control of each power channel’s output current,
the average channel current is also used to provide CORE
voltage “droop” compensation. Average full channel current
is defined as 50A. By selecting an input resistor, RIN, the
amount of voltage droop required at full load current can be
programmed. The average current driven into the FB pin
results in a voltage increase across resistor RIN that is in the
direction to make the Error Amplifier “see” a higher voltage
at the inverting input, resulting in the Error Amplifier
adjusting the output voltage lower. The voltage developed
across RIN is equal to the “droop” voltage. See the “Current
Sensing and Balancing” section for more details.
Applications and Convertor Start-Up
Each PWM power channel’s current is regulated. This
enables the PWM channels to accurately share the load
current for enhanced reliability. The HIP6601, HIP6602 or
HIP6603 MOSFET driver interfaces with the ISL6223. For
more information, see the HIP6601, HIP6602 or HIP6603
data sheets.
The ISL6223 controls the two PWM power channels 180°
out of phase. Figure 2 shows the out of phase relationship
between the two PWM channels.
PWM 1
PWM 2
FIGURE 2. TWO PHASE PWM OUTPUT AT 500kHz
Power supply ripple frequency is determined by the channel
frequency, FSW, multiplied by the number of active channels.
For example, if the channel frequency is set to 250kHz, the
ripple frequency is 500kHz with two channels.
The IC monitors and precisely regulates the CORE voltage
of a microprocessor. After initial start-up, the controller also
provides protection for the load and the power supply. The
following section discusses these features.
Initialization
The ISL6223 operates from a 5V power supply. Many
functions are initiated by the rising supply voltage to the VCC
pin of the ISL6223. Oscillator, Sawtooth Generator, SoftStart and other functions are initialized during this interval.
These circuits are controlled by POR, Power-On Reset.
During this interval, the PWM outputs are driven to a three
7
Soft-Start
After the POR function is completed with VCC reaching
4.375V, the Soft-Start sequence is initiated. Soft-Start, by its
slow rise in CORE voltage from zero, avoids an overcurrent
condition by slowly charging the discharged output
capacitors. This voltage rise is initiated by an internal DAC
that slowly raises the reference voltage to the error amplifier
input. The voltage rise is controlled by the oscillator
frequency and the DAC within the ISL6223, therefore, the
output voltage is effectively regulated as it rises to the final
programmed CORE voltage value.
For the first 64 PWM switching cycles, the DAC output
remains inhibited and the PWM outputs remain three stated.
From the 65th cycle and for another, approximately 300
cycles the PWM output remains low, clamping the lower
output MOSFETs to ground, see Figure 3. The time
variability is due to the Error Amplifier, Sawtooth Generator
and Comparators moving into their active regions. After this
short interval, the PWM outputs are enabled and increment
the PWM pulse width from zero duty cycle to operational
pulse width, thus allowing the output voltage to slowly reach
the CORE voltage. The CORE voltage will reach its
programmed value before the 4096 cycles, but the PGOOD
output will not be initiated until the 4096th switching cycle.
The Soft-Start time or delay time, DT = 4096/FSW. For an
oscillator frequency, FSW, of 200kHz, the first 64 cycles or
320s, the PWM outputs are held in a three state level as
explained above. After this period and a short interval
described above, the PWM outputs are initiated and the
voltage rises in 20.16ms, for a total delay time DT of
20.48ms.
Figure 3 shows the start-up sequence as initiated by an
enable (EN) switch, applied to the ISL6223. The start-up is
enabled at the falling edge of the EN switch output.
Figure 4 shows the waveforms when the regulator is
operating at 200kHz. Note that the Soft-Start duration is a
function of the Channel Frequency as explained previously.
Also note the pulses on the COMP terminal. These pulses
are the current correction signal feeding into the comparator
input (see the “Block Diagram” on page 2).
Figure 5 shows the regulator operating from a 12V battery
supply. In this system, the battery voltage is available before
any other voltages, including the 5V bias voltage VCC for the
controller IC. In this figure, note the slight rise in PGOOD as
FN9013.3
July 9, 2008
ISL6223
the 5V supply rises. The PGOOD output stage is made up of
NMOS and PMOS transistors. On the rising VCC, the PMOS
device becomes active slightly before the NMOS transistor
pulls “down”, generating the slight rise in the PGOOD
voltage.
.
12V
SUPPLY
Note that Figure 5 shows the 12V battery voltage available
before the 5V supply to the ISL6223 has reached its
threshold level. If conditions were reversed and the 5V
supply was to rise first, the start-up sequence would be
different. In this case, the ISL6223 will sense an overcurrent
condition due to charging the output capacitors. The supply
will then restart and go through the normal Soft-Start cycle.
DELAY TIME
PGOOD
VCORE
5V
SUPPLY
FIGURE 5. SUPPLY POWERED BY ATX SUPPLY
PWM 1
OUTPUT
PGOOD
DELAY TIME
VCORE
Fault Protection
The ISL6223 protects the microprocessor and the entire
power system from damaging stress levels. Within the
ISL6223 both Overvoltage and Overcurrent circuits are
incorporated to protect the load and regulator.
Overvoltage
The VSEN pin is connected to the microprocessor CORE
voltage. A CORE overvoltage condition is detected when the
VSEN pin goes above 2.35V.
EN
SWITCH
VIN = 12V
FIGURE 3. START-UP OF A SYSTEM OPERATING AT 200kHz
V COMP
PGOOD
DELAY TIME
VCORE
EN
SWITCH
VIN = 12V
FIGURE 4. START-UP A SYSTEM OPERATING AT 200kHz
8
The overvoltage condition is latched, disabling normal PWM
operation, and causing PGOOD to go low. The latch can
only be reset by lowering and returning VCC high to initiate a
POR and Soft-Start sequence.
During a latched overvoltage, the PWM outputs will be
driven either low or three state, depending upon the VSEN
input. PWM outputs are driven low when the VSEN pin
detects that the CORE voltage is above 2.35V. This
condition drives the PWM outputs low, resulting in the lower
or synchronous rectifier MOSFETs to conduct and shunt the
CORE voltage to ground to protect the load.
If after this event, the CORE voltage falls below 1.7V, the
PWM outputs will be three state. The HIP6601 family of
drivers pass the three state information along, and shut off
both upper and lower MOSFETs. This prevents “dumping” of
the output capacitors back through the lower MOSFETs,
avoiding a possibly destructive ringing of the capacitors and
output inductors. If the conditions that caused the
overvoltage still persist, the PWM outputs will be cycled
between three state and VCORE clamped to ground, as a
hysteretic shunt regulator.
Undervoltage
The VSEN pin also detects when the CORE voltage falls
below 0.9V level. This causes PGOOD to go low, but has no
other effect on operation and is not latched. There is also
hysteresis in this detection point.
FN9013.3
July 9, 2008
ISL6223
Overcurrent
DACOUT Pin
In the event of an overcurrent condition, the overcurrent
protection circuit reduces the average current delivered to
less than 25% of the current limit. When an overcurrent
condition is detected, the controller forces all PWM outputs
into a three state mode. This condition results in the gate
driver removing drive to the output stages. The ISL6223
goes into a wait delay timing cycle that is equal to the
Soft-Start ramp time. PGOOD also goes “low” during this
time due to VSEN going below its threshold voltage. To
lower the average output dissipation, the Soft-Start initial
wait time is increased from 64 to 4096 cycles, then the SoftStart ramp is initiated. At a PWM frequency of 200kHz, for
instance, an overcurrent detection would cause a dead time
of 20.48ms, then a ramp of 20.16ms.
The internal DAC output is brought out to pin 11, DACOUT,
in ISL6223. The typical output impedance of the DAC is
1.7k. The DACOUT pin allows the user to connect a
capacitor between this pin and the ground to form an RC
filter to slow down the voltage transition at the non-inverting
input of the error amplifier, when the VID code is being
changed. Slower voltage transition reduces the inrush
current to avoid tripping the overcurrent protection during the
transition. Typical systems require the transition to be
finished within 100µs, therefore, a time constant of 30µs to
40µs is a good tradeoff between the inrush current and the
transition time. Connecting a 22nF capacitor to the DACOUT
results in a time constant of 37µs for the RC filter. Figure 7
shows the waveforms for the VID changes from 1.3V to 1.6V.
From top to bottom, the waveforms are the core voltage, the
DACOUT, and the two inductor currents.
At the end of the delay, PWM outputs are restarted and the
Soft-Start ramp is initiated. If a short is present at that time,
the cycle is repeated. This is the hiccup mode.
Figure 6 shows the supply shorted under operation and the
hiccup operating mode described above. Note that due to
the high short circuit current, overcurrent is detected before
completion of the start-up sequence so the delay is not quite
as long as the normal Soft-Start cycle.
VOLTAGE IDENTIFICATION CODE AT
PROCESSOR PINS
VID4
VID3
VID2
VID1
VID0
VCCCORE
(VDC)
1
1
1
1
1
Shutdown
1
1
1
1
0
0.925
1
1
1
0
1
0.950
1
1
1
0
0
0.975
1
1
0
1
1
1.000
1
1
0
1
0
1.025
1
1
0
0
1
1.050
SHORT
1
1
0
0
0
1.075
CURRENT
1
0
1
1
1
1.100
50A/DIV.
1
0
1
1
0
1.125
1
0
1
0
1
1.150
1
0
1
0
0
1.175
1
0
0
1
1
1.200
1
0
0
1
0
1.225
1
0
0
0
1
1.250
1
0
0
0
0
1.275
0
1
1
1
1
Shutdown
0
1
1
1
0
1.300
0
1
1
0
1
1.350
0
1
1
0
0
1.400
0
1
0
1
1
1.450
0
1
0
1
0
1.500
0
1
0
0
1
1.550
0
1
0
0
0
1.600
0
0
1
1
1
1.650
0
0
1
1
0
1.700
0
0
1
0
1
1.750
0
0
1
0
0
1.800
0
0
0
1
1
1.850
0
0
0
1
0
1.900
0
0
0
0
1
1.950
0
0
0
0
0
2.000
SHORT APPLIED HERE
PGOOD
FIGURE 6. SHORT APPLIED TO SUPPLY AFTER POWER-UP
VCORE
DACOUT
INDUCTOR
CURRENTS
FIGURE 7. VID CHANGES FROM 1.3V TO 1.6V. THE LOAD
CURRENT IS SET TO 10A.
9
TABLE 1. VOLTAGE IDENTIFICATION CODES
FN9013.3
July 9, 2008
ISL6223
RIN
Cc
RFB
COMP
FB
VIN
ISL6223
GENERATOR
-
+
CORRECTION
+
-
L01
Q1
PWM
CIRCUIT
PWM
HIP6601
IL
+
Q2
REFERENCE
DAC
VCORE
RLOAD
ERROR
AMPLIFIER
COMPARATOR
COUT
SAWTOOTH
PHASE
DIFFERENCE
+
ISEN
CURRENT
RISEN
SENSING
TO OTHER
CHANNEL
AVERAGING
TO OVER
CURRENT
TRIP
+
-
REFERENCE
COMPARATOR
CURRENT
SENSING
FROM
OTHER
CHANNEL
ONLY ONE OUTPUT
STAGE SHOWN
INDUCTOR
CURRENT
FROM
OTHER
CHANNEL
FIGURE 8. SIMPLIFIED FUNCTIONAL BLOCK DIAGRAM SHOWING CURRENT AND VOLTAGE SAMPLING
CORE Voltage Programming
The voltage identification pins (VID0, VID1, VID2, VID3 and
VID4) set the CORE output voltage. Each VID pin is pulled to
VCC by an internal 12µA current source and accepts
open-collector/open-drain/open-switch-to-ground or
standard low-voltage TTL or CMOS signals.
Table 1 shows the nominal DAC voltage as a function of the
VID codes. The power supply system is 1% accurate over
the operating temperature and voltage range.
Current Sensing and Balancing
Overview
The ISL6223 samples the on-state voltage drop across each
synchronous rectifier FET, Q2, as an indication of the
inductor current in that phase, see Figure 8. Neglecting AC
effects (to be discussed later), the voltage drop across Q2 is
simply rDS(ON)(Q2) x inductor current (IL). Note that IL, the
inductor current, is 1/2 of the total current (ILT).
The voltage at Q2’s drain, the PHASE node, is applied to the
RISEN resistor to develop the IISEN current to the ISL6223
ISEN pin. This pin is held at virtual ground, so the current
through RISEN is IL x rDS(ON)(Q2) / RISEN.
The IISEN current provides information to perform the
following functions:
1. Detection of an overcurrent condition
10
2. Reduce the regulator output voltage with increasing load
current (droop)
3. Balance the IL currents in the two phases
Overcurrent, Selecting RISEN
The current detected through the RISEN resistor is averaged
with the current detected in the other channel. The averaged
current is compared with a trimmed, internally generated
current, and used to detect an overcurrent condition.
The nominal current through the RISEN resistor should be
50A at full output load current, and the nominal trip point for
overcurrent detection is 150% of that value, or 75µA.
Therefore:
R ISEN = I L  r DS  ON   Q2   50A
(EQ. 1)
For a full load of 25A per phase, and an rDS(ON) (Q2) of
4m, RISEN = 2k.
The overcurrent trip point would be 150% of 25A, or
approximately 37.5A per phase. The RISEN value can be
adjusted to change the overcurrent trip point, but it is
suggested to stay within 25% of nominal.
Droop, Selection of RIN
The average of the currents detected through the RISEN
resistors is also steered to the FB pin. There is no DC return
path connected to the FB pin except for RIN, so the average
current creates a voltage drop across RIN. This drop increases
the apparent VCORE voltage with increasing load current,
causing the system to decrease VCORE to maintain balance at
FN9013.3
July 9, 2008
ISL6223
With a high dv/dt load transient, typical of high performance
microprocessors, the largest deviations in output voltage
occur at the leading and trailing edges of the load transient. In
order to fully utilize the output-voltage tolerance range, the
output voltage is positioned in the upper half of the range
when the output is unloaded and in the lower half of the range
when the controller is under full load. This droop
compensation allows larger transient voltage deviations and
thus reduces the size and cost of the output filter components.
RIN should be selected to give the desired “droop” voltage at
the normal full load current 50µA applied through the RISEN
resistor (or at a different full load current if adjusted as under
“Overcurrent, Selecting RISEN” above).
R IN = V DROOP  50A
Example:
If VCORE = 1.6V, VIN = 12V, L = 1.3µH and FSW = 250kHz;
then iP-P = 4.3A.
25
20
AMPERES
the FB pin. This is the desired “droop” voltage used to maintain
VCORE within limits under transient conditions.
15
10
5
0
(EQ. 2)
For a VDROOP of 80mV, RIN = 1.6k
The AC feedback components, RFB and Cc, are scaled in
relation to RIN.
FIGURE 9. TWO CHANNEL MULTIPHASE SYSTEM WITH CURRENT
BALANCING DISABLED
Current Balancing
The detected currents are also used to balance the phase
currents.
The balancing circuit can not make up for a difference in
rDS(ON) between synchronous rectifiers. If a FET has a
higher rDS(ON), the current through that phase will be
reduced.
25
20
AMPERES
Each phase’s current is compared to the average of the two
phase currents, and the difference is used to create an offset
in that phase’s PWM comparator. The offset is in a direction
to reduce the imbalance.
15
10
5
0
Figures 9 and 10 show the inductor current of a two phase
system without and with current balancing.
Inductor Current
The inductor current in each phase of a multi-phase Buck
converter has two components. There is a current equal to
the load current divided by the number of phases (ILT / n),
and a sawtooth current (iPK-PK), resulting from switching.
The sawtooth component is dependent on the size of the
inductors, the switching frequency of each phase, and the
values of the input and output voltage. Ignoring secondary
effects, such as series resistance, the peak to peak value of
the sawtooth current can be described by Equation 3:
i P-P =  V IN  V CORE – V CORE 2    L  F SW  V IN 
Where:
VCORE = DC value of the output or VID voltage
VIN = DC value of the input or supply voltage
L = value of the inductor
FSW = switching frequency
11
(EQ. 3)
FIGURE 10. TWO CHANNEL MULTIPHASE SYSTEM WITH CURRENT
BALANCING ENABLED
The inductor, or load current, flows alternately from VIN
through Q1 and from ground through Q2. The ISL6223
samples the on-state voltage drop across each Q2 transistor
to indicate the inductor current in that phase. The voltage
drop is sampled 1/3 of a switching period, 1/FSW, after Q1 is
turned OFF and Q2 is turned on. Because of the sawtooth
current component, the sampled current is different from the
average current per phase. Neglecting secondary effects,
the sampled current (ISAMPLE) can be related to the load
current (ILT) by Equation 4:
I SAMPLE = I LT  n +  V IN  V CORE – 3V CORE 2    6L  F SW  V IN 
(EQ. 4)
FN9013.3
July 9, 2008
ISL6223
Where:
ILT = total load current
n = the number of channels
Example: Using the previously given conditions, if ILT = 50A
and n = 2; then ISAMPLE = 25.49A.
As discussed previously, the voltage drop across each Q2
transistor at the point in time when current is sampled is
rDS(ON) (Q2) x ISAMPLE. The voltage at Q2’s drain, the
PHASE node, is applied through the RISEN resistor to the
ISL6223 ISEN pin. This pin is held at virtual ground, so the
current into ISEN is:
I SENSE = I SAMPLE  r DS  ON   Q2   R ISEN
(EQ. 5)
R ISEN = I SAMPLE  r DS  ON   Q2    50A 
(EQ. 6)
Example: From the previous conditions, if ILT = 50A,
ISAMPLE = 25.49A and rDS(ON) (Q2) = 4mthen
RISEN = 2.04k, ICURRENT TRIP = 150%, and Short circuit
ILT = 75A.
1,000
There are two sets of critical components in a DC/DC
converter using a ISL6223 controller and a HIP6601 gate
driver. The power components are the most critical because
they switch large amounts of energy. Next are small signal
components that connect to sensitive nodes or supply critical
bypassing current and signal coupling.
The power components should be placed first. Locate the
input capacitors close to the power switches. Minimize the
length of the connections between the input capacitors, CIN,
and the power switches. Locate the output inductors and
output capacitors between the MOSFETs and the load.
Locate the gate driver close to the MOSFETs.
500
200
100
RT (k)
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit
and lead to device overvoltage stress. Careful component
layout and printed circuit design minimizes the voltage
spikes in the converter. Consider, as an example, the turnoff
transition of the upper PWM MOSFET. Prior to turnoff, the
upper MOSFET was carrying channel current. During the
turnoff, current stops flowing in the upper MOSFET and is
picked up by the lower MOSFET. Any inductance in the
switched current path generates a large voltage spike during
the switching interval. Careful component selection, tight
layout of the critical components, and short, wide circuit
traces minimize the magnitude of voltage spikes. Contact
Intersil for evaluation board drawings of the component
placement and printed circuit board.
50
20
10
5
2
1
10
20
50 100 200
500 1,000 2,000 5,000 10,000
CHANNEL OSCILLATOR FREQUENCY, FSW (kHz)
FIGURE 11. RESISTANCE RT vs FREQUENCY
Channel Frequency Oscillator
The channel oscillator frequency is set by placing a resistor,
RT, to ground from the FS/DIS pin. Figure 11 is a curve
showing the relationship between frequency, FSW, and
resistor RT. To avoid pickup by the FS/DIS pin, it is important
to place this resistor next to the pin. If this pin is also used to
disable the converter, it is also important to locate the
pull-down device next to this pin.
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting
12
The critical small components include the bypass capacitors
for VCC and PVCC on the gate driver ICs. Locate the
bypass capacitor, CBP, for the ISL6223 controller close to
the device. It is especially important to locate the resistors
associated with the input to the amplifiers close to their
respective pins, since they represent the input to feedback
amplifiers. Resistor RT, that sets the oscillator frequency
should also be located next to the associated pin. It is
especially important to place the RSEN resistor(s) at the
respective terminals of the ISL6223.
A multi-layer printed circuit board is recommended. Figure 12
shows the connections of the critical components for one
output channel of the converter. Note that capacitors CIN and
COUT could each represent numerous physical capacitors.
Dedicate one solid layer, usually the middle layer of the PC
board, for a ground plane and make all critical component
ground connections with vias to this layer. Dedicate another
solid layer as a power plane and break this plane into smaller
islands of common voltage levels. Keep the metal runs from
the PHASE terminal to inductor LO1 short. The power plane
should support the input power and output power nodes. Use
copper filled polygons on the top and bottom circuit layers for
the phase nodes. Use the remaining printed circuit layers for
small signal wiring. The wiring traces from the driver IC to the
MOSFET gate and source should be sized to carry at least
one ampere of current.
FN9013.3
July 9, 2008
ISL6223
+5VIN
USE INDIVIDUAL METAL RUNS
FOR EACH CHANNEL TO HELP
ISOLATE OUTPUT STAGES
+12V
CBP
LOCATE NEXT TO IC PIN(S)
VCC
CBP
PWM
CBOOT
LOCATE NEXT
TO FB PIN
LOCATE NEAR TRANSISTOR
LO1
VCORE
PHASE
COUT
RT
ISL6223
RFB
CIN
HIP6601
COMP FS/DIS
CT
FB
LOCATE NEXT TO IC PIN
RSEN
RIN
VSEN
KEY
VCC PVCC
ISEN
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 12. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
The output capacitor is selected to meet both the dynamic
load requirements and the voltage ripple requirements. The
load transient for the microprocessor CORE is characterized
by high slew rate (di/dt) current demands. In general,
multiple high quality capacitors of different size and dielectric
are paralleled to meet the design constraints.
Modern microprocessors produce severe transient load rates.
High frequency capacitors supply the initially transient current
and slow the load rate-of-change seen by the bulk capacitors.
The bulk filter capacitor values are generally determined by
the ESR (effective series resistance) and voltage rating
requirements rather than actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR determines the output ripple voltage
and the initial voltage drop following a high slew-rate
transient’s edge. In most cases, multiple capacitors of small
case size perform better than a single large case capacitor.
Bulk capacitor choices include aluminum electrolytic, OS-Con,
Tantalum and even ceramic dielectrics. An aluminum
electrolytic capacitor’s ESR value is related to the case size
with lower ESR available in larger case sizes. However, the
equivalent series inductance (ESL) of these capacitors
increases with case size and can reduce the usefulness of the
capacitor to high slew-rate transient loading. Unfortunately,
13
Output Inductor Selection
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Small inductors in a multi-phase converter reduces
the response time without significant increases in total ripple
current.
The output inductor of each power channel controls the
ripple current. The control IC is stable for channel ripple
current (peak-to-peak) up to twice the average current. A
single channel’s ripple current is approximately:
V IN – V OUT V OUT
I = --------------------------------  ---------------F SW xL
V IN
(EQ. 7)
1.0
SINGLE
CHANNEL
0.8
VO / (LX FSW)
Output Capacitor Selection
ESL is not a specified parameter. Consult the capacitor
manufacturer and measure the capacitor’s impedance with
frequency to select a suitable component. For surface mount
designs, solid tantalum capacitors or Panasonic Speciality
Polymer (SP) capacitors can be used.
RIPPLE CURRENT (APEAK-PEAK)
Component Selection Guidelines
0.6
2 CHANNEL
0.4
3 CHANNEL
0.2
4 CHANNEL
0
0
0.1
0.2
0.3
0.4
0.5
DUTY CYCLE (VO/VIN)
FIGURE 13. RIPPLE CURRENT vs DUTY CYCLE
FN9013.3
July 9, 2008
ISL6223
For bulk capacitance, several electrolytic capacitors (Panasonic
HFQ series or Nichicon PL series or Sanyo MV-GX or
equivalent) may be needed. For surface mount designs, solid
tantalum capacitors or Panasonic Speciality Polymer (SP)
capacitors can be used. Caution must be exercised with regard
to the capacitor surge current rating when using the Tantalum
capacitors. These capacitors must be capable of handling the
surge-current at power-up. The TPS series available from AVX,
and the 593D series from Sprague are both surge current
tested.
CURRENT MULTIPLIER
0.5
SINGLE
CHANNEL
0.4
0.3
2 CHANNEL
0.2
3 CHANNEL
0.1
0
4 CHANNEL
0
0.1
MOSFET Selection and Considerations
0.2
0.3
0.4
0.5
DUTY CYCLE (VO/VIN)
FIGURE 14. CURRENT MULTIPLIER vs DUTY CYCLE
The current from multiple channels tend to cancel each other
and reduce the total ripple current. Figure 13 gives the total
ripple current as a function of duty cycle, normalized to the
parameter  Vo    L  F S  at zero duty cycle. To determine the
total ripple current from the number of channels and the duty
cycle, multiply the y-axis value by  Vo    LxF SW  .
Small values of output inductance can cause excessive power
dissipation. The ISL6223 is designed for stable operation for
ripple currents up to twice the load current. However, for this
condition, the RMS current is 115% above the value shown in
the following MOSFET Selection and Considerations section.
With all else fixed, decreasing the inductance could increase
the power dissipated in the MOSFETs by 30%.
Input Capacitor Selection
The important parameters for the bulk input capacitors are
the voltage rating and the RMS current rating. For reliable
operation, select bulk input capacitors with voltage and
current ratings above the maximum input voltage and
largest RMS current required by the circuit. The capacitor
voltage rating should be at least 1.25 times greater than the
maximum input voltage and a voltage rating of 1.5 times is
a conservative guideline. The RMS current required for a
multi-phase converter can be approximated with the aid of
Figure 14.
First determine the operating duty ratio as the ratio of the
output voltage divided by the input voltage. Find the Current
Multiplier from the curve with the appropriate power
channels. Multiply the current multiplier by the full load
output current. The resulting value is the RMS current rating
required by the input capacitor.
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use ceramic capacitance
for the high frequency decoupling and bulk capacitors to
supply the RMS current. Small ceramic capacitors should
be placed very close to the drain of the upper MOSFET to
suppress the voltage induced in the parasitic circuit
impedances.
14
In high-current PWM applications, the MOSFET power
dissipation, package selection and heatsink are the
dominant design factors. The power dissipation includes two
loss components; conduction loss and switching loss. These
losses are distributed between the upper and lower
MOSFETs according to duty factor (see Equations 8 and 9).
The conduction losses are the main component of power
dissipation for the lower MOSFETs, Q2 and Q4 of Figure 1.
Only the upper MOSFETs, Q1 and Q3 have significant
switching losses, since the lower device turns on and off into
near zero voltage.
The equations assume linear voltage-current transitions and
do not model power loss due to the reverse-recovery of the
lower MOSFETs body diode. The reverse-recovery loss can
be a significant portion of the upper MOSFETs. The gatecharge losses are dissipated by the Driver IC and don't heat
the MOSFETs. However, large gate-charge increases the
switching time, tSW which increases the upper MOSFET
switching losses. Ensure that both MOSFETs are within their
maximum junction temperature at high ambient temperature
by calculating the temperature rise according to package
thermal-resistance specifications. A separate heatsink may
be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
2
I O  r DS  ON   V OUT I O  V IN  t SW  F SW
P UPPER = ------------------------------------------------------------ + ---------------------------------------------------------V IN
2
(EQ. 8)
2
I O  r DS  ON    V IN – V OUT 
P LOWER = --------------------------------------------------------------------------------V IN
(EQ. 9)
A diode, anode to ground, may be placed across Q2 and Q4
of Figure 1. These diodes function as a clamp that catches
the negative inductor swing during the dead time between
the turn off of the lower MOSFETs and the turn on of the
upper MOSFETs. The diodes must be a Schottky type to
prevent the lossy parasitic MOSFET body diode from
conducting. It is usually acceptable to omit the diodes and let
the body diodes of the lower MOSFETs clamp the negative
inductor swing, but efficiency could drop one or two percent
as a result. The diode's rated reverse breakdown voltage
must be greater than the maximum input voltage.
FN9013.3
July 9, 2008
ISL6223
15
FN9013.3
July 9, 2008
ISL6223
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
N
INDEX
AREA
H
0.25(0.010) M
2
GAUGE
PLANE
INCHES
SYMBOL
3
0.25
0.010
SEATING PLANE
-A-
20 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
E
-B-
1
M20.15
B M
A
D
L
h x 45°
-C-

e
A2
A1
B
C
0.10(0.004)
0.17(0.007) M
C A M
B S
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm
(0.006 inch) per side.
MIN
MILLIMETERS
MAX
MIN
MAX
NOTES
A
0.053
0.069
1.35
1.75
-
A1
0.004
0.010
0.10
0.25
-
A2
-
0.061
-
1.54
-
B
0.008
0.012
0.20
0.30
9
C
0.007
0.010
0.18
0.25
-
D
0.337
0.344
8.56
8.74
3
E
0.150
0.157
3.81
3.98
4
e
0.025 BSC
0.635 BSC
-
H
0.228
0.244
5.80
6.19
-
h
0.0099
0.0196
0.26
0.49
5
L
0.016
0.050
0.41
1.27
6
N

20
0°
20
8°
0°
7
8°
Rev. 1 6/04
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per
side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of “B” dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9001 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
16
FN9013.3
July 9, 2008
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