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T
88-IN
1-8
Data
Sheet
July 2004
®
ACPI Regulator/Controller for
Dual Channel DDR Memory Systems
The switching PWM controller drives two N-Channel
MOSFETs in a synchronous-rectified buck converter
topology. The synchronous buck converter uses voltagemode control with fast transient response. Both the switching
regulator and standby LDO provide a maximum static
regulation tolerance of 2% over line, load, and temperature
ranges. The output is user-adjustable by means of external
resistors down to 0.8V.
Switching memory core output between the PWM regulator
and the standby LDO during state transitions is
accomplished smoothly via the internal ACPI control
circuitry. The NCH signal provides synchronized switching of
a backfeed blocking switch during the transitions eliminating
the need to route 5V Dual to the memory supply.
An integrated soft-start feature brings all outputs into
regulation in a controlled manner when returning to S0/S1
state from any sleep state. During S0 the PGOOD signal
indicates VTT is within spec and operational.
Each output is monitored for under and over-voltage events.
The switching regulator has over current protection. Thermal
shutdown is integrated.
• Generates 3 Regulated Voltages
- Synchronous Buck PWM Controller with Standby LDO
- 3A Integrated Sink/Source Linear Regulator with
Accurate VDDQ/2 Divider Reference.
- Glitch-free Transitions During State Changes
- LDO Regulator for 1.5V Video and Core voltage
• ACPI compliant sleep state control
• Integrated VREF Buffer
• PWM Controller Drives Low Cost N-Channel MOSFETs
• 250kHz Constant Frequency Operation
• Tight Output Voltage Regulation
- All Outputs: 2% Over Temperature
• 5V or 3.3V Down Conversion
• Fully-Adjustable Outputs with Wide Voltage Range: Down
to 0.8V supports DDR and DDR2 Specifications
• Simple Single-Loop Voltage-Mode PWM Control Design
• Fast PWM Converter Transient Response
• Under and Over-voltage Monitoring on All Outputs
• OCP on the Switching Regulator and VTT
• Integrated Thermal Shutdown Protection
• QFN Package Option
- QFN Compliant to JEDEC PUB95 MO-220 QFN - Quad
Flat No Leads - Product Outline
- QFN Near Chip Scale Package Footprint; Improves
PCB Efficiency, Thinner in Profile
• Pb-free available
Applications
• Single and Dual Channel DDR Memory Power Systems in
ACPI compliant PCs
• Graphics cards - GPU and memory supplies
• ASIC power supplies
NCH
S3#
S5#
P12V
UGATE
LGATE
GNDP
ISL6532C (QFN) TOP VIEW
FN9121.2
Features
The ISL6532C provides a complete ACPI compliant power
solution for up to 4 DIMM dual channel DDR/DDR2 Memory
systems. Included are both a synchronous buck controller
and integrated LDO to supply VDDQ with high current during
S0/S1 states and standby current during S3 state. During
S0/S1 state, a fully integrated sink-source regulator
generates an accurate (VDDQ/2) high current VTT voltage
without the need for a negative supply. A buffered version of
the VDDQ/2 reference is provided as VREF. An LDO
controller is also integrated for AGP core voltage regulation.
Pinout
ISL6532C
• Embedded processor and I/O supplies
28 27 26 25 24 23 22
• DSP supplies
GNDP 1
21 PGOOD
5VSBY 2
20 PHASE
GNDQ 3
19 DRIVE2
GNDQ 4
18 FB2
PART NUMBER
VTT 5
17 GNDA
ISL6532CCR
0 to 70
28 Ld 6x6 QFN L28.6x6
VTT 6
16 COMP
ISL6532CCRZ
(See Note)
0 to 70
28 Ld 6x6 QFN L28.6x6
(Pb-free)
VDDQ 7
15 FB
1
VREF_IN
VREF_OUT
OCSET
10 11 12 13 14
P5VSBY
VDDQ
9
VTTSNS
VDDQ
8
Ordering Information
TEMP. RANGE
(oC)
PACKAGE
PKG. DWG. #
*Add “-T” suffix to part number for tape and reel packaging.
NOTE: Intersil Pb-free products employ special Pb-free material sets; molding
compounds/die attach materials and 100% matte tin plate termination finish, which
is compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that meet or
exceed the Pb-free requirements of IPC/JEDEC J Std-020B.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2002-2004. All Rights Reserved.
All other trademarks mentioned are the property of their respective owners.
Block Diagram
P5VSBY
S3#
VDDQ S3
REGULATOR
+
-
S5#
5VSBY
VOLTAGE
REFERENCE
0.800V
0.680V (-15%)
VDDQ(3)
5V
0.920V (+15%)
12VCC
POR
2
VTTSNS
+
VTT
- REG
+
VTT(2)
S3
GNDQ
S0
DISABLE
+
-
VREF_IN
{
RL
+
FB2
UV/OV3
NCH
UV/OV
PWM ENABLE
12V
POR
SOFT-START
RU
OSCILLATOR
+
PWM
P12V
PWM
LOGIC
UGATE
250kHz
+
+
-
VREF_OUT
+
COMP
EA1
-
GNDA
DRIVE2
650 OUTPUT
IMPEDANCE
UV/OV1
PHASE
+
-
OC
COMP
20A
LGATE
UV/OV2
PGOOD
FB
COMP
OCSET
GNDP
ISL6532C
{
S0/S3
SLEEP,
SOFT-START,
PGOOD,
AND FAULT
LOGIC
EA2
ISL6532C
Simplified Power System Diagram
12V
5VSBY
5V
ISL6532C
NCH
SLEEP
STATE
LOGIC
SLP_S3
SLP_S5
Q1
VDDQ
PWM
CONTROLLER
+
Q2
5VSBY/3V3SBY
STANDBY
LDO
VDDQ
VREF
LINEAR
CONTROLLER
Q3
VTT
REGULATOR
VAGP
VTT
+
+
Typical Application - 5V or 3.3V Input
5VSBY
+12V
+3.3V
+5V or +3.3V
P12V
P5VSBY
5VSBY
CBP
RNCH
PGOOD
VDDQ
S3#
SLP_S3
NCH
S5#
SLP_S5
VREF_OUT
VREF
ROCSET
UGATE
+
ISL6532C
LGATE
VTT
VDDQ
VDDQ
VDDQ
VTTSNS
GNDQ
GNDQ
+
CVTT_OUT
DRIVE2
Q3
FB
COMP
VAGP
1.5V
FB2
+
GNDP
COUT2
3
GNDA
CIN
Q1
PHASE
VTT
VDDQ
+
OCSET
VREF_IN
VTT
Q4
VDDQ
LOUT
2.5V
+
Q2
CVDDQ_OUT
ISL6532C
Typical Application - Input From 5V Dual
5VSBY
+12V
5V Dual
P12V
P5VSBY
CBP
5VSBY
+3.3V
PGOOD
VDDQ
S3#
SLP_S3
NCH
S5#
SLP_S5
VREF_OUT
VREF
+
OCSET
ROCSET
VREF_IN
UGATE
Q1
PHASE
VTT
VTT
ISL6532C
LGATE
VTT
+
VDDQ
VDDQ
VDDQ
CVTT_OUT
VDDQ
GNDQ
GNDQ
VTTSNS
DRIVE2
Q3
FB
VAGP
COMP
1.5V
FB2
+
GNDP
COUT2
4
GNDA
CIN
VDDQ
LOUT
2.5V
+
Q2
CVDDQ_OUT
ISL6532C
Absolute Maximum Ratings
Thermal Information
5VSBY, P5VSBY . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +7V
P12V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +14V
UGATE, LGATE, NCH . . . . . . . . . . . . . . GND - 0.3V to P12V + 0.3V
All other Pins . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5VCC + 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Level 1
Thermal Resistance (Typical, Notes 1, 2)
Recommended Operating Conditions
JA (oC/W) JC (oC/W)
QFN Package . . . . . . . . . . . . . . . . . . .
32
5
Maximum Junction Temperature (Plastic Package) . . . . . . . 150oC
Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC
(SOIC - Lead Tips Only)
Supply Voltage on 5VSBY . . . . . . . . . . . . . . . . . . . . . . . . +5V 10%
Supply Voltage on P12V . . . . . . . . . . . . . . . . . . . . . . . . +12V 10%
Supply Voltage on P5VSBY . . . . . . . . . . . . . . . . . . . . . . . +5V 10%
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . 0oC to 70oC
Junction Temperature Range. . . . . . . . . . . . . . . . . . . 0oC to 125oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams and Typical Application Schematics
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
5VSBY SUPPLY CURRENT
Nominal Supply Current
ICC_S0
S3# & S5# HIGH, UGATE/LGATE Open
3.00
5.25
7.25
mA
ICC_S3
S3# LOW, S5# HIGH, UGATE/LGATE
Open
3.50
-
4.75
mA
ICC_S5
S5# LOW, S3# Don’t Care,
UGATE/LGATE Open
300
-
800
A
Rising 5VSBY POR Threshold
4.00
-
4.35
V
Falling 5VSBY POR Threshold
3.60
-
3.95
V
Rising P12V POR Threshold
10.0
-
10.5
V
Falling P12V POR Threshold
8.80
-
9.75
V
POWER-ON RESET
OSCILLATOR AND SOFT-START
PWM Frequency
fOSC
220
250
280
kHz
Ramp Amplitude
VOSC
-
1.5
-
V
Error Amp Reset Time
tRESET
Mechanical Off/S5 to S0
6.5
-
9.5
ms
tSS
Mechanical Off/S5 to S0
6.5
-
9.5
ms
-
0.800
-
V
-2.0
-
+2.0
%
-
80
-
dB
GBWP
15
-
-
MHz
SR
-
6
-
V/s
S3# Transition Level
VS3
-
1.5
-
V
S5# Transition Level
VS5
-
1.5
-
V
VDDQ Soft-Start Interval
REFERENCE VOLTAGE
Reference Voltage
VREF
System Accuracy
PWM CONTROLLER ERROR AMPLIFIER
DC Gain
Guaranteed By Design
Gain-Bandwidth Product
Slew Rate
STATE LOGIC
5
ISL6532C
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams and Typical Application Schematics (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
PWM CONTROLLER GATE DRIVERS
UGATE and LGATE Source
IGATE
-
-0.8
-
A
UGATE and LGATE Sink
IGATE
-
0.8
-
A
-
-
6
mA
9.0
9.5
10.0
V
P5VSBY = 5.0V
-
-
650
mA
P5VSBY = 3.3V
-
-
550
mA
NCH BACKFEED CONTROL
NCH Current Sink
INCH
NCH Trip Level
VNCH
NCH = 0.8V
VDDQ STANDBY LDO
Output Drive Current
VTT REGULATOR
Upper Divider Impedance
RU
-
2.5
-
k
Lower Divider Impedance
RL
-
2.5
-
k
IVREF_OUT
-
-
2
mA
-3
-
3
A
-3.3
-
3.3
A
-
80
-
dB
GBWP
15
-
-
MHz
SR
-
6
-
V/s
DRIVE2 High Output Voltage
10.0
10.2
-
V
DRIVE2 Low Output Voltage
-
0.16
0.40
V
DRIVE2 High Output Source Current
-.5
-1.4
-
mA
DRIVE2 Low Output Sink Current
.85
1.3
-
mA
VREF_OUT Buffer Source Current
Maximum VTT Load Current
IVTT_MAX
Periodic load applied with 30% duty cycle
and 10ms period using ISL6532CEVAL1
evaluation board (see Application Note
AN1056)
VTT Over Current Trip
ITRIP_VTT
By Design
LINEAR REGULATOR
DC GAIN
Guaranteed By Design
Gain Bandwidth Product
Slew Rate
PGOOD
PGOOD Rising Threshold
VVTTSNS/VVDDQ S0
-
57.5
-
%
PGOOD Falling Threshold
VVTTSNS/VVDDQ S0
-
45.0
-
%
17
20
22
A
PROTECTION
OCSET Current Source
IOCSET
VDDQ OV Level
VFB/VREF
S0
-
115
-
%
VDDQ UV Level
VFB/VREF
S0
-
85
-
%
Linear Regulator OV Level
VFB2/VREF
S0
-
115
-
%
Linear Regulator UV Level
VFB2/VREF
S0
-
85
-
%
By Design
-
140
-
°C
Thermal Shutdown Limit
TSD
6
ISL6532C
Functional Pin Description
5VSBY (Pin 2)
5VSBY is the bias supply of the ISL6532C. It is typically
connected to the 5V standby rail of an ATX power supply.
During S4/S5 sleep states the ISL6532C enters a reduced
power mode and draws less than 1mA (ICC_S5) from the
5VSBY supply. The supply to 5VSBY should be locally
bypassed using a 0.1F capacitor.
P12V (Pin 25)
P12V provides the gate drive to the switching MOSFETs of
the PWM power stage. The VTT regulation circuit and the
Linear Driver are also powered by P12V. P12V is not
required except during S0/S1/S2 operation. P12V is typically
connected to the +12V rail of an ATX power supply.
The FB pin is also monitored for under and over-voltage
events.
PHASE (Pin 20)
Connect this pin to the upper MOSFET’s source. This pin is
used to monitor the voltage drop across the upper MOSFET
for over-current protection.
OCSET (Pin 12)
Connect a resistor (ROCSET) from this pin to the drain of the
upper MOSFET. ROCSET, an internal 20A current source
(IOCSET), and the upper MOSFET on-resistance (rDS(ON))
set the converter over-current (OC) trip point according to
the following equation:
I OCSET xR OCSET
I PEAK = -----------------------------------------------r DS  ON 
5VSBY (Pin 11)
An over-current trip cycles the soft-start function.
This pin provides the VDDQ output power during S3 sleep
state. The regulator is capable of providing standby VDDQ
power from either the 5VSBY or 3.3VSBY rail. It is
recommended that the 5VSBY rail be used as the output
current handling capability of the standby LDO is higher than
with the 3.3VSBY rail.
VDDQ (Pins 7, 8, 9)
The VDDQ pins should be connected externally together to
the regulated VDDQ output. During S0/S1 states, the VDDQ
pins serve as inputs to the VTT regulator and to the VTT
Reference precision divider. During S3 state, the VDDQ pins
serve as an output from the integrated standby LDO.
GND, GNDA, GNDP, GNDQ (Pins 1, 3, 4, 17, 29)
VTT (Pins 5, 6)
The GND terminals of the ISL6532C provide the return path
for the VTT LDO, standby LDO and switching MOSFET gate
drivers. High ground currents are conducted directly through
the exposed paddle of the QFN package which must be
electrically connected to the ground plane through a path as
low in inductance as possible. GNDA is the Analog ground
pin, GNDQ is the return for the VTT regulator and GNDP is
the return for the upper and lower gate drives.
The VTT pins should be connect externally together. During
S0/S1 states, the VTT pins serve as the outputs of the VTT
linear regulator. During S3 state, the VTT regulator is
disabled.
UGATE (Pin 26)
UGATE drives the upper (control) FET of the VDDQ
synchronous buck switching regulator. UGATE is driven
between GND and P12V.
LGATE (Pin 27)
LGATE drives the lower (synchronous) FET of the VDDQ
synchronous buck switching regulator. LGATE is driven
between GND and P12V.
FB (Pin 15) and COMP (Pin 16)
The VDDQ switching regulator employs a single voltage
control loop. FB is the negative input to the voltage loop error
amplifier. The positive input of the error amplifier is
connected to a precision 0.8V reference and the output of
the error amplifier is connected to the COMP pin. The VDDQ
output voltage is set by an external resistor divider
connected to FB. With a properly selected divider, VDDQ can
be set to any voltage between the power rail (reduced by
converter losses) and the 0.8V reference. Loop
compensation is achieved by connecting an AC network
across COMP and FB.
7
VTTSNS (Pin 10)
VTTSNS is used as the feedback for control of the VTT linear
regulator. Connect this pin to the VTT output at the physical
point of desired regulation.
VREF_OUT (Pin 13)
VREF_OUT is a buffered version of VTT and also acts as the
reference voltage for the VTT linear regulator. It is
recommended that a minimum capacitance of 0.1F is
connected between VDDQ and VREF_OUT and also
between VREF_OUT and ground for proper operation.
VREF_IN (Pin 14)
A capacitor, CSS, connected between VREF_IN and ground
is required. This capacitor and the parallel combination of
the Upper and Lower Divider Impedance (RU||RL), sets the
time constant for the start up ramp when transitioning from
S3 to S0/S1/S2.
The minimum value for CSS can be found through the
following equation:
C VTTOUT  V DDQ
C SS  -----------------------------------------------10  2A  R U  R L
The calculated capacitance, CSS, will charge the output
capacitor bank on the VTT rail in a controlled manner without
reaching the current limit of the VTT LDO.
ISL6532C
NCH (Pin 22)
Initialization
NCH is an open-drain output that controls the MOSFET
blocking backfeed from VDDQ to the input rail during sleep
states. A 2k or larger resistor is to be tied between the 12V
rail and the NCH pin. Until the voltage on the NCH pin
reaches the NCH trip level, the PWM is disabled.
The ISL6532C automatically initializes upon receipt of input
power. Special sequencing of the input supplies is not
necessary. The Power-On Reset (POR) function continually
monitors the input bias supply voltages. The POR monitors
the bias voltage at the 5VSBY and P12V pins. The POR
function initiates soft-start operation after the bias supply
voltages exceed their POR thresholds.
If NCH is not actively utilized, it still must be tied to the 12V
rail through a resistor. For systems using 5V dual as the
input to the switching regulator, a time constant, in the form
of a capacitor, can be added to the NCH pad to delay start of
the PWM switcher until the 5V dual has switched from
5VSBY to 5VATX.
PGOOD (Pin 21)
Power Good is an open-drain logic output that changes to a
logic low if any of the three regulators are out of regulation in
S0/S1/S2 state. PGOOD will always be low in any state
other than S0/S1/S2.
SLP_S5# (Pin 24)
This pin accepts the SLP_S5# sleep state signal.
SLP_S3# (Pin 23)
This pin accepts the SLP_S3# sleep state signal.
FB2 (Pin 18)
Connect the output of the external linear regulator to this pin
through a properly sized resistor divider. The voltage at this
pin is regulated to 0.8V. This pin is monitored for under and
over-voltage events.
DRIVE2 (Pin 19)
Connect this pin to the gate terminal of an external NChannel MOSFET transistor. This pin provides the gate
voltage for the linear regulator pass transistor. It also
provides a means of compensating the error amplifier for
applications requiring the transient response of the linear
regulator to be optimized.
Functional Description
ACPI State Transitions
Cold Start (S4/S5 to S0 Transition)
At the onset of a mechanical start, the ISL6532C receives its
bias voltage from the 5V Standby bus (5VSBY). As soon as
the SLP_S3 and SLP_S5 have transitioned HIGH, the
ISL6532C starts an internal counter. Following a cold start or
any subsequent S4/S5 state, state transitions are ignored
until the system enters S0/S1. None of the regulators will
begin the soft start procedure until the 5V Standby bus has
exceeded POR, the 12V bus has exceeded POR and VNCH
has exceeded the trip level.
Once all of these conditions are met, the PWM error
amplifier will first be reset by internally shorting the COMP
pin to the FB pin. This reset lasts for 2048 clock cycles,
which is typically 8.2ms (one clock cycle = 1/fOSC). The
digital soft start sequence will then begin.
The PWM error amplifier reference input is clamped to a
level proportional to the soft-start voltage. As the soft-start
voltage slews up, the PWM comparator generates PHASE
pulses of increasing width that charge the output
capacitor(s). The internal VTT LDO will also soft start
through the reference that tracks the output of the PWM
regulator. The reference for the AGP LDO controller will rise
relative to the soft start reference. The soft start lasts for
2048 clock cycles, which is typically 8.2ms. This method
provides a rapid and controlled output voltage rise.
S3
S5
Overview
The ISL6532C provides complete control, drive, protection
and ACPI compliance for a regulator powering DDR memory
systems. It is primarily designed for computer applications
powered from an ATX power supply. A 250kHz Synchronous
Buck Regulator with a precision 0.8V reference provides the
proper Core voltage to the system memory of the computer.
An internal LDO regulator with the ability to both sink and
source current and an externally available buffered
reference that tracks the VDDQ output by 50% provides the
VTT termination voltage. The ISL6532C also features an
LDO regulator for 1.5V AGP Video and Core voltage.
ACPI compliance is realized through the SLP_S3 and
SLP_S5 sleep signals and through monitoring of the 12V
ATX bus.
8
12VATX 2V/DIV
5VSBY
1V/DIV
VDDQ
500mV/DIV
VAGP
500mV/DIV
VTT
500mV/DIV
PGOOD
5V/DIV
2048 CLOCK
CYCLES
12V POR
2048 CLOCK
CYCLES
SOFT START ENDS
SOFT START
INITIATES PGOOD COMPARATOR
ENABLED
FIGURE 1. TYPICAL COLD START
ISL6532C
Figure 1 shows the soft start sequence for a typical cold
start. Due to the soft start capacitance, CSS, on the
VREF_IN pin, the S5 to S0 transition profile of the VTT rail
will have a more rounded features at the start and end of the
soft start whereas the VDDQ profile has distinct starting and
ending points to the ramp up.
By directly monitoring 12VATX and the SLP_S3 and SLP_S5
signals the ISL6532C can achieve PGOOD status
significantly faster than other devices that depend on
Latched_Backfeed_Cut for timing.
S3
S5
12VATX 2V/DIV
VAGP
500mV/DIV
VTT_FLOAT
VTT
500mV/DIV
Active to Sleep (S0 to S3 Transition)
When SLP_S3 goes LOW with SLP_S5 still HIGH, the
ISL6532C will disable the VTT linear regulator and the AGP
LDO controller. The VDDQ standby regulator will be enabled
and the VDDQ switching regulator will be disabled. NCH is
pulled low to disable the backfeed blocking MOSFET.
PGOOD will also transition LOW. When VTT is disabled, the
internal reference for the VTT regulator is internally shorted
to the VTT rail. This allows the VTT rail to float. When
floating, the voltage on the VTT rail will depend on the
leakage characteristics of the memory and MCH I/O pins. It
is important to note that the VTT rail may not bleed down to 0V.
The VDDQ rail will be supported in the S3 state through the
standby VDDQ LDO. When S3 transitions LOW, the Standby
regulator is immediately enabled. The switching regulator is
disabled synchronous to the switching waveform. The shut
off time will range between 4 and 8s. The standby LDO is
capable of supporting up to 650mA of load with P5VSBY tied
to the 5V Standby Rail. The standby LDO may receive input
from either the 3.3V Standby rail or the 5V Standby rail
through the P5VSBY pin. It is recommended that the 5V
Standby rail be used as the current delivery capability of the
LDO is greater.
Sleep to Active (S3 to S0 Transition)
When SLP_S3 transitions from LOW to HIGH with SLP_S5
held HIGH and after the 12V rail exceeds POR, the
ISL6532C will enable the VDDQ switching regulator, disable
the VDDQ standby regulator, enable the VTT LDO and force
the NCH pin to a high impedance state turning on the
blocking MOSFET. The AGP LDO goes through a 2048
clock cycle soft-start. The internal short between the VTT
reference and the VTT rail is released. Upon release of the
short, the capacitor on VREF_IN is then charged up through
the internal resistor divider network. The VTT output will
follow this capacitor charge up, and acting as the S3 to S0
transition soft start for the VTT rail. The PGOOD comparator
is enabled only after 2048 clock cycles, or typically 8.2ms,
have passed following the S3 transition to a HIGH state.
Figure 2 illustrates a typical state transition from S3 to S0. It
should be noted that the soft start profile of the VTT LDO
output will vary according to the value of the capacitor on the
VREF_IN pin.
9
VDDQ
500mV/DIV
PGOOD
5V/DIV
2048 CLOCK
CYCLES
12V POR
PGOOD COMPARATOR
ENABLED
FIGURE 2. TYPICAL S3 TO S0 STATE TRANSITION
Active to Shutdown (S0 to S5 Transition)
When the system transitions from active, S0, state to
shutdown, S4/S5, state, the ISL6532C IC disables all
regulators and forces the PGOOD pin and the NCH pin LOW.
VDDQ Over Current Protection (S0 State)
The over-current function protects the switching converter
from a shorted output by using the upper MOSFET onresistance, rDS(ON), to monitor the current. This method
enhances the converter’s efficiency and reduces cost by
eliminating a current sensing resistor.
The over-current function cycles the soft-start function in a
hiccup mode to provide fault protection. A resistor (ROCSET)
programs the over-current trip level (see Typical Application
diagrams on pages 3 and 4). An internal 20A (typical)
current sink develops a voltage across ROCSET that is
referenced to the converter input voltage. When the voltage
across the upper MOSFET (also referenced to the converter
input voltage) exceeds the voltage across ROCSET, the overcurrent function initiates a soft-start sequence. The initiation
of soft start will affect all regulators. The VTT regulator is
directly affected as it receives it’s reference from VDDQ. The
AGP LDO will also be soft started, and as such, the AGP
LDO voltage will be disabled while the VDDQ regulator is
disabled.
Figure 3 illustrates the protection feature responding to an
over current event. At time T0, an over current condition is
sensed across the upper MOSFET. As a result, the regulator
is quickly shutdown and the internal soft-start function begins
producing soft-start ramps. The delay interval seen by the
output is equivalent to three soft-start cycles. The fourth
internal soft-start cycle initiates a normal soft-start ramp of
the output, at time T1. The output is brought back into
regulation by time T2, as long as the over current event has
cleared.
ISL6532C
Had the cause of the over current still been present after the
delay interval, the over current condition would be sensed
and the regulator would be shut down again for another
delay interval of three soft start cycles. The resulting hiccup
mode style of protection would continue to repeat indefinitely.
VDDQ
VTT Over Current Protection
The internal VTT LDO is protected from fault conditions
through a 3.3A current limit. This current limit protects the
ISL6532C if the LDO is sinking or sourcing current. During
an overcurrent event on the VTT LDO, only the VTT LDO is
disabled. Once the over current condition on the VTT rail is
removed, VTT will recover.
Over/Under Voltage Protection
All three regulators are protected from faults through internal
Over/Under voltage detection circuitry. If the any rail falls
below 85% of the targeted voltage, then an undervoltage
event is tripped. An under voltage will disable all three
regulators for a period of 3 soft-start cycles, after which a
normal soft-start is initiated. If the output is still under 85% of
target, the regulators will continue to be disabled and softstarted in a hiccup mode until the fault is cleared. This
protection feature works much the same as the VDDQ PWM
over current protection works. See Figure 3.
VAGP
VTT
500mV/DIV
INTERNAL SOFT-START FUNCTION
If the any rail exceeds 115% of the targeted voltage, then all
three outputs are immediately disabled. The ISL6532C will
not re-enable the outputs until either the bias voltage is
toggled in order to initiate a POR or the S5 signal is forced
LOW and then back to HIGH.
DELAY INTERVAL
T0
T1
TIME
T2
FIGURE 3. VDDQ and VTT OVER CURRENT PROTECTION
AND VTT/VAGP LDO UNDER VOLTAGE
PROTECTION RESPONSES
The over-current function will trip at a peak inductor current
(IPEAK) determined by:
Thermal Protection (S0/S3 State)
If the ISL6532C IC junction temperature reaches a nominal
temperature of 140oC, all regulators will be disabled. The
ISL6532C will not re-enable the outputs until the junction
temperature drops below 110oC and either the bias voltage
is toggled in order to initiate a POR or the SLP_S5 signal is
forced LOW and then back to HIGH.
Shoot-Through Protection
I OCSET x R OCSET
I PEAK = ---------------------------------------------------r DS  ON 
where IOCSET is the internal OCSET current source (20A
typical). The OC trip point varies mainly due to the MOSFET
rDS(ON) variations. To avoid over-current tripping in the
normal operating load range, find the ROCSET resistor from
the equation above with:
1. The maximum rDS(ON) at the highest junction
temperature.
2. The minimum IOCSET from the specification table.
3. Determine IPEAK for:
 I 
I PEAK > I OUT  MAX  + ---------- , whereI is
2
the output inductor ripple current.
For an equation for the ripple current see the section under
component guidelines titled ‘Output Inductor Selection’.
A small ceramic capacitor should be placed in parallel with
ROCSET to smooth the voltage across ROCSET in the
presence of switching noise on the input voltage.
10
A shoot-through condition occurs when both the upper and
lower MOSFETs are turned on simultaneously, effectively
shorting the input voltage to ground. To protect from a shootthrough condition, the ISL6532C incorporates specialized
circuitry which insures that complementary MOSFETs are
not ON simultaneously.
The adaptive shoot-through protection utilized by the VDDQ
regulator looks at the lower gate drive pin, LGATE, and the
upper gate drive pin, UGATE, to determine whether a
MOSFET is ON or OFF. If the voltage from UGATE or from
LGATE to GND is less than 0.8V, then the respective
MOSFET is defined as being OFF and the other MOSFET is
allowed to turned ON. This method allows the VDDQ
regulator to both source and sink current.
Since the voltage of the MOSFET gates are being measured
to determine the state of the MOSFET, the designer is
encouraged to consider the repercussions of introducing
external components between the gate drivers and their
respective MOSFET gates before actually implementing
such measures. Doing so may interfere with the shootthrough protection.
ISL6532C
12VATX
CBP
P12V
GNDP
Layout Considerations
Layout is very important in high frequency switching
converter design. With power devices switching efficiently at
250kHz, the resulting current transitions from one device to
another cause voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit,
and lead to device over-voltage stress. Careful component
layout and printed circuit board design minimizes these
voltage spikes.
NCH
5VSBY
GNDP
VDDQ
PHASE
LGATE
COMP
C2
C1
R2
R1
FB
C3 R3
R4
VDDQ(3)
VDDQ
VTT(2)
VTT
COUT2
VIN_AGP
Q3
DRIVE2
GND PAD
COUT1
Q2
R5
VAGP
FB2
R6
COUT3
LOAD
11
Q1 LOUT
LOAD
The switching components should be placed close to the
ISL6532C first. Minimize the length of the connections
between the input capacitors, CIN, and the power switches
by placing them nearby. Position both the ceramic and bulk
input capacitors as close to the upper MOSFET drain as
CIN
CBP
UGATE
There are two sets of critical components in the ISL6532C
switching converter. The switching components are the most
critical because they switch large amounts of energy, and
therefore tend to generate large amounts of noise. Next are
the small signal components which connect to sensitive
nodes or supply critical bypass current and signal coupling.
In order to dissipate heat generated by the internal VTT
LDO, the ground pad, pin 29, should be connected to the
internal ground plane through at least four vias. This allows
the heat to move away from the IC and also ties the pad to
the ground plane through a low impedance path.
5VSBY
P5VSBY
As an example, consider the turn-off transition of the control
MOSFET. Prior to turn-off, the MOSFET is carrying the full
load current. During turn-off, current stops flowing in the
MOSFET and is picked up by the lower MOSFET. Any
parasitic inductance in the switched current path generates a
large voltage spike during the switching interval. Careful
component selection, tight layout of the critical components,
and short, wide traces minimizes the magnitude of voltage
spikes.
A multi-layer printed circuit board is recommended. Figure 4
shows the connections of the critical components in the
converter. Note that capacitors CIN and COUT could each
represent numerous physical capacitors. Dedicate one solid
layer, usually a middle layer of the PC board, for a ground
plane and make all critical component ground connections
with vias to this layer. Dedicate another solid layer as a
power plane and break this plane into smaller islands of
common voltage levels. Keep the metal runs from the
PHASE terminals to the output inductor short. The power
plane should support the input power and output power
nodes. Use copper filled polygons on the top and bottom
circuit layers for the phase nodes. Use the remaining printed
circuit layers for small signal wiring. The wiring traces from
the GATE pins to the MOSFET gates should be kept short
and wide enough to easily handle the 1A of drive current.
VIN_DDR
ISL6532C
LOAD
Application Guidelines
KEY
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 4. PRINTED CIRCUIT BOARD POWER PLANES
AND ISLANDS
possible. Position the output inductor and output capacitors
between the upper and lower MOSFETs and the load.
The critical small signal components include any bypass
capacitors, feedback components, and compensation
components. Place the PWM converter compensation
components close to the FB and COMP pins. The feedback
resistors should be located as close as possible to the FB
pin with vias tied straight to the ground plane as required.
Feedback Compensation - PWM Buck Converter
Figure 5 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The error
amplifier output (VE/A) is compared with the oscillator (OSC)
triangular wave to provide a pulse-width modulated (PWM)
wave with an amplitude of VIN at the PHASE node.
ISL6532C
The PWM wave is smoothed by the output filter (LO and
CO).
VIN
DRIVER
OSC
PWM
COMPARATOR
DRIVER
+
5. Place 2ND Pole at Half the Switching Frequency.
6. Check Gain against Error Amplifier’s Open-Loop Gain.
LO
-
VOSC
4. Place 1ST Pole at the ESR Zero.
VDDQ
PHASE
CO
7. Estimate Phase Margin - Repeat if Necessary.
Compensation Break Frequency Equations
ESR
(PARASITIC)
ZFB
VE/A
-
ZIN
REFERENCE
DETAILED COMPENSATION COMPONENTS
ZFB
C1
C2
VDDQ
ZIN
C3
R2
R3
R1
COMP
FB
+
R4
ISL6532C
REFERENCE
R 

V DDQ = 0.8   1 + ------1-
R 4

FIGURE 5. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN AND OUTPUT
VOLTAGE SELECTION
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR . The DC Gain of
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage VOSC .
Modulator Break Frequency Equations
1
F LC = ------------------------------------------2 x L O x C O
1
F ESR = -------------------------------------------2 x ESR x C O
The compensation network consists of the error amplifier
(internal to the ISL6532C) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R1 , R2 ,
R3 , C1 , C2 , and C3) in Figure 5. Use these guidelines for
locating the poles and zeros of the compensation network:
1. Pick Gain (R2/R1) for desired converter bandwidth.
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).
3. Place 2ND Zero at Filter’s Double Pole.
12
1
F P1 = -------------------------------------------------------- C 1 x C 2
2 x R 2 x  ----------------------
 C1 + C2 
1
F Z2 = ------------------------------------------------------2 x  R 1 + R 3  x C 3
1
F P2 = -----------------------------------2 x R 3 x C 3
Figure 6 shows an asymptotic plot of the DC-DC converter’s
gain vs. frequency. The actual Modulator Gain has a high
gain peak due to the high Q factor of the output filter and is
not shown in Figure 6. Using the above guidelines should
give a Compensation Gain similar to the curve plotted. The
open loop error amplifier gain bounds the compensation
gain. Check the compensation gain at FP2 with the
capabilities of the error amplifier. The Closed Loop Gain is
constructed on the graph of Figure 6 by adding the
Modulator Gain (in dB) to the Compensation Gain (in dB).
This is equivalent to multiplying the modulator transfer
function to the compensation transfer function and plotting
the gain.
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW) overall
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
100
FZ1 FZ2
FP1
FP2
80
OPEN LOOP
ERROR AMP GAIN
60
GAIN (dB)
+
ERROR
AMP
1
F Z1 = -----------------------------------2 x R 2 x C 2
40
20
20LOG
(R2/R1)
20LOG
(VIN/VOSC)
0
COMPENSATION
GAIN
MODULATOR
GAIN
-20
-40
-60
CLOSED LOOP
GAIN
FLC
10
100
1K
FESR
10K
100K
1M
10M
FREQUENCY (Hz)
FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Feedback Compensation - AGP LDO Controller
Figure 7 shows the AGP LDO power and control stage. This
LDO, which uses a MOSFET as the linear pass element,
requires feedback compensation to insure stability of the
system. The LDO requires compensation because of the
output impedance of the error amplifier.
ISL6532C
ISL6532C
is 249. The voltage programming resistor, R9 can be
calculated based on the following equation:
VDDQ
0.8V
REFERENCE
R 8  0.8V
R 9 = ---------------------------------V AGP – 0.8V
650
+
-
OUTPUT
IMPEDANCE
DRIVE2
Component Selection Guidelines
C25
VAGP
R10
FB2
R8
R9
ESR
R 

V AGP = 0.8   1 + ------8-
R 9

COUT
RLOAD
+
FIGURE 7. COMPENSATION AND OUTPUT VOLTAGE
SELECTION OF THE LINEAR
To properly compensate the LDO system, a 100k 1%
resistor and a 680pF X5R ceramic capacitor, represented as
R10 and C25 in Figure 7, are used. This compensation will
insure a stable system with any MOSFET given the following
conditions:
 = C OUT  ESR  10s
R FB = R 8 = 249
Maximum bandwidth will be realized at full load while
minimum bandwidth will be realized at no load. Bandwidth at
no load will be maximized as  becomes closer to 10s.
Output Voltage Selection
The output voltage of the VDDQ PWM converter can be
programmed to any level between VIN and the internal
reference, 0.8V. An external resistor divider is used to scale
the output voltage relative to the reference voltage and feed
it back to the inverting input of the error amplifier, see
Figure 5. However, since the value of R1 affects the values of
the rest of the compensation components, it is advisable to
keep its value less than 5k. Depending on the value chosen
for R1, R4 can be calculated based on the following equation:
R1  0.8V
R4 = ----------------------------------V DDQ – 0.8V
If the output voltage desired is 0.8V, simply route VDDQ back
to the FB pin through R1, but do not populate R4.
The output voltage for the internal VTT linear regulator is set
internal to the ISL6532C to track the VDDQ voltage by 50%.
There is no need for external programming resistors.
As with the VDDQ PWM regulator, the AGP linear regulator
output voltage is set by means of an external resistor divider
as shown in Figure 7. For stability concerns described
earlier, the recommended value of the feedback resistor, R8,
13
Output Capacitor Selection - PWM Buck Converter
An output capacitor is required to filter the inductor current
and supply the load transient current. The filtering
requirements are a function of the switching frequency and
the ripple current. The load transient requirements are a
function of the slew rate (di/dt) and the magnitude of the
transient load current. These requirements are generally met
with a mix of capacitors and careful layout.
DDR memory systems are capable of producing transient
load rates above 1A/ns. High frequency capacitors initially
supply the transient and slow the current load rate seen by
the bulk capacitors. The bulk filter capacitor values are
generally determined by the ESR (Effective Series
Resistance) and voltage rating requirements rather than
actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient.
An aluminum electrolytic capacitor’s ESR value is related to
the case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient
loading. Unfortunately, ESL is not a specified parameter.
Work with your capacitor supplier and measure the
capacitor’s impedance with frequency to select a suitable
component. In most cases, multiple electrolytic capacitors of
small case size perform better than a single large case
capacitor.
Output Capacitor Selection - LDO Regulators
The output capacitors used in LDO regulators are used to
provide dynamic load current. The amount of capacitance
and type of capacitor should be chosen with this criteria in
mind.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
ISL6532C
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
I =
VIN - VOUT
Fs x L
x
VOUT
VIN
VOUT = I x ESR
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6532C will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
VOUT
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Input Capacitor Selection - PWM Buck Converter
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time the upper MOSFET
turns on. Place the small ceramic capacitors physically close
to the MOSFETs and between the drain of upper MOSFET
and the source of lower MOSFET.
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. Their voltage rating should be
at least 1.25 times greater than the maximum input voltage,
while a voltage rating of 1.5 times is a conservative
guideline. For most cases, the RMS current rating
requirement for the input capacitor of a buck regulator is
approximately 1/2 the DC load current.
The maximum RMS current required by the regulator may be
closely approximated through the following equation:
I RMS
MAX
=
V OUT 
V IN – V OUT V OUT 2
2
1
--------------  I OUT
+ ------   -----------------------------  -------------- 

V IN
V IN  
12  L  f s
MAX
For a through hole design, several electrolytic capacitors
may be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge-current at
power-up. Some capacitor series available from reputable
manufacturers are surge current tested.
MOSFET Selection - PWM Buck Converter
The ISL6532C requires 2 N-Channel power MOSFETs for
switching power and a third MOSFET to block backfeed from
VDDQ to the Input in S3 Mode. These should be selected
based upon rDS(ON) , gate supply requirements, and thermal
management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor. The switching losses
seen when sourcing current will be different from the switching
losses seen when sinking current. When sourcing current, the
upper MOSFET realizes most of the switching losses. The lower
switch realizes most of the switching losses when the converter
is sinking current (see the equations below). These equations
assume linear voltage-current transitions and do not adequately
model power loss due the reverse-recovery of the upper and
lower MOSFET’s body diode. The gate-charge losses are
dissipated in part by the ISL6532C and do not significantly heat
the MOSFETs. However, large gate-charge increases the
switching interval, tSW which increases the MOSFET switching
losses. Ensure that both MOSFETs are within their maximum
junction temperature at high ambient temperature by calculating
the temperature rise according to package thermal-resistance
specifications. A separate heatsink may be necessary
depending upon MOSFET power, package type, ambient
temperature and air flow.
Approximate Losses while Sourcing current
2
1
P UPPER = Io  r DS  ON   D + ---  Io  V IN  t SW  f s
2
PLOWER = Io2 x rDS(ON) x (1 - D)
Approximate Losses while Sinking current
PUPPER = Io2 x rDS(ON) x D
2
1
P LOWER = Io  r DS  ON    1 – D  + ---  Io  V IN  t SW  f s
2
Where: D is the duty cycle = VOUT / VIN ,
tSW is the combined switch ON and OFF time, and
fs is the switching frequency.
14
ISL6532C
MOSFET Selection - AGP LDO
ISL6532C Application Circuit
The main criteria for selection of the linear regulator pass
transistor is package selection for efficient removal of heat.
Select a package and heatsink that maintains the junction
temperature below the rating with a maximum expected
ambient temperature.
Figure 8 shows an application circuit utilizing the ISL6532C.
Detailed information on the circuit, including a complete Billof-Materials and circuit board description, can be found in
Application Note AN1056.
The power dissipated in the linear regulator is:
P LINEAR  I O   V IN – V OUT 
where IO is the maximum output current and VOUT is the
nominal output voltage of the linear regulator.
VCC5
5VSBY
PGOOD
VDDQ
S5#
SLP_S3
S3#
5VSBY
SLP_S5
+
VTT
VTT
C21
220F
C23
220F
VDDQ
Q2,4
R4
1.74k
GNDQ
GNDQ
FB
R9
287
COMP
C15
1000pF
C14
6.8nF
R3
19.1k
C13
56nF
R6
825
FIGURE 8. DDR SDRAM AND AGP VOLTAGE REGULATOR USING THE ISL6532C
15
C1-3
2200F
2.5V 15AMAX
100k
R8
249
+
Q1,3
VDDQ
VDDQ
VDDQ
DRIVE2
R10
FB2
+
8.87k
LGATE
VTTSNS
C25 680pF
1.5V
C4,5
1F
L2
2.1H
GNDA
VAGP
R7
1000pF
UGATE
GNDP
Q4
C22
ISL6532C
GNDP
VDDQ
L1
2.1H
PHASE
C20 +
220F
VTT
1.25V
C24
1F
VREF_IN
C19
0.47F
VDDQ
Q5
NCH
OCSET
VREF_OUT
C27
0.1F
R1
4.99k
C16
1F
PGOOD
C26
0.1F
VREF
C17,18
1F
P12V
R2
10.0k
VCC12
P5VSBY
+3.3V
R5
22.6
+
C6-8
1800F
C9-12
22F
ISL6532C
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L28.6x6
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VJJC ISSUE C)
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
9
A3
b
0.20 REF
0.23
D
0.35
5, 8
6.00 BSC
D1
D2
0.28
9
-
5.75 BSC
3.95
4.10
9
4.25
7, 8
E
6.00 BSC
-
E1
5.75 BSC
9
E2
3.95
e
4.10
4.25
7, 8
0.65 BSC
-
k
0.25
-
-
-
L
0.35
0.60
0.75
8
L1
-
-
0.15
10
N
28
2
Nd
7
3
Ne
7
3
P
-
-
0.60
9

-
-
12
9
Rev. 1 10/02
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P &  are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
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Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
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16
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