DATASHEET

HIP6304
TM
Data Sheet
Microprocessor CORE Voltage Regulator
Multi-Phase Buck PWM Controller
The HIP6304 multi-phase PWM control IC together with its
companion gate drivers, the HIP6601, HIP6602 or HIP6603
and Intersil MOSFETs provides a precision voltage
regulation system for advanced microprocessors.
Multiphase power conversion is a marked departure from
earlier single phase converter configurations previously
employed to satisfy the ever increasing current demands of
modern microprocessors. Multi-phase convertors, by
distributing the power and load current results in smaller and
lower cost transistors with fewer input and output capacitors.
These reductions accrue from the higher effective
conversion frequency with higher frequency ripple current
due to the phase interleaving process of this topology. For
example, a two phase convertor operating at 350kHz will
have a ripple frequency of 700kHz. Moreover, greater
convertor bandwidth of this design results in faster response
to load transients.
Outstanding features of this controller IC include
programmable VID codes from the microprocessor that
range from 1.30V to 2.05V with a system accuracy of ±1%.
Pull up currents on these VID pins eliminates the need for
external pull up resistors. In addition “droop” compensation,
used to reduce the overshoot or undershoot of the CORE
voltage, is easily programmed with a single resistor.
Another feature of this controller IC is the PGOOD monitor
circuit which is held low until the CORE voltage increases,
during its Soft-Start sequence, to within 10% of the
programmed voltage. Over-voltage, 15% above programmed
CORE voltage, results in the converter shutting down and
turning the lower MOSFETs ON to clamp and protect the
microprocessor. Under voltage is also detected and results
in PGOOD low if the CORE voltage falls 10% below the
programmed level. Over-current protection reduces the
regulator current to less than 25% of the programmed trip
value. These features provide monitoring and protection for
the microprocessor and power system.
March 2000
4840
Features
• AMD Athlon Compatible Multi-Phase Power Conversion
• Precision Channel Current Sharing
- Loss Less Current Sampling - Uses rDS(ON)
• Precision CORE Voltage Regulation
- ±1% System Accuracy Over Temperature
• Microprocessor Voltage Identification Input
- 4-Bit VID Input
- 1.30V to 2.05V in 50mV Steps
- Programmable “Droop” Voltage
• Fast Transient Recovery Time
• Over Current Protection
• High Ripple Frequency, (Channel Frequency) Times
Number Channels . . . . . . . . . . . . . . . . . 100kHz to 3MHz
Ordering Information
PART NUMBER
TEMP. (oC)
HIP6304CB
0 to 70
PACKAGE
16 Ld SOIC
HIP6304CB-T
16 Ld SOIC Tape and Reel
HIP6304EVAL1
Evaluation Platform
PKG. NO.
M16.15
Pinout
HIP6304 (SOIC)
TOP VIEW
VID3 1
16 VCC
VID2 2
15 PGOOD
VID1 3
14 ISEN1
VID0 4
13 PWM1
EN 5
12 PWM2
COMP 6
11 ISEN2
FB 7
10 VSEN
FS/DIS 8
1
File Number
9 GND
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
Athlon™ is a trademark of Advanced Micro Devices, Inc.
1-888-INTERSIL or 321-724-7143 | Intersil and Design is a trademark of Intersil Corporation. | Copyright © Intersil Corporation 2000
HIP6304
Block Diagram
VCC
PGOOD
VSEN
+
POWER-ON
RESET (POR)
UV
THREE
STATE
-
X 0.9
OV
LATCH
CLOCK AND
SAWTOOTH
GENERATOR
S
+
OVP
FS/EN
-
X1.15
+
SOFTSTART
AND FAULT
LOGIC
EN
∑
+
-
PWM1
PWM
COMP
VID0
+
VID1
∑
+
D/A
VID2
PWM2
PWM
+
E/A
VID3
-
-
CURRENT
CORRECTION
FB
I_TOT
∑
+
OC
-
+
+
I_TRIP
ISEN1
ISEN2
GND
Simplified Power System Diagram
VSEN
PWM 1
SYNCHRONOUS
RECTIFIED BUCK
CHANNEL
HIP6304
MICROPROCESSOR
PWM 2
VID
2
SYNCHRONOUS
RECTIFIED BUCK
CHANNEL
HIP6304
Typical Application - Two Phase Converter Using HIP6601 Gate Drivers
+12V
PVCC
VCC
+5V
PWM
VIN = +5V
BOOT
UGATE
PHASE
DRIVER
HIP6601
LGATE
GND
COMP
FB
+VCORE
+12V
VCC
VSEN
PWM2
VID3
VID2
VID1
PVCC
ISEN2
EN
VIN = +5V
BOOT
PGOOD
UGATE
VCC
MAIN
CONTROL
HIP6304
PWM
PWM1
VID0
PHASE
DRIVER
HIP6601
LGATE
GND
FS/DIS
ISEN1
GND
Typical Application - Two Phase Converter Using a HIP6602 Gate Driver
+5V
BOOT1
+12V
FB
V IN = +12V
UGATE1
COMP
VCC
VCC
VSEN
PHASE1
ISEN1
PGOOD
PWM1
EN
VID3
VID2
LGATE1
PWM1
DUAL
DRIVER
HIP6602
MAIN
CONTROL
HIP6304
PVCC
BOOT2
VID1
PWM2
VID0
PWM2
PHASE2
ISEN2
FS/DIS
UGATE2
LGATE2
GND
GND
3
PGND
+VCORE
+5V
V IN +12V
HIP6304
RIN
VIN
FB
HIP6304
ERROR
AMPLIFIER
COMPARATOR
CORRECTION
-
∑
+
+
Q1
-
PWM
CIRCUIT
+
L01
PWM1
HIP6601
IL1
Q2
PROGRAMMABLE
REFERENCE
DAC
+
∑
CURRENT
PHASE
ISEN1
SENSING
I AVERAGE
RISEN1
CURRENT
AVERAGING
-
∑
+
CURRENT
+
SENSING
-
VCORE
RISEN2
ISEN2
VIN
COMPARATOR
∑
+
CORRECTION
-
Q3
PWM
CIRCUIT
PHASE
COUT
RLOAD
L02
PWM2
HIP6601
IL2
Q4
FIGURE 1. SIMPLIFIED BLOCK DIAGRAM OF THE HIP6304 VOLTAGE AND CURRENT CONTROL LOOPS FOR A TWO POWER
CHANNEL REGULATOR
Functional Pin Description
FS/DIS (Pin 8)
VID3 1
16 VCC
VID2 2
15 PGOOD
VID1 3
14 ISEN1
Channel frequency, FSW, select and disable. A resistor from
this pin to ground sets the switching frequency of the
converter. Pulling this pin to ground disables the converter
and three states the PWM outputs. See Figure 10.
VID0 4
13 PWM1
GND (Pin 9)
EN 5
12 PWM2
COMP 6
11 ISEN2
Bias and reference ground. All signals are referenced to this
pin.
FB 7
10 VSEN
9 GND
FS/DIS 8
VID3 (Pin 1), VID2 (Pin 2), VID1 (Pin 3) and VID0
(Pin 4)
VSEN (Pin 10)
Power good monitor input. Connect to the microprocessorCORE voltage.
ISEN2 (Pin 11) and ISEN1 (Pin 14)
Voltage Identification inputs from microprocessor. These pins
respond to TTL and 3.3V logic signals. The HIP6304 decodes
VID bits to establish the output voltage. See Table 1.
Current sense inputs from the individual converter channel’s
phase nodes.
EN (Pin 5)
PWM outputs for each driven channel in use. Connect these
pins to the PWM input of a HIP6601/2/3 driver.
Enable pin normal operation is with input open or high. A low
input disables the regulator and three states the PWM outputs.
COMP (Pin 6)
PWM2 (Pin 12) and PWM1 (Pin 13)
PGOOD (Pin 15)
Output of the internal error amplifier. Connect this pin to the
external feedback and compensation network.
Power good. This pin provides a logic-high signal when the
microprocessor CORE voltage (VSEN pin) is within specified
limits and Soft-Start has timed out.
FB (Pin 7)
VCC (Pin 16)
Inverting input of the internal error amplifier.
Bias supply. Connect this pin to a 5V supply.
4
HIP6304
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +7V
Input, Output, or I/O Voltage . . . . . . . . . . GND -0.3V to VCC + 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class TBD
Thermal Resistance (Typical, Note 1)
θJA (oC/W)
SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
106
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . .150oC
Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC
(SOIC - Lead Tips Only)
Recommended Operating Conditions
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5%
Ambient Temperature. . . . . . . . . . . . . . . . . . . . . . . . . . . 0oC to 70oC
CAUTION: Stress above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational section of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on an evaluation PC board in free air.
Operating Conditions: VCC = 5V, TA = 0oC to 70oC, Unless Otherwise Specified
Electrical Specifications
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
-
10
15
mA
INPUT SUPPLY POWER
Input Supply Current
RT = 100kΩ, Active and Disabled Maximum Limit
POR (Power-On Reset) Threshold
VCC Rising
4.25
4.38
4.5
V
VCC Falling
3.75
3.88
4.00
V
REFERENCE AND DAC
System Accuracy
Percent system deviation from programmed VID Codes
-1
-
1
%
DAC (VID0 - VID3) Input Low Voltage
DAC Programming Input Low Threshold Voltage
-
-
0.8
V
DAC (VID0 - VID3) Input High Voltage
DAC Programming Input High Threshold Voltage
2.0
-
-
V
VID Pull-Up
VIDx = 0V or VIDx = 3V
10
20
40
µA
Frequency, FSW
RT = 100kΩ, ±1%
245
275
305
kHz
Adjustment Range
See Figure 10
0.05
-
1.5
MHz
Disable Voltage
Maximum voltage at FS/DIS to disable controller. IFS/DIS = 1mA.
-
-
1.0
V
DC Gain
RL = 10K to Ground
-
72
-
dB
Gain-Bandwidth Product
CL = 100pF, RL = 10K to Ground
-
18
-
MHz
Slew Rate
CL = 100pF, Load = ±400µA
-
5.3
-
V/µs
Maximum Output Voltage
RL = 10K to ground, Load = 400µA
3.6
4.1
-
V
Minimum Output Voltage
RL = 10K to ground, Load = -400µA
-
0.16
0.5
V
Full Scale Input Current
-
50
-
µA
Over-Current Trip Level
-
82.5
-
µA
CHANNEL GENERATOR
ERROR AMPLIFIER
ISEN
POWER GOOD MONITOR
Under-Voltage Threshold
VSEN Rising
-
0.92
-
VDAC
Under-Voltage Threshold
VSEN Falling
-
0.90
-
VDAC
PGOOD Low Output Voltage
IPGOOD = 4mA
-
0.18
0.4
V
1.12
1.15
1.2
VDAC
-
2
-
%
PROTECTION
Over-Voltage Threshold
VSEN Rising
Percent Over-Voltage Hysteresis
VSEN Falling after Over-Voltage
5
HIP6304
Operation
Figure 1 shows a simplified diagram of the voltage regulation
and current control loops. Both voltage and current feedback
are used to precisely regulate voltage and tightly control
output currents, IL1 and IL2, of the two power channels. The
voltage loop comprises the Error Amplifier, Comparators,
gate drivers and output MOSFETs. The Error Amplifier is
essentially connected as a voltage follower that has as an
input, the Programmable Reference DAC and an output that
is the CORE voltage.
Voltage Loop
Feedback from the CORE voltage is applied via resistor RIN
to the inverting input of the Error Amplifier. This signal can
drive the Error Amplifier output either high or low, depending
upon the CORE voltage. Low CORE voltage makes the
amplifier output move towards a higher output voltage level.
Amplifier output voltage is applied to the positive inputs of
the Comparators via the Correction summing networks. Outof-phase sawtooth signals are applied to the two
Comparators inverting inputs. Increasing Error Amplifier
voltage results in increased Comparator output duty cycle.
This increased duty cycle signal is passed through the PWM
CIRCUIT with no phase reversal and on to the HIP6601,
again with no phase reversal for gate drive to the upper
MOSFETs, Q1 and Q3. Increased duty cycle or ON time for
the MOSFET transistors results in increased output voltage
to compensate for the low output voltage sensed.
Current Loop
The current control loop works in a similar fashion to the
voltage control loop, but with current control information applied
individually to each channel’s Comparator. The information
used for this control is the voltage that is developed across
rDS(ON) of each lower MOSFET, Q2 and Q4, when they are
conducting. A single resistor converts and scales the voltage
across the MOSFETs to a current that is applied to the Current
Sensing circuit within the HIP6304. Output from these sensing
circuits is applied to the current averaging circuit. Each PWM
channel receives the difference current signal from the
summing circuit that compares the average sensed current to
the individual channel current. When a power channel’s current
is greater than the average current, the signal applied via the
summing Correction circuit to the Comparator, reduces the
output pulse width of the Comparator to compensate for the
detected “above average” current in that channel.
Droop Compensation
In addition to control of each power channel’s output current,
the average channel current is also used to provide CORE
voltage “droop” compensation. Average full channel current
is defined as 50µA. By selecting an input resistor, RIN, the
amount of voltage droop required at full load current can be
programmed. The average current driven into the FB pin
results in a voltage increase across resistor RIN that is in the
direction to make the Error Amplifier “see” a higher voltage
6
at the inverting input, resulting in the Error Amplifier
adjusting the output voltage lower. The voltage developed
across RIN is equal to the “droop” voltage. See the “Current
Sensing and Balancing” section for more details.
Applications and Convertor Start-Up
Each PWM power channel’s current is regulated. This
enables the PWM channels to accurately share the load
current for enhanced reliability. The HIP6601, HIP6602 or
HIP6603 MOSFET driver interfaces with the HIP6304. For
more information, see the HIP6601, HIP6602 or HIP6603
data sheets.
The HIP6304 controls the two PWM power channels 180
degrees out of phase. Figure 2 shows the out of phase
relationship between the two PWM channels.
PWM 1
PWM 2
FIGURE 2. TWO PHASE PWM OUTPUT AT 500kHz
Power supply ripple frequency is determined by the channel
frequency, FSW, multiplied by the number of active channels.
For example, if the channel frequency is set to 250kHz, the
ripple frequency is 500kHz.
The IC monitors and precisely regulates the CORE voltage
of a microprocessor. After initial start-up, the controller also
provides protection for the load and the power supply. The
following section discusses these features.
Initialization
The HIP6304 usually operates from an ATX power supply.
Many functions are initiated by the rising supply voltage to the
VCC pin of the HIP6304. Oscillator, Sawtooth Generator, SoftStart and other functions are initialized during this interval.
These circuits are controlled by POR, Power-On Reset. During
this interval, the PWM outputs are driven to a three state
condition that makes these outputs essentially open. This state
results in no gate drive to the output MOSFETs.
Once the VCC voltage reaches 4.375V (+125mV), a voltage
level to insure proper internal function, the PWM outputs are
enabled and the Soft-Start sequence is initiated. If for any
reason, the VCC voltage drops below 3.875V (+125mV). The
POR circuit shuts the converter down and again three states
the PWM outputs.
Soft-Start
After the POR function is completed with VCC reaching
4.375V, the Soft-Start sequence is initiated. Soft-Start, by its
slow rise in CORE voltage from zero, avoids an over-current
HIP6304
condition by slowly charging the discharged output
capacitors. This voltage rise is initiated by an internal DAC
that slowly raises the reference voltage to the error amplifier
input. The voltage rise is controlled by the oscillator
frequency and the DAC within the HIP6304, therefore, the
output voltage is effectively regulated as it rises to the final
programmed CORE voltage value.
PWM 1
OUTPUT
DELAY TIME
PGOOD
For the first 32 PWM switching cycles, the DAC output
remains inhibited and the PWM outputs remain three stated.
From the 33rd cycle and for another, approximately 150
cycles the PWM output remains low, clamping the lower
output MOSFETs to ground, see Figure 3. The time
variability is due to the Error Amplifier, Sawtooth Generator
and Comparators moving into their active regions. After this
short interval, the PWM outputs are enabled and increment
the PWM pulse width from zero duty cycle to operational
pulse width, thus allowing the output voltage to slowly reach
the CORE voltage. The CORE voltage will reach its
programmed value before the 2048 cycles, but the PGOOD
output will not be initiated until the 2048th PWM switching
cycle.
VCORE
5V
VCC
VIN = 12V
FIGURE 3. START-UP OF A SYSTEM OPERATING AT 500kHz
V COMP
The Soft-Start time or delay time, DT = 2048 / FSW. For an
oscillator frequency, FSW, of 200kHz, the first 32 cycles or
160µs, the PWM outputs are held in a three state level as
explained above. After this period and a short interval
described above, the PWM outputs are initiated and the
voltage rises in 10.08ms, for a total delay time DT of
10.24ms.
DELAY TIME
PGOOD
VCORE
Figure 3 shows the start-up sequence as initiated by a fast
rising 5V supply, VCC, applied to the HIP6304. Note the
short rise to the three state level in PWM 1 output during first
32 PWM cycles.
Figure 4 shows the waveforms when the regulator is
operating at 200kHz. Note that the Soft-Start duration is a
function of the Channel Frequency as explained previously.
Also note the pulses on the COMP terminal. These pulses
are the current correction signal feeding into the comparator
input (see the Block Diagram on page 2).
5V
VCC
VIN = 12V
FIGURE 4. START-UP A SYSTEM OPERATING AT 200kHz
.
12V ATX
SUPPLY
Figure 5 shows the regulator operating from an ATX supply.
In this figure, note the slight rise in PGOOD as the 5V supply
rises. The PGOOD output stage is made up of NMOS and
PMOS transistors. On the rising VCC, the PMOS device
becomes active slightly before the NMOS transistor pulls
“down”, generating the slight rise in the PGOOD voltage.
Note that Figure 5 shows the 12V gate driver voltage
available before the 5V supply to the HIP6304 has reached
its threshold level. If conditions were reversed and the 5V
supply was to rise first, the start-up sequence would be
different. In this case the HIP6303 will sense an over-current
condition due to charging the output capacitors. The supply
will then restart and go through the normal Soft-Start cycle.
PGOOD
VCORE
5 V ATX
SUPPLY
VIN = 5V, CORE LOAD CURRENT = 31A
FREQUENCY 200kHz
ATX SUPPLY ACTIVATED BY ATX “PS-ON PIN”
FIGURE 5. SUPPLY POWERED BY ATX SUPPLY
7
HIP6304
Fault Protection
The HIP6304 protects the microprocessor and the entire
power system from damaging stress levels. Within the
HIP6304 both Over-Voltage and Over-Current circuits are
incorporated to protect the load and regulator.
Over-Voltage
The VSEN pin is connected to the microprocessor CORE
voltage. A CORE over-voltage condition is detected when
the VSEN pin goes more than 15% above the programmed
VID level.
The over-voltage condition is latched, disabling normal PWM
operation, and causing PGOOD to go low. The latch can
only be reset by lowering and returning VCC high to initiate a
POR and Soft-Start sequence.
During a latched over-voltage, the PWM outputs will be
driven either low or three state, depending upon the VSEN
input. PWM outputs are driven low when the VSEN pin
detects that the CORE voltage is 15% above the
programmed VID level. This condition drives the PWM
outputs low, resulting in the lower or synchronous rectifier
MOSFETs to conduct and shunt the CORE voltage to
ground to protect the load.
If after this event, the CORE voltage falls below the overvoltage limit (plus some hysteresis), the PWM outputs will
three state. The HIP6601 family drivers pass the three state
information along, and shuts off both upper and lower
MOSFETs. This prevents “dumping” of the output capacitors
back through the lower MOSFETs, avoiding a possibly
destructive ringing of the capacitors and output inductors. If
the conditions that caused the over-voltage still persist, the
PWM outputs will be cycled between three state and VCORE
clamped to ground, as a hysteretic shunt regulator.
Under-Voltage
The VSEN pin also detects when the CORE voltage falls
more than 10% below the VID programmed level. This
causes PGOOD to go low, but has no other effect on
operation and is not latched. There is also hysteresis in this
detection point.
over-current detection would cause a dead time of 10.24ms,
then a ramp of 10.08ms.
At the end of the delay, PWM outputs are restarted and the
Soft-Start ramp is initiated. If a short is present at that time,
the cycle is repeated. This is the hiccup mode.
Figure 6 shows the supply shorted under operation and the
hiccup operating mode described above. Note that due to
the high short circuit current, over-current is detected before
completion of the start-up sequence so the delay is not quite
as long as the normal Soft-Start cycle.
SHORT APPLIED HERE
PGOOD
SHORT
CURRENT
50A/DIV
HICCUP MODE. SUPPLY POWERED BY ATX SUPPLY
CORE LOAD CURRENT = 31A, 5V LOAD = 5A
SUPPLY FREQUENCY = 200kHz, V IN = 12V
ATX SUPPLY ACTIVATED BY ATX “PS-ON PIN”
FIGURE 6. SHORT APPLIED TO SUPPLY AFTER POWER-UP
CORE Voltage Programming
The voltage identification pins (VID0, VID1, VID2, and VID3)
set the CORE output voltage. Each VID pin is pulled to VCC
by an internal 20µA current source and accepts
open-collector/open-drain/open-switch-to-ground or
standard low-voltage TTL or CMOS signals.
Table 1 shows the nominal DAC voltage as a function of the
VID codes. The power supply system is ±1% accurate over
the operating temperature and voltage range.
TABLE 1. VOLTAGE IDENTIFICATION CODES
VID3
VID2
VID1
VID0
VDAC
1
1
1
1
1.30
1
1
1
0
1.35
1
1
0
1
1.40
Over-Current
1
1
0
0
1.45
In the event of an over-current condition, the over-current
protection circuit reduces the average current delivered to
less than 25% of the current limit. When an over-current
condition is detected, the controller forces all PWM outputs
into a three state mode. This condition results in the gate
driver removing drive to the output stages. The HIP6304
goes into a wait delay timing cycle that is equal to the SoftStart ramp time. PGOOD also goes “low” during this time
due to VSEN going below its threshold voltage. To lower the
average output dissipation, the Soft-Start initial wait time is
increased from 32 to 2048 cycles, then the Soft-Start ramp is
initiated. At a PWM frequency of 200kHz, for instance, an
1
0
1
1
1.50
1
0
1
0
1.55
1
0
0
1
1.60
1
0
0
0
1.65
0
1
1
1
1.70
0
1
1
0
1.75
0
1
0
1
1.80
0
1
0
0
1.85
0
0
1
1
1.90
0
0
1
0
1.95
0
0
0
1
2.00
0
0
0
0
2.05
8
HIP6304
Current Sensing and Balancing
over-current detection is 165% of that value, or 82.5µA.
Therefore, RISEN = IL x rDS(ON) (Q2) / 50µA.
Overview
For a full load of 25A per phase, and an rDS(ON) (Q2) of
4mΩ, RISEN = 2kΩ .
The HIP6304 samples the on-state voltage drop across each
synchronous rectifier FET, Q2, as an indication of the
inductor current in that phase, see Figure 7. Neglecting AC
effects (to be discussed later), the voltage drop across Q2 is
simply rDS(ON) (Q2) x inductor current (IL). Note that IL, the
inductor current, is 1/2 of the total current (ILT).
The over-current trip point would be 165% of 25A, or ~ 41A
per phase. The RISEN value can be adjusted to change the
over-current trip point, but it is suggested to stay within ±25%
of nominal.
The voltage at Q2’s drain, the PHASE node, is applied to the
RISEN resistor to develop the IISEN current to the HIP6304
ISEN pin. This pin is held at virtual ground, so the current
through RISEN is IL x rDS(ON)(Q2) / RISEN.
Droop, Selection of RIN
The average of the currents detected through the RISEN
resistors is also steered to the FB pin. There is no DC return
path connected to the FB pin except for RIN, so the average
current creates a voltage drop across RIN. This drop increases
the apparent VCORE voltage with increasing load current,
causing the system to decrease VCORE to maintain balance at
the FB pin. This is the desired “droop” voltage used to maintain
VCORE within limits under transient conditions.
The IISEN current provides information to perform the
following functions:
1. Detection of an over-current condition
2. Reduce the regulator output voltage with increasing load
current (droop)
With a high dv/dt load transient, typical of high performance
microprocessors, the largest deviations in output voltage
occur at the leading and trailing edges of the load transient. In
order to fully utilize the output-voltage tolerance range, the
output voltage is positioned in the upper half of the range
when the output is unloaded and in the lower half of the range
when the controller is under full load. This droop
compensation allows larger transient voltage deviations and
thus reduces the size and cost of the output filter components.
3. Balance the IL currents in the two phases
Over-Current, Selecting RISEN
The current detected through the RISEN resistor is averaged
with the current detected in the other channel. The averaged
current is compared with a trimmed, internally generated
current, and used to detect an over-current condition.
The nominal current through the RISEN resistor should be
50µA at full output load current, and the nominal trip point for
RIN
Cc
VIN
COMP
SAWTOOTH
ERROR
AMPLIFIER
COMPARATOR
GENERATOR
+
CORRECTION
+
-
HIP6304
L01
Q1
PWM
CIRCUIT
VCORE
HIP6601
PWM
IL
Q2
+
-
PHASE
DIFFERENCE
+
REFERENCE
DAC
CURRENT
ISEN
RISEN
SENSING
CURRENT
SENSING
FROM
OTHER
CHANNEL
TO OTHER
CHANNEL
AVERAGING
TO OVER
CURRENT
TRIP
+
ONLY ONE OUTPUT
STAGE SHOWN
INDUCTOR
CURRENT
FROM
OTHER
CHANNEL
COMPARATOR
REFERENCE
FIGURE 7. SIMPLIFIED FUNCTIONAL BLOCK DIAGRAM SHOWING CURRENT AND VOLTAGE SAMPLING
9
RLOAD
RFB
COUT
FB
HIP6304
RIN should be selected to give the desired “droop” voltage at
the normal full load current 50µA applied through the RISEN
resistor (or at a different full load current if adjusted as under
“Over-Current, Selecting RISEN” above).
25
For a Vdroop of 80mV, RIN = 1.6kΩ
The AC feedback components, RFB and Cc, are scaled in
relation to RIN.
Current Balancing
The detected currents are also used to balance the phase
currents.
Each phase’s current is compared to the average of the two
phase currents, and the difference is used to create an offset
in that phase’s PWM comparator. The offset is in a direction
to reduce the imbalance.
AMPERES
20
RIN = Vdroop / 50µA
15
10
5
0
FIGURE 8. TWO CHANNEL MULTIPHASE SYSTEM WITH
CURRENT BALANCING DISABLED
The balancing circuit can not make up for a difference in
rDS(ON) between synchronous rectifiers. If a FET has a higher
rDS(ON), the current through that phase will be reduced.
25
Figures 8 and 9 show the inductor current of a two phase
system without and with current balancing.
The inductor current in each phase of a multi-phase Buck
converter has two components. There is a current equal to
the load current divided by the number of phases (ILT / n),
and a sawtooth current, (iPK-PK) resulting from switching.
The sawtooth component is dependent on the size of the
inductors, the switching frequency of each phase, and the
values of the input and output voltage. Ignoring secondary
effects, such as series resistance, the peak to peak value of
the sawtooth current can be described by:
iPK-PK = (VIN x VCORE - VCORE2) / (L x FSW x VIN)
Where: VCORE = DC value of the output or VID voltage
VIN = DC value of the input or supply voltage
L = value of the inductor
FSW = switching frequency
Example: For VCORE = 1.6V,
VIN = 12V,
L = 1.3µH,
FSW = 250kHz,
Then iPK-PK = 4.3A
AMPERES
Inductor Current
20
15
10
5
0
FIGURE 9. TWO CHANNEL MULTIPHASE SYSTEM WITH
CURRENT BALANCING ENABLED
The inductor, or load current, flows alternately from VIN
through Q1 and from ground through Q2. The HIP6304
samples the on-state voltage drop across each Q2 transistor
to indicate the inductor current in that phase. The voltage
drop is sampled 1/3 of a switching period, 1/FSW, after Q1 is
turned OFF and Q2 is turned on. Because of the sawtooth
current component, the sampled current is different from the
average current per phase. Neglecting secondary effects,
the sampled current (ISAMPLE) can be related to the load
current (ILT) by:
ISAMPLE = ILT / n + (VINVCORE - 3VCORE2) / (6L x FSW x VIN)
Where: ILT
10
= total load current
n = the number of channels
HIP6304
Example: Using the previously given conditions, and
For ILT = 50A,
n =2
Then ISAMPLE = 25.49A
As discussed previously, the voltage drop across each Q2
transistor at the point in time when current is sampled is
rDSON (Q2) x ISAMPLE. The voltage at Q2’s drain, the
PHASE node, is applied through the RISEN resistor to the
HIP6304 ISEN pin. This pin is held at virtual ground, so the
current into ISEN is:
ISENSE = ISAMPLE x rDS(ON) (Q2) / RISEN.
= ISAMPLE x rDS(ON) (Q2) / 50µA
RIsen
Example: From the previous conditions,
where ILT
= 50A,
ISAMPLE
= 25.49A,
rDS(ON) (Q2)
= 4mΩ
Then: RISEN
= 2.04K and
ICURRENT TRIP
= 165%
Short circuit ILT
= 82.5A.
Channel Frequency Oscillator
The channel oscillator frequency is set by placing a resistor,
RT, to ground from the FS/DIS pin. Figure 10 is a curve
showing the relationship between frequency, FSW, and
resistor RT. To avoid pickup by the FS/DIS pin, it is important
to place this resistor next to the pin. If this pin is also used to
disable the converter, it is also important to locate the pulldown device next to this pin.
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit
and lead to device over-voltage stress. Careful component
layout and printed circuit design minimizes the voltage
spikes in the converter. Consider, as an example, the turnoff
transition of the upper PWM MOSFET. Prior to turnoff, the
upper MOSFET was carrying channel current. During the
turnoff, current stops flowing in the upper MOSFET and is
picked up by the lower MOSFET. Any inductance in the
switched current path generates a large voltage spike during
the switching interval. Careful component selection, tight
layout of the critical components, and short, wide circuit
traces minimize the magnitude of voltage spikes. Contact
Intersil for evaluation board drawings of the component
placement and printed circuit board.
There are two sets of critical components in a DC-DC
converter using a HIP6304 controller and a HIP6601 gate
driver. The power components are the most critical because
they switch large amounts of energy. Next are small signal
components that connect to sensitive nodes or supply
critical bypassing current and signal coupling.
The power components should be placed first. Locate the
input capacitors close to the power switches. Minimize the
length of the connections between the input capacitors, CIN,
and the power switches. Locate the output inductors and
output capacitors between the MOSFETs and the load.
Locate the gate driver close to the MOSFETs.
The critical small components include the bypass capacitors
for VCC and PVCC on the gate driver ICs. Locate the
bypass capacitor, CBP, for the HIP6304 controller close to
the device. It is especially important to locate the resistors
associated with the input to the amplifiers close to their
respective pins, since they represent the input to feedback
amplifiers. Resistor RT, that sets the oscillator frequency
should also be located next to the associated pin. It is
especially important to place the RSEN resistor(s) at the
respective terminals of the HIP6304.
1,000
500
200
100
50
RT (kΩ)
Layout Considerations
20
10
5
2
1
10
20
50
100
200
500 1,000 2,000 5,000 10,000
CHANNEL OSCILLATOR FREQUENCY, FSW (kHz)
FIGURE 10. RESISTANCE RT vs FREQUENCY
11
A multi-layer printed circuit board is recommended. Figure 11
shows the connections of the critical components for one
output channel of the converter. Note that capacitors CIN and
COUT could each represent numerous physical capacitors.
Dedicate one solid layer, usually the middle layer of the PC
board, for a ground plane and make all critical component
ground connections with vias to this layer. Dedicate another
solid layer as a power plane and break this plane into smaller
islands of common voltage levels. Keep the metal runs from
the PHASE terminal to inductor LO1 short. The power plane
should support the input power and output power nodes. Use
copper filled polygons on the top and bottom circuit layers for
HIP6304
with lower ESR available in larger case sizes. However, the
equivalent series inductance (ESL) of these capacitors
increases with case size and can reduce the usefulness of
the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Consult the
capacitor manufacturer and measure the capacitor’s
impedance with frequency to select a suitable component.
the phase nodes. Use the remaining printed circuit layers for
small signal wiring. The wiring traces from the driver IC to the
MOSFET gate and source should be sized to carry at least
one ampere of current.
Component Selection Guidelines
Output Capacitor Selection
The output capacitor is selected to meet both the dynamic
load requirements and the voltage ripple requirements. The
load transient for the microprocessor CORE is characterized
by high slew rate (di/dt) current demands. In general,
multiple high quality capacitors of different size and dielectric
are paralleled to meet the design constraints.
Output Inductor Selection
Modern microprocessors produce severe transient load rates.
High frequency capacitors supply the initially transient current
and slow the load rate-of-change seen by the bulk capacitors.
The bulk filter capacitor values are generally determined by
the ESR (effective series resistance) and voltage rating
requirements rather than actual capacitance requirements.
The output inductor of each power channel controls the
ripple current. The control IC is stable for channel ripple
current (peak-to-peak) up to twice the average current. A
single channel’s ripple current is approximately:
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Small inductors in a multi-phase converter reduces
the response time without significant increases in total ripple
current.
V IN – V OUT V OUT
∆I = -------------------------------- × ---------------V IN
F SW xL
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
The current from multiple channels tend to cancel each other
and reduce the total ripple current. Figure 12 gives the total
ripple current as a function of duty cycle, normalized to the
parameter ( Vo ) ⁄ ( L ⋅ F S ) at zero duty cycle. To determine the
total ripple current from the number of channels and the duty
cycle, multiply the y-axis value by ( Vo ) ⁄ ( LxF SW ) .
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR determines the output ripple voltage
and the initial voltage drop following a high slew-rate
transient’s edge. In most cases, multiple capacitors of small
case size perform better than a single large case capacitor.
Small values of output inductance can cause excessive power
dissipation. The HIP6303 is designed for stable operation for
ripple currents up to twice the load current. However, for this
condition, the RMS current is 115% above the value shown in
the following MOSFET Selection and Considerations section.
With all else fixed, decreasing the inductance could increase
the power dissipated in the MOSFETs by 30%.
Bulk capacitor choices include aluminum electrolytic, OSCon, Tantalum and even ceramic dielectrics. An aluminum
electrolytic capacitor’s ESR value is related to the case size
+5VIN
USE INDIVIDUAL METAL RUNS
FOR EACH CHANNEL TO HELP
ISOLATE OUTPUT STAGES
+12V
CBP
VCC PVCC
LOCATE NEXT TO IC PIN(S)
CBOOT
VCC
CBP
PWM
HIP6304
RFB
LOCATE NEXT
TO FB PIN
VCORE
PHASE
COUT
RT
FB
LOCATE NEXT TO IC PIN
RSEN
RIN
VSEN
LEGEND
LOCATE NEAR TRANSISTOR
LO1
HIP6601
COMP FS/DIS
CT
CIN
ISEN
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 11. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
12
HIP6304
supply the RMS current. Small ceramic capacitors should
be placed very close to the drain of the upper MOSFET to
suppress the voltage induced in the parasitic circuit
impedances.
SINGLE
CHANNEL
0.8
VO / (LX FSW)
RIPPLE CURRENT (APEAK-PEAK)
1.0
For bulk capacitance, several electrolytic capacitors (Panasonic
HFQ series or Nichicon PL series or Sanyo MV-GX or
equivalent) may be needed. For surface mount designs, solid
tantalum capacitors can be used, but caution must be
exercised with regard to the capacitor surge current rating.
These capacitors must be capable of handling the surgecurrent at power-up. The TPS series available from AVX, and
the 593D series from Sprague are both surge current tested.
0.6
2 CHANNEL
0.4
3 CHANNEL
0.2
4 CHANNEL
0
0
0.1
0.2
0.3
0.4
0.5
DUTY CYCLE (VO/VIN)
FIGURE 12. RIPPLE CURRENT vs DUTY CYCLE
Input Capacitor Selection
The important parameters for the bulk input capacitors are
the voltage rating and the RMS current rating. For reliable
operation, select bulk input capacitors with voltage and
current ratings above the maximum input voltage and
largest RMS current required by the circuit. The capacitor
voltage rating should be at least 1.25 times greater than the
maximum input voltage and a voltage rating of 1.5 times is
a conservative guideline. The RMS current required for a
multi-phase converter can be approximated with the aid of
Figure 13.
CURRENT MULTIPLIER
0.5
SINGLE
CHANNEL
0.4
0.3
2 CHANNEL
0.2
3 CHANNEL
0.1
MOSFET Selection and Considerations
In high-current PWM applications, the MOSFET power
dissipation, package selection and heatsink are the
dominant design factors. The power dissipation includes two
loss components; conduction loss and switching loss. These
losses are distributed between the upper and lower
MOSFETs according to duty factor (see the following
equations). The conduction losses are the main component
of power dissipation for the lower MOSFETs, Q2 and Q4 of
Figure 1. Only the upper MOSFETs, Q1 and Q3 have
significant switching losses, since the lower device turns on
and off into near zero voltage.
The equations assume linear voltage-current transitions and
do not model power loss due to the reverse-recovery of the
lower MOSFETs body diode. The gate-charge losses are
dissipated by the Driver IC and don't heat the MOSFETs.
However, large gate-charge increases the switching time,
tSW which increases the upper MOSFET switching losses.
Ensure that both MOSFETs are within their maximum
junction temperature at high ambient temperature by
calculating the temperature rise according to package
thermal-resistance specifications. A separate heatsink may
be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
2
I O × r DS ( ON ) × V OUT I O × V IN × t SW × F SW
P UPPER = ------------------------------------------------------------ + ---------------------------------------------------------V IN
2
4 CHANNEL
2
0
0
0.1
0.2
0.3
0.4
0.5
DUTY CYCLE (VO/VIN)
FIGURE 13. CURRENT MULTIPLIER vs DUTY CYCLE
First determine the operating duty ratio as the ratio of the
output voltage divided by the input voltage. Find the Current
Multiplier from the curve with the appropriate power
channels. Multiply the current multiplier by the full load
output current. The resulting value is the RMS current rating
required by the input capacitor.
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use ceramic capacitance
for the high frequency decoupling and bulk capacitors to
13
I O × r DS ( ON ) × ( V IN – V OUT )
P LOWER = --------------------------------------------------------------------------------V IN
A diode, anode to ground, may be placed across Q2 and Q4
of Figure 1. These diodes function as a clamp that catches
the negative inductor swing during the dead time between
the turn off of the lower MOSFETs and the turn on of the
upper MOSFETs. The diodes must be a Schottky type to
prevent the lossy parasitic MOSFET body diode from
conducting. It is usually acceptable to omit the diodes and let
the body diodes of the lower MOSFETs clamp the negative
inductor swing, but efficiency could drop one or two percent
as a result. The diode's rated reverse breakdown voltage
must be greater than the maximum input voltage.
HIP6304
Small Outline Plastic Packages (SOIC)
N
INDEX
AREA
H
0.25(0.010) M
M16.15 (JEDEC MS-012-AC ISSUE C)
B M
16 LEAD NARROW BODY SMALL OUTLINE PLASTIC
PACKAGE
E
-B1
2
INCHES
3
L
SEATING PLANE
-A-
h x 45o
A
D
-C-
α
e
A1
B
0.25(0.010) M
C
0.10(0.004)
C A M
B S
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006
inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead
flash and protrusions shall not exceed 0.25mm (0.010 inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above
the seating plane, shall not exceed a maximum value of 0.61mm
(0.024 inch).
10. Controlling dimension: MILLIMETER. Converted inch dimensions are
not necessarily exact.
MILLIMETERS
SYMBOL
MIN
MAX
MIN
MAX
NOTES
A
0.0532
0.0688
1.35
1.75
-
A1
0.0040
0.0098
0.10
0.25
-
B
0.013
0.020
0.33
0.51
9
C
0.0075
0.0098
0.19
0.25
-
D
0.3859
0.3937
9.80
10.00
3
E
0.1497
0.1574
3.80
4.00
4
e
0.050 BSC
1.27 BSC
-
H
0.2284
0.2440
5.80
6.20
-
h
0.0099
0.0196
0.25
0.50
5
L
0.016
0.050
0.40
N
α
16
0o
1.27
6
16
8o
0o
7
8o
Rev. 0 12/93
All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification.
Intersil semiconductor products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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Intersil Corporation
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TEL: (321) 724-7000
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14
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