DATASHEET

HIP6006
®
Data Sheet
April 1, 2005
Buck and Synchronous-Rectifier
Pulse-Width Modulator (PWM) Controller
The HIP6006 provides complete control and protection for a
DC-DC converter optimized for high-performance
microprocessor applications. It is designed to drive two
N Channel MOSFETs in a synchronous-rectified buck
topology. The HIP6006 integrates all of the control, output
adjustment, monitoring and protection functions into a single
package.
The output voltage of the converter can be precisely
regulated to as low as 1.27V, with a maximum tolerance of
±1% over temperature and line voltage variations.
The HIP6006 provides simple, single feedback loop, voltagemode control with fast transient response. It includes a
200kHz free-running triangle-wave oscillator that is
adjustable from below 50kHz to over 1MHz. The error
amplifier features a 15MHz gain-bandwidth product and
6V/µs slew rate which enables high converter bandwidth for
fast transient performance. The resulting PWM duty ratio
ranges from 0% to 100%.
FN4306.3
Features
• Drives Two N-Channel MOSFETs
• Operates From +5V or +12V Input
• Simple Single-Loop Control Design
- Voltage-Mode PWM Control
• Fast Transient Response
- High-Bandwidth Error Amplifier
- Full 0% to 100% Duty Ratio
• Excellent Output Voltage Regulation
- 1.27V Internal Reference
- ±1% Over Line Voltage and Temperature
• Over-Current Fault Monitor
- Does Not Require Extra Current Sensing Element
- Uses MOSFETs rDS(ON)
• Small Converter Size
- Constant Frequency Operation
- 200kHz Free-Running Oscillator Programmable from
50kHz to Over 1MHz
The HIP6006 protects against over-current conditions by
inhibiting PWM operation. The HIP6006 monitors the current
by using the rDS(ON) of the upper MOSFET which eliminates
the need for a current sensing resistor.
• 14 Pin, SOIC and TSSOP Package
Pinout
• Power Supply for Pentium®, Pentium Pro, PowerPC™
and Alpha™ Microprocessors
HIP6006
(SOIC, TSSOP)
TOP VIEW
• Pb-Free Available (RoHS Compliant)
Applications
• High-Power 5V to 3.xV DC-DC Regulators
• Low-Voltage Distributed Power Supplies
RT
1
14 VCC
OCSET
2
13 PVCC
SS
Ordering Information
TEMP.
RANGE (oC)
PKG.
DWG. #
3
12 LGATE
COMP
4
11 PGND
FB
5
10 BOOT
HIP6006CB
0 to 70
14 Ld SOIC
M14.15
EN
6
9
UGATE
HIP6006CB-T
0 to 70
14 Ld SOIC) (T&R)
M14.15
GND
7
8
PHASE
HIP6006CBZ (Note)
0 to 70
14 Ld SOIC (Pb-free) M14.15
HIP6006CBZ-T
(Note)
0 to 70
14 Ld SOIC) (T&R)
(Pb-free)
M14.15
HIP6006CV
0 to 70
14 Ld TSSOP
M14.173
HIP6006CV-T
0 to 70
14 Ld TSSOP (T&R) M14.173
PART NUMBER
PACKAGE
NOTE: Intersil Pb-free products employ special Pb-free material sets;
molding compounds/die attach materials and 100% matte tin plate
termination finish, which are RoHS compliant and compatible with both
SnPb and Pb-free soldering operations. Intersil Pb-free products are
MSL classified at Pb-free peak reflow temperatures that meet or exceed
the Pb-free requirements of IPC/JEDEC J STD-020.
PowerPC™ is a trademark of IBM.
Alpha™ is a trademark of Digital Equipment Corporation.
Pentium® is a registered trademark of Intel Corporation.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-352-6832 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2001, 2005. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
HIP6006
Typical Application
12V
+5V OR +12V
VCC
SS
OCSET
MONITOR AND
PROTECTION
EN
BOOT
RT
OSC
UGATE
PHASE
HIP6006
REF
+
FB
-
-
+VO
+12V
PVCC
LGATE
+
PGND
GND
COMP
Block Diagram
VCC
POWER-ON
RESET (POR)
EN
10µA
+
-
OCSET
OVERCURRENT
SOFTSTART
SS
BOOT
4V
200µA
UGATE
PHASE
REFERENCE
PWM
COMPARATOR
1.27 VREF
+
-
+
-
ERROR
AMP
FB
INHIBIT
PWM
GATE
CONTROL
LOGIC
PVCC
LGATE
PGND
COMP
GND
OSCILLATOR
RT
2
HIP6006
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +15.0V
Boot Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . . . . +15.0V
Input, Output or I/O Voltage . . . . . . . . . . . . GND -0.3V to VCC +0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
Thermal Resistance (Typical, Note 1)
θJA (oC/W)
SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
85
TSSOP Package . . . . . . . . . . . . . . . . . . . . . . . . . . .
95
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150oC
Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC
(Lead tips only)
Operating Conditions
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10%
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . 0oC to 70oC
Junction Temperature Range. . . . . . . . . . . . . . . . . . . . 0oC to 125oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
EN = VCC; UGATE and LGATE Open
-
5
-
mA
EN = 0V
-
50
100
µA
Rising VCC Threshold
VOCSET = 4.5VDC
-
-
10.4
V
Falling VCC Threshold
VOCSET = 4.5VDC
8.2
-
-
V
Enable - Input threshold Voltage
VOCSET = 4.5VDC
0.8
-
2.0
V
-
1.27
-
V
VCC SUPPLY CURRENT
Nominal Supply
ICC
Shutdown Supply
POWER-ON RESET
Rising VOCSET Threshold
OSCILLATOR
Free Running Frequency
RT = OPEN, VCC = 12
185
200
215
kHz
Total Variation
6kΩ < RT to GND < 200kΩ
-15
-
+15
%
-
1.9
-
VP-P
1.258
1.270
1.282
V
-
88
-
dB
-
15
-
MHz
-
6
-
V/µs
350
500
-
mA
-
5.5
10
Ω
300
450
-
mA
-
3.5
6.5
Ω
170
200
230
µA
-
10
-
µA
∆VOSC
Ramp Amplitude
RT = OPEN
REFERENCE
Reference Voltage
ERROR AMPLIFIER
DC Gain
Gain-Bandwidth Product
GBW
Slew Rate
SR
COMP = 10pF
GATE DRIVERS
Upper Gate Source
IUGATE
VBOOT - VPHASE = 12V, VUGATE = 6V
Upper Gate Sink
RUGATE
ILGATE = 0.3A
Lower Gate Source
ILGATE
VCC = 12V, VLGATE = 6V
Lower Gate Sink
RLGATE
ILGATE = 0.3A
IOCSET
VOCSET = 4.5VDC
PROTECTION
OCSET Current Source
Soft Start Current
ISS
3
HIP6006
Typical Performance Curves
80
70
RT PULLUP
TO +12V
60
IVCC (mA)
RESISTANCE (kΩ)
1000
100
RT PULLDOWN
TO VSS
CGATE = 3300pF
50
40
CGATE = 1000pF
30
20
10
CGATE = 10pF
10
10
100
SWITCHING FREQUENCY (kHz)
1000
FIGURE 1. RT RESISTANCE vs FREQUENCY
Functional Pin Description
0
100
200
300 400 500 600 700 800
SWITCHING FREQUENCY (kHz)
900
1000
FIGURE 2. BIAS SUPPLY CURRENT vs FREQUENCY
SS (Pin 3)
Connect a capacitor from this pin to ground. This capacitor,
along with an internal 10µA current source, sets the soft-start
interval of the converter.
RT
1
14 VCC
OCSET
2
13 PVCC
SS
3
12 LGATE
COMP (Pin 4) and FB (Pin 5)
COMP
4
11 PGND
FB
5
10 BOOT
EN
6
9
UGATE
GND
7
8
PHASE
COMP and FB are the available external pins of the error
amplifier. The FB pin is the inverting input of the error
amplifier and the COMP pin is the error amplifier output.
These pins are used to compensate the voltage-control
feedback loop of the converter.
RT (Pin 1)
This pin provides oscillator switching frequency adjustment.
By placing a resistor (RT) from this pin to GND, the nominal
200kHz switching frequency is increased according to the
following equation:
(RT to GND)
Conversely, connecting a pull-up resistor (RT) from this pin
to VCC reduces the switching frequency according to the
following equation.:
7
4 • 10
Fs ≈ 200kHz – --------------------R T ( kΩ )
This pin is the open-collector enable pin. Pull this pin below
1V to disable the converter. In shutdown, the soft start pin is
discharged and the UGATE and LGATE pins are held low.
GND (Pin 7)
6
5 • 10
Fs ≈ 200kHz + --------------------R T ( kΩ )
EN (Pin 6)
(RT to 12V)
Signal ground for the IC. All voltage levels are measured
with respect to this pin.
PHASE (Pin 8)
Connect the PHASE pin to the upper MOSFET source. This
pin is used to monitor the voltage drop across the MOSFET
for over-current protection. This pin also provides the return
path for the upper gate drive.
UGATE (Pin 9)
OCSET (Pin 2)
Connect a resistor (ROCSET) from this pin to the drain of the
upper MOSFET. ROCSET, an internal 200µA current source
(IOCS), and the upper MOSFET on-resistance (rDS(ON)) set
the converter over-current (OC) trip point according to the
following equation:
I OCS • R OCSET
I PEAK = -------------------------------------------r DS ( ON )
Connect UGATE to the upper MOSFET gate. This pin
provides the gate drive for the upper MOSFET.
BOOT (Pin 10)
This pin provides bias voltage to the upper MOSFET driver.
A bootstrap circuit may be used to create a BOOT voltage
suitable to drive a standard N-Channel MOSFET.
PGND (Pin 11)
An over-current trip cycles the soft-start function.
4
This is the power ground connection. Tie the lower MOSFET
source to this pin.
HIP6006
LGATE (Pin 12)
Connect LGATE to the lower MOSFET gate. This pin
provides the gate drive for the lower MOSFET.
PVCC (Pin 13)
SOFT-START
(1V/DIV)
Provide a bias supply for the lower gate drive to this pin.
VCC (Pin 14)
Provide a 12V bias supply for the chip to this pin.
Functional Description
0V
Initialization
0V
The Power-On Reset (POR) function inhibits operation with
the chip disabled (EN pin low). With both input supplies
above their POR thresholds, transitioning the EN pin high
initiates a soft start interval.
t1
t2
t3
TIME (5ms/DIV)
SOFT-START
FIGURE 3. SOFT-START INTERVAL
OUTPUT INDUCTOR
The HIP6006 automatically initializes upon receipt of power.
Special sequencing of the input supplies is not necessary.
The Power-On Reset (POR) function continually monitors
the input supply voltages and the enable (EN) pin. The POR
monitors the bias voltage at the VCC pin and the input
voltage (VIN) on the OCSET pin. The level on OCSET is
equal to VIN Less a fixed voltage drop (see over-current
protection). With the EN pin held to VCC, the POR function
initiates soft start operation after both input supply voltages
exceed their POR thresholds. For operation with a single
+12V power source, VIN and VCC are equivalent and the
+12V power source must exceed the rising VCC threshold
before POR initiates operation.
OUTPUT
VOLTAGE
(1V/DIV)
4V
2V
0V
15A
10A
5A
0A
Soft Start
The POR function initiates the soft start sequence. An internal
10µA current source charges an external capacitor (CSS) on
the SS pin to 4V. Soft start clamps the error amplifier output
(COMP pin) and reference input (+ terminal of error amp) to
the SS pin voltage. Figure 3 shows the soft start interval with
CSS = 0.1µF. Initially the clamp on the error amplifier (COMP
pin) controls the converter’s output voltage. At t1 in Figure 3,
the SS voltage reaches the valley of the oscillator’s triangle
wave. The oscillator’s triangular waveform is compared to the
ramping error amplifier voltage. This generates PHASE
pulses of increasing width that charge the output capacitor(s).
This interval of increasing pulse width continues to t2. With
sufficient output voltage, the clamp on the reference input
controls the output voltage. This is the interval between t2 and
t3 in Figure 3. At t3 the SS voltage exceeds the reference
voltage and the output voltage is in regulation. This method
provides a rapid and controlled output voltage rise.
5
TIME (20ms/DIV)
FIGURE 4. OVER-CURRENT OPERATION
Over-Current Protection
The over-current function protects the converter from a
shorted output by using the upper MOSFETs on-resistance,
rDS(ON) to monitor the current. This method enhances the
converter’s efficiency and reduces cost by eliminating a
current sensing resistor.
The over-current function cycles the soft-start function in a
hiccup mode to provide fault protection. A resistor (ROCSET)
programs the over-current trip level. An internal 200µA
(typical) current sink develops a voltage across ROCSET that
is reference to VIN. When the voltage across the upper
MOSFET (also referenced to VIN) exceeds the voltage
across ROCSET, the over-current function initiates a softstart sequence. The soft-start function discharges CSS with
a 10µA current sink and inhibits PWM operation. The softstart function recharges CSS, and PWM operation resumes
with the error amplifier clamped to the SS voltage. Should an
overload occur while recharging CSS, the soft start function
inhibits PWM operation while fully charging CSS to 4V to
HIP6006
VIN
HIP6006
UGATE
The over-current function will trip at a peak inductor current
(IPEAK) determined by:
Q1
LO
VOUT
PHASE
I OCSET • R OCSET
I PEAK = --------------------------------------------------r DS ( ON )
Q2
LGATE
D2
CIN
LOAD
complete its cycle. Figure 4 shows this operation with an
overload condition. Note that the inductor current increases
to over 15A during the CSS charging interval and causes an
over-current trip. The converter dissipates very little power
with this method. The measured input power for the
conditions of Figure 4 is 2.5W.
CO
PGND
1. The maximum rDS(ON) at the highest junction
temperature.
2. The minimum IOCSET from the specification table.
3. Determine I PEAK for I PEAK > I OUT ( MAX ) + ( ∆I ) ⁄ 2
where ∆I is the output inductor ripple current.
,
RETURN
FIGURE 5. PRINTED CIRCUIT BOARD POWER AND
GROUND PLANES OR ISLANDS
current paths on the SS PIN and locate the capacitor, Css
close to the SS pin because the internal current source is
only 10µA. Provide local VCC decoupling between VCC and
GND pins. Locate the capacitor, CBOOT as close as practical
to the BOOT and PHASE pins.
A small ceramic capacitor should be placed in parallel with
ROCSET to smooth the voltage across ROCSET in the
presence of switching noise on the input voltage.
+VIN
BOOT
For an equation for the ripple current see the section under
component guidelines titled ‘Output Inductor Selection’.
D1
CBOOT
HIP6006
Q1
LO
VOUT
PHASE
SS
+12V
Q2
CO
LOAD
where IOCSET is the internal OCSET current source (200µA
- typical). The OC trip point varies mainly due to the
MOSFETs rDS(ON) variations. To avoid over-current tripping
in the normal operating load range, find the ROCSET resistor
from the equation above with:
VCC
Application Guidelines
Layout Considerations
CVCC
CSS
GND
As in any high frequency switching converter, layout is very
important. Switching current from one power device to
another can generate voltage transients across the
impedances of the interconnecting bond wires and circuit
traces. These interconnecting impedances should be
minimized by using wide, short printed circuit traces. The
critical components should be located as close together as
possible using ground plane construction or single point
grounding.
Figure 5 shows the critical power components of the
converter. To minimize the voltage overshoot the
interconnecting wires indicated by heavy lines should be part
of ground or power plane in a printed circuit board. The
components shown in Figure 6 should be located as close
together as possible. Please note that the capacitors CIN
and CO each represent numerous physical capacitors.
Locate the HIP6006 within 3 inches of the MOSFETs, Q1
and Q2. The circuit traces for the MOSFETs’ gate and
source connections from the HIP6006 must be sized to
handle up to 1A peak current.
Figure 6 shows the circuit traces that require additional
layout consideration. Use single point and ground plane
construction for the circuits shown. Minimize any leakage
6
FIGURE 6. PRINTED CIRCUIT BOARD SMALL SIGNAL
LAYOUT GUIDELINES
Feedback Compensation
Figure 7 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(Vout) is regulated to the Reference voltage level. The error
amplifier (Error Amp) output (VE/A) is compared with the
oscillator (OSC) triangular wave to provide a pulse-width
modulated (PWM) wave with an amplitude of VIN at the
PHASE node. The PWM wave is smoothed by the output filter
(LO and CO).
The modulator transfer function is the small-signal transfer
function of Vout/VE/A. This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR. The DC Gain of
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage ∆VOSC.
HIP6006
1. Pick Gain (R2/R1) for desired converter bandwidth
2. Place 1ST Zero Below Filter’s Double Pole
(~75% FLC)
VIN
OSC
DRIVER
PWM
COMPARATOR
LO
-
DRIVER
+
∆VOSC
VOUT
PHASE
CO
ESR
(PARASITIC)
ZFB
VE/A
-
ZIN
+
REFERENCE
ERROR
AMP
DETAILED COMPENSATION COMPONENTS
ZFB
C2
C1
VOUT
ZIN
C3
R2
R3
R1
COMP
3. Place 2ND Zero at Filter’s Double Pole
4. Place 1ST Pole at the ESR Zero
5. Place 2ND Pole at Half the Switching Frequency
6. Check Gain against Error Amplifier’s Open-Loop Gain
7. Estimate Phase Margin - Repeat if Necessary
Figure 8 shows an asymptotic plot of the DC-DC converter’s
gain vs frequency. The actual Modulator Gain has a high
gain peak do to the high Q factor of the output filter and is
not shown in Figure 8. Using the above guidelines should
give a Compensation Gain similar to the curve plotted. The
open loop error amplifier gain bounds the compensation
gain. Check the compensation gain at FP2 with the
capabilities of the error amplifier. The Closed Loop Gain is
constructed on the log-log graph of Figure 8 by adding the
Modulator Gain (in dB) to the Compensation Gain (in dB).
This is equivalent to multiplying the modulator transfer
function to the compensation transfer function and plotting
the gain.
FB
-
100
+
REF
FIGURE 7. VOLTAGE - MODE BUCK CONVERTER
COMPENSATION DESIGN
Modulator Break Frequency Equations
1
F LC = --------------------------------------2π • L O • C O
FP1
FP2
1
F ESR = --------------------------------------------2π • ( ESR • C O )
The compensation network consists of the error amplifier
(internal to the HIP6006) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180o. The equations below relate the compensation
network’s poles, zeros and gain to the components (R1, R2,
R3, C1, C2, and C3) in Figure 8. Use these guidelines for
locating the poles and zeros of the compensation network:
Compensation Break Frequency Equations
1
F Z1 = ---------------------------------2π • R 2 • C1
1
F P1 = ------------------------------------------------------C1 • C2
2π • R2 •  ----------------------
 C1 + C2
1
F Z2 = -----------------------------------------------------2π • ( R1 + R3 ) • C3
1
F P2 = ---------------------------------2π • R3 • C3
7
OPEN LOOP
ERROR AMP GAIN
60
GAIN (dB)
HIP6006
FZ1 FZ2
80
40
20
20LOG
(R2/R1)
20LOG
(VIN/∆VOSC)
0
-40
-60
COMPENSATION
GAIN
MODULATOR
GAIN
-20
CLOSED LOOP
GAIN
FLC
10
100
1K
FESR
10K
100K
1M
10M
FREQUENCY (Hz)
FIGURE 8. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW) overall
loop. A stable control loop has a gain crossing with 20dB/decade slope and a phase margin greater than 45o.
Include worst case component variations when determining
phase margin.
Component Selection Guidelines
Output Capacitor Selection
An output capacitor is required to filter the output and supply
the load transient current. The filtering requirements are a
function of the switching frequency and the ripple current.
The load transient requirements are a function of the slew
rate (di/dt) and the magnitude of the transient load current.
These requirements are generally met with a mix of
capacitors and careful layout.
HIP6006
Modern microprocessors produce transient load rates above
1A/ns. High frequency capacitors initially supply the transient
and slow the current load rate seen by the bulk capacitors.
The bulk filter capacitor values are generally determined by
the ESR (effective series resistance) and voltage rating
requirements rather than actual capacitance requirements.
Minimizing the response time can minimize the output
capacitance required.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements. For example, Intel
recommends that the high frequency decoupling for the
Pentium Pro be composed of at least forty (40) 1.0µF
ceramic capacitors in the 1206 surface-mount package.
L O × I TRAN
t RISE = -------------------------------V IN – V OUT
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors.
The bulk capacitor’s ESR will determine the output ripple
voltage and the initial voltage drop after a high slew-rate
transient. An aluminum electrolytic capacitor's ESR value is
related to the case size with lower ESR available in larger
case sizes. However, the equivalent series inductance
(ESL) of these capacitors increases with case size and can
reduce the usefulness of the capacitor to high slew-rate
transient loading. Unfortunately, ESL is not a specified
parameter. Work with your capacitor supplier and measure
the capacitor’s impedance with frequency to select a
suitable component. In most cases, multiple electrolytic
capacitors of small case size perform better than a single
large case capacitor.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
V IN - V OUT V OUT
∆I = -------------------------------- • ---------------Fs x L
V IN
∆VOUT= ∆I x ESR
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
HIP6006 will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
8
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
L O × I TRAN
t FALL = ------------------------------V OUT
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. With a +5V input
source, the worst case response time can be either at the
application or removal of load and dependent upon the
output voltage setting. Be sure to check both of these
equations at the minimum and maximum output levels for
the worst case response time.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time Q1 turns on. Place
the small ceramic capacitors physically close to the
MOSFETs and between the drain of Q1 and the source of
Q2.
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the maximum
input voltage and a voltage rating of 1.5 times is a
conservative guideline. The RMS current rating requirement
for the input capacitor of a buck regulator is approximately
1/2 the DC load current.
For a through hole design, several electrolytic capacitors
(Panasonic HFQ series or Nichicon PL series or Sanyo MVGX or equivalent) may be needed. For surface mount
designs, solid tantalum capacitors can be used, but caution
must be exercised with regard to the capacitor surge current
rating. These capacitors must be capable of handling the
surge-current at power-up. The TPS series available from
AVX, and the 593D series from Sprague are both surge
current tested.
MOSFET Selection/Considerations
The HIP6006 requires 2 N-Channel power MOSFETs. These
should be selected based upon rDS(ON), gate supply
requirements, and thermal management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss
components; conduction loss and switching loss. The
HIP6006
conduction losses are the largest component of power
dissipation for both the upper and the lower MOSFETs.
These losses are distributed between the two MOSFETs
according to duty factor (see the equations below). Only the
upper MOSFET has switching losses, since the Schottky
rectifier clamps the switching node before the synchronous
rectifier turns on.
+12V
DBOOT
+
VCC
These equations assume linear voltage-current transitions
and do not adequately model power loss due the reverserecovery of the lower MOSFETs body diode. The
gate-charge losses are dissipated by the HIP6006 and don't
heat the MOSFETs. However, large gate-charge increases
the switching interval, tSW which increases the upper
MOSFET switching losses. Ensure that both MOSFETs are
within their maximum junction temperature at high ambient
temperature by calculating the temperature rise according to
package thermal-resistance specifications. A separate
heatsink may be necessary depending upon MOSFET
power, package type, ambient temperature and air flow.
Standard-gate MOSFETs are normally recommended for
use with the HIP6006. However, logic-level gate MOSFETs
can be used under special circumstances. The input voltage,
upper gate drive level, and the MOSFETs absolute gate-tosource voltage rating determine whether logic-level
MOSFETs are appropriate.
Figure 9 shows the upper gate drive (BOOT pin) supplied by
a bootstrap circuit from VCC . The boot capacitor, CBOOT
develops a floating supply voltage referenced to the PHASE
pin. This supply is refreshed each cycle to a voltage of VCC
less the boot diode drop (VD) when the lower MOSFET, Q2
turns on. A logic-level MOSFET can only be used for Q1 if
the MOSFETs absolute gate-to-source voltage rating
exceeds the maximum voltage applied to VCC . For Q2, a
logic-level MOSFET can be used if its absolute gate-tosource voltage rating exceeds the maximum voltage applied
to PVCC.
Figure 10 shows the upper gate drive supplied by a direct
connection to VCC . This option should only be used in
converter systems where the main input voltage is +5 VDC
or less. The peak upper gate-to-source voltage is
approximately VCC less the input supply. For +5V main
power and +12 VDC for the bias, the gate-to-source voltage
of Q1 is 7V. A logic-level MOSFET is a good choice for Q1
and a logic-level MOSFET can be used for Q2 if its absolute
gate-to-source voltage rating exceeds the maximum voltage
applied to PVCC.
9
+5V OR +12V
-
BOOT
HIP6006
CBOOT
UGATE
PUPPER = IO2 x rDS(ON) x D + 1 Io x VIN x tSW x Fs
2
2
PLOWER = IO x rDS(ON) x (1 - D)
Where: D is the duty cycle = VO / VIN,
tSW is the switching interval, and
Fs is the switching frequency.
VD
Q1
PHASE
NOTE:
VG-S ≈ VCC - VD
+5V
PVCC OR +12V
LGATE
+
Q2
D2
NOTE:
VG-S ≈ PVCC
PGND
GND
FIGURE 9. UPPER GATE DRIVE - BOOTSTRAP OPTION
+12V
+5V OR LESS
VCC
HIP6006
BOOT
UGATE
Q1
NOTE:
VG-S ≈ VCC - 5V
PHASE
PVCC
+
+5V
OR +12V
LGATE
PGND
Q2
D2
NOTE:
VG-S ≈ PVCC
GND
FIGURE 10. UPPER GATE DRIVE - DIRECT VCC DRIVE OPTION
Schottky Selection
Rectifier D2 is a clamp that catches the negative inductor
swing during the dead time between turning off the lower
MOSFET and turning on the upper MOSFET. The diode must
be a Schottky type to prevent the lossy parasitic MOSFET
body diode from conducting. It is acceptable to omit the diode
and let the body diode of the lower MOSFET clamp the
negative inductor swing, but efficiency will drop one or two
percent as a result. The diode's rated reverse breakdown
voltage must be greater than the maximum input voltage.
HIP6006
HIP6006 DC-DC Converter Application Circuit
Materials and circuit board description, can be found in
application note AN9722. See Intersil’s home page on the
web: www.intersil.com.
The figure below shows an application circuit of a DC-DC
Converter for a microprocessor application. Detailed
information on the circuit, including a complete Bill-of-
12VCC
VIN
C17-18
2x 1µF
1206
C1-3
3x 680µF
RTN
C12
1µF
1206
R7
10k
C19
VCC
6
ENABLE
2 OCSET
MONITOR AND
PROTECTION
SS 3
3.01k
PHASE
TP2
Q1
U1
HIP6006
REF
5
C14
33pF
C15
R5
0.01µF 15k
L1
8 PHASE
VOUT
13 PVCC
Q2
12 LGATE
-+
+
++
--
CR2
MBR
340
11 PGND
4
R2
1k
C20
0.1µF
9 UGATE
OSC
R1
SPARE
FB
R6
10 BOOT
RT 1
C13
0.1µF
CR1
4148
1000pF
14
7
COMP
GND
JP1
COMP
TP1
C16
R3
1k
SPARE
R4
SPARE
Component Selection Notes:
C1-C3 - 3 each 680µF 25W VDC, Sanyo MV-GX or equivalent
C6-C9 - 4 each 1000µF 6.3W VDC, Sanyo MV-GX or equivalent
L1 - Core: Micrometals T50-52B; Winding: 10 Turns of 17AWG
CR1 - 1N4148 or equivalent
CR2 - 3A, 40V Schottky, Motorola MBR340 or equivalent
Q1, Q2 - Intersil MOSFET; RFP25N05
FIGURE 11. DC-DC CONVERTER APPLICATION CIRCUIT
10
C6-9
4x 1000µF
RTN
HIP6006
Thin Shrink Small Outline Plastic Packages (TSSOP)
M14.173
N
INDEX
AREA
E
0.25(0.010) M
E1
2
SYMBOL
3
0.05(0.002)
-A-
INCHES
GAUGE
PLANE
-B1
14 LEAD THIN SHRINK SMALL OUTLINE PLASTIC
PACKAGE
B M
L
A
D
-C-
e
α
A1
b
0.10(0.004) M
0.25
0.010
SEATING PLANE
A2
c
0.10(0.004)
C A M
B S
MIN
1. These package dimensions are within allowable dimensions of
JEDEC MO-153-AC, Issue E.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm
(0.006 inch) per side.
4. Dimension “E1” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.15mm (0.006 inch) per
side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “b” does not include dambar protrusion. Allowable dambar
protrusion shall be 0.08mm (0.003 inch) total in excess of “b” dimension at maximum material condition. Minimum space between protrusion and adjacent lead is 0.07mm (0.0027 inch).
10. Controlling dimension: MILLIMETER. Converted inch dimensions
are not necessarily exact. (Angles in degrees)
11
MILLIMETERS
MIN
MAX
NOTES
A
-
0.047
-
1.20
-
A1
0.002
0.006
0.05
0.15
-
A2
0.031
0.051
0.80
1.05
-
b
0.0075
0.0118
0.19
0.30
9
c
0.0035
0.0079
0.09
0.20
-
D
0.195
0.199
4.95
5.05
3
E1
0.169
0.177
4.30
4.50
4
e
0.026 BSC
0.65 BSC
-
E
0.246
0.256
6.25
6.50
-
L
0.0177
0.0295
0.45
0.75
6
8o
0o
N
NOTES:
MAX
α
14
0o
14
7
8o
Rev. 1 6/00
HIP6006
Small Outline Plastic Packages (SOIC)
M14.15 (JEDEC MS-012-AB ISSUE C)
N
INDEX
AREA
0.25(0.010) M
H
14 LEAD NARROW BODY SMALL OUTLINE PLASTIC
PACKAGE
B M
E
INCHES
-B-
1
2
3
L
SEATING PLANE
-A-
h x 45o
A
D
-C-
µα
e
A1
B
0.25(0.010) M
C A M
SYMBOL
MIN
MAX
MIN
MAX
NOTES
A
0.0532
0.0688
1.35
1.75
-
A1
0.0040
0.0098
0.10
0.25
-
B
0.013
0.020
0.33
0.51
9
C
0.0075
0.0098
0.19
0.25
-
D
0.3367
0.3444
8.55
8.75
3
E
0.1497
0.1574
3.80
4.00
4
e
C
0.10(0.004)
B S
0.050 BSC
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
1.27 BSC
-
H
0.2284
0.2440
5.80
6.20
-
h
0.0099
0.0196
0.25
0.50
5
L
0.016
0.050
0.40
1.27
6
N
NOTES:
MILLIMETERS
α
14
0o
14
8o
0o
7
8o
Rev. 0 12/93
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006
inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead
flash and protrusions shall not exceed 0.25mm (0.010 inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch).
10. Controlling dimension: MILLIMETER. Converted inch dimensions
are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
12
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