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January 3, 2006
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contact Data
ERSIL o
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8
8
1-8
ACPI Regulator/Controller for
Dual Channel DDR Memory Systems
ISL6548
FN9188.2
Features
The ISL6548 provides a complete ACPI compliant power
solution for up to 4 DIMM dual channel DDR/DDR2 Memory
systems. Included are both a synchronous buck controller to
supply VDDQ during S0/S1 and S3 states. During S0/S1
state, a fully integrated sink-source regulator generates an
accurate (VDDQ/2) high current VTT voltage without the
need for a negative supply. Two LDO controllers are also
integrated for the GMCH core voltage regulation and for the
GMCH/CPU VTT termination voltage regulation.
The switching PWM controller drives two N-Channel
MOSFETs in a synchronous-rectified buck converter
topology. The synchronous buck converter uses voltagemode control with fast transient response. The switching
regulator provides a maximum static regulation tolerance of
2% over line, load, and temperature ranges. The output is
user-adjustable by means of external resistors down to 0.8V.
An integrated soft-start feature brings all outputs into
regulation in a controlled manner when returning to S0/S1
state from any sleep state. During S0 the VIDPGD signal
indicates that the GMCH/CPU VTT termination voltage is
within spec and operational.
Each output is monitored for undervoltage events. The
switching regulator also has overvoltage and overcurrent
protection. Thermal shutdown is integrated.
• Generates 4 Regulated Voltages
- Synchronous Buck PWM Controller for DDR VDDQ
- 3A Integrated Sink/Source Linear Regulator with
Accurate VDDQ/2 Divider Reference for DDR VTT
- LDO Regulator for GMCH Core
- Sink/Source LDO Regulator for CPU/GMCH VTT
Termination
• ACPI Compliant Sleep State Control
• Glitch-free Transitions During State Changes
• PWM Controller Drives Low Cost N-Channel MOSFETs
• 250kHz Constant Frequency Operation
• Tight Output Voltage Regulation
- All Outputs: 2% Over Temperature
• Fully-Adjustable Outputs with Wide Voltage Range: Down
to 0.8V supports DDR and DDR2 Specifications
• Simple Single-Loop Voltage-Mode PWM Control Design
• Fast PWM Converter Transient Response
• Under and Overvoltage Monitoring on All Outputs
• OCP on the Switching Regulator
• Integrated Thermal Shutdown Protection
• Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
• Single and Dual Channel DDR Memory Power Systems in
ACPI compliant PCs
Pinout
ISL6548 (6x6 QFN)
TOP VIEW
LGATE
GND
UGATE
BOOT
PHASE
S5#
OCSET
• Graphics Cards - GPU and Memory Supplies
28
27
26
25
24
23
22
• ASIC Power Supplies
• Embedded Processor and I/O Supplies
• DSP Supplies
5VSBY
1
21 DRIVE4
S3#
2
20 REFADJ4
P12V
3
19 DRIVE3
PART
PART NUMBER MARKING
GND
4
18 FB3
DDR_VTT
5
DDR_VTT
VDDQ
GND
29
Ordering Information
TEMP.
RANGE
(°C)
PACKAGE
PKG.
DWG. #
7
15 FB
28 Ld 6x6 QFN L28.6x6
Tape and Reel
(Pb-free)
8
9
10
11
12
13
14
VREF_IN
0 to 70
DRIVE2_L
ISL6548CRZA-T ISL6548CRZ
(Note)
VIDPGD
16 COMP
FB2
6
DRIVE2_U
28 Ld 6x6 QFN L28.6x6
(Pb-free)
DDR_VTTSNS
0 to 70
VDDQ
ISL6548CRZ
17 FB4
ISL6548CRZA
(Note)
1
NOTE: Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and 100% matte
tin plate termination finish, which are RoHS compliant and compatible
with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2005, 2006. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
Block Diagram
5VSBY
S3#
S5#
FB
COMP
VDDQ
P12V
BOOT
RGU
EA4
EA1
DRIVE4
2
RGL
FB4
P12V
POR
GMCH DUAL LDO
UGATE
PWM
REFADJ4
5VSBY
P12V
MONITOR AND CONTROL
EA3
FAULT
DRIVE3
250kHz
P12V
EA2
DRIVE2_U
PHASE
OC
COMP
VOLTAGE
REFERENCE
OCSET
20A
S3
0.800V
VTTSNS
0.680V (-15%)
0.920V (+15%)
FB2
P12V
VDDQ(2)
VTT
REG
VTT(2)
UV/OV
RU
UV
UV
VREF_IN
UV/OV
RL
VIDPGD
GND PAD
GND(2)
ISL6548
SOFTSTART & ENABLE A
SOFTSTART & ENABLE B
SOFTSTART & ENABLE C
ENABLE VIDPGD
ENABLE DDR_VTT
FB3
DRIVE2_L
LGATE
OSCILLATOR
FN9188.2
January 3, 2006
ISL6548
Simplified Power System Diagram
12V
5VSBY
VDDQ
5VDUAL
SLEEP
STATE
LOGIC
SLP_S3
SLP_S5
Q1
VDDQ
PWM
CONTROLLER
Q3
Q2
TWO STAGE
LINEAR
CONTROLLER
Q4
+
VGMCH
ISL6548
+
VREF
VTT
REGULATOR
LINEAR
CONTROLLER
Q5
VTT
VTT_GMCH/CPU
+
+
Q6
Typical Application
5VSBY
12V
SLP_S5
SLP_S3
S5#
DBOOT
BOOT
ROCSET
OCSET
S3#
DRIVE4
Q3
P12V
VIDPGD
VDDQ_DDR
5VSBY
5VDUAL
FB4
ISL6548
CBOOT
Q1
UGATE
VDDQ_DDR
PHASE
+
REFADJ4
DRIVE3
Q4
VGMCH
R5
DDR_VDDQ(x2)
COMP
FB3
Q2
LGATE
R2
C1
C2
R3
C3
R6
FB
R4
DRIVE2_U
Q5
R7
VTT_GMCH/CPU
VREF_IN
FB2
R8
DDR_VTT(x2)
DRIVE2_L
Q6
R1
VTT_DDR
DDR_VTTSNS
GND
3
FN9188.2
January 3, 2006
ISL6548
Absolute Maximum Ratings
Thermal Information
5VSBY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +7V
P12V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +14V
Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . +15.0V
Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . +6.0V
All other Pins . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5VCC + 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
Thermal Resistance
JA (°C/W) JC (°C/W)
QFN Package (Notes 1, 2) . . . . . . . . .
32
4
Maximum Junction Temperature (Plastic Package) . . . . . . . 150°C
Maximum Storage Temperature Range . . . . . . . . . . -65°C to 150°C
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300°C
Recommended Operating Conditions
Supply Voltage on 5VSBY . . . . . . . . . . . . . . . . . . . . . . . . +5V ±10%
Supply Voltage on P12V . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10%
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C
Junction Temperature Range. . . . . . . . . . . . . . . . . . . . 0°C to 125°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams and Typical Application Schematics
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
5.50
7.00
8.00
mA
-
700
850
A
Rising 5VSBY POR Threshold
4.10
-
4.45
V
Falling 5VSBY POR Threshold
3.60
-
3.95
V
Rising P12V POR Threshold
10.0
-
10.5
V
Falling P12V POR Threshold
8.80
-
9.75
V
5VSBY SUPPLY CURRENT
Nominal Supply Current
ICC_S0
S3# & S5# HIGH, UGATE/LGATE Open
ICC_S5
S5# LOW, S3# Don’t Care, UGATE/LGATE Open
POWER-ON RESET
OSCILLATOR AND SOFT-START
PWM Frequency
fOSC
220
250
280
kHz
Ramp Amplitude
VOSC
-
1.5
-
V
Soft-Start Interval
tSS
6.5
8.2
9.5
ms
VREF
-
0.800
-
V
-2.0
-
+2.0
%
-
80
-
dB
GBWP
15
-
-
MHz
SR
-
6
-
V/s
Low Level Input Threshold
0.75
-
-
V
High Level Input Threshold
-
-
2.2
V
REFERENCE VOLTAGE
Reference Voltage
System Accuracy
VDDQ PWM CONTROLLER ERROR AMPLIFIER
DC Gain
Guaranteed By Design
Gain-Bandwidth Product
Slew Rate
CONTROL I/O (S3# and S5#)
4
FN9188.2
January 3, 2006
ISL6548
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams and Typical Application Schematics (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
PWM CONTROLLER GATE DRIVERS
UGATE and LGATE Source
IGATE
-
-0.8
-
A
UGATE and LGATE Sink
IGATE
-
0.8
-
A
Upper Divider Impedance
RU
-
2.5
-
k
Lower Divider Impedance
RL
-
2.5
-
k
Periodic load applied with 30% duty cycle and
10ms period using ISL6548_6506EVAL1
evaluation board (see Application Note AN1123)
-3
-
3
A
Guaranteed By Design
-
80
-
dB
GBWP
15
-
-
MHz
SR
-
6
-
V/s
9.75
10.0
-
V
-
0.16
0.50
V
VTT REGULATOR
Maximum VTT Load Current
IVTT_MAX
LINEAR REGULATORS
DC GAIN
Gain Bandwidth Product
Slew Rate
DRIVEn High Output Voltage
DRIVEn unloaded
DRIVEn Low Output Voltage
DRIVEn High Output Source Current
VFB = 770mV; VDRIVEn = 0V
-
1.7
-
mA
DRIVEn Low Output Sink Current
VFB = 830mV; VDRIVEn = 10V
-
1.2
-
mA
VIDPGD
VTT_GMCH/CPU Rising Threshold
S0
0.725
0.74
-
V
VTT_GMCH/CPU Falling Threshold
S0
-
0.70
0.715
V
18
20
22
A
-3.3
-
3.3
A
PROTECTION
OCSET Current Source
IOCSET
VTT_DDR Current Limit
By Design
VDDQ OV Level
VFB/VREF
S0/S3
-
115
-
%
VDDQ UV Level
VFB/VREF
S0/S3
-
85
-
%
VTT_DDR OV Level
VTT/VVREF_IN
S0
-
115
-
%
VTT_DDR UV Level
VTT/VVREF_IN
S0
-
85
-
%
VGMCH UV Level
VFB4/VREF
S0
-
85
-
%
VTT_GMCH/CPU UV Level
VFB2/VREF
S0
-
85
-
%
By Design
-
140
-
°C
Thermal Shutdown Limit
TSD
5
FN9188.2
January 3, 2006
ISL6548
Functional Pin Description
5VSBY (Pin 1)
PHASE (Pin 24)
5VSBY is the bias supply of the ISL6548. It is typically
connected to the 5V standby rail of an ATX power supply.
During S4/S5 sleep states the ISL6548 enters a reduced
power mode and draws less than 1mA (ICC_S5) from the
5VSBY supply. The supply to 5VSBY should be locally
bypassed using a 0.1F capacitor.
Connect this pin to the upper MOSFET’s source. This pin is
used to monitor the voltage drop across the upper MOSFET
for overcurrent protection.
P12V (Pin 3)
The VTT regulation circuit and the Linear Drivers are
powered by P12V. P12V is not required during S3/S4/S5
operation. P12V is typically connected to the +12V rail of an
ATX power supply.
GND (Pins 4, 27, 29)
OCSET (Pin 22)
Connect a resistor (ROCSET) from this pin to the drain of the
upper MOSFET. ROCSET, an internal 20A current source
(IOCSET), and the upper MOSFET on-resistance (rDS(ON))
set the converter overcurrent (OC) trip point according to the
following equation:
I OCSET xR OCSET
I PEAK = ------------------------------------------------r DS  ON 
An overcurrent trip cycles the soft-start function.
The GND terminals of the ISL6548 provide the return path
for the VTT LDO, and switching MOSFET gate drivers. High
ground currents are conducted directly through the exposed
paddle of the QFN package which must be electrically
connected to the ground plane through a path as low in
inductance as possible.
The VDDQ pins should be connected externally together to
the regulated VDDQ output. During S0/S1 states, the VDDQ
pins serve as inputs to the VTT regulator and to the VTT
Reference precision divider.
UGATE (Pin 26)
DDR_VTT (Pins 5, 6)
Connect this pin to the upper MOSFET’s gate. This pin
provides the PWM-controlled gate drive for the upper
MOSFET. This pin is also monitored by the adaptive shootthrough protection circuitry to determine when the upper
MOSFET has turned off. Do not insert any circuitry between
this pin and the gate of the upper MOSFET, as it may
interfere with the internal adaptive shoot-through protection
circuitry and render it ineffective.
The DDR_VTT pins should be connect externally together.
During S0/S1 states, the DDR_VTT pins serve as the
outputs of the VTT linear regulator. During S3 state, the VTT
regulator is disabled.
LGATE (Pin 28)
Connect this pin to the lower MOSFET’s gate. This pin
provides the PWM-controlled gate drive for the lower
MOSFET. This pin is also monitored by the adaptive shootthrough protection circuitry to determine when the lower
MOSFET has turned off. Do not insert any circuitry between
this pin and the gate of the lower MOSFET, as it may
interfere with the internal adaptive shoot-through protection
circuitry and render it ineffective.
FB (Pin 15) and COMP (Pin 16)
The VDDQ switching regulator employs a single voltage
control loop. FB is the negative input to the voltage loop error
amplifier. The VDDQ output voltage is set by an external
resistor divider connected to FB. With a properly selected
divider, VDDQ can be set to any voltage between the power
rail (reduced by converter losses) and the 0.8V reference.
Loop compensation is achieved by connecting an AC
network across COMP and FB.
The FB pin is also monitored for under and overvoltage
events.
6
VDDQ (Pins 7, 8)
DDR_VTTSNS (Pin 9)
VTTSNS is used as the feedback for control of the VTT linear
regulator. Connect this pin to the VTT output at the physical
point of desired regulation.
VREF_IN (Pin 14)
A capacitor, CSS, connected between VREF_IN and ground
is required. This capacitor and the parallel combination of
the Upper and Lower Divider Impedance (RU||RL), sets the
time constant for the start up ramp when transitioning from
S3/S4/S5 to S0/S1/S2.
The minimum value for CSS can be found through the
following equation:
C VTTOUT  V DDQ
C SS  -----------------------------------------------10  2A  R U  R L
The calculated capacitance, CSS, will charge the output
capacitor bank on the VTT rail in a controlled manner without
reaching the current limit of the VTT LDO.
BOOT (Pin 25)
This pin provides ground referenced bias voltage to the
upper MOSFET driver. A bootstrap circuit is used to create a
voltage suitable to drive a logic-level N-channel MOSFET.
FN9188.2
January 3, 2006
ISL6548
Functional Description
FB2 (Pin 11)
Connect the output of the VTT_GMCH/CPU linear regulator to
this pin through a properly sized resistor divider. The voltage
at this pin is regulated to 0.8V. This pin is monitored for
undervoltage events.
DRIVE2 _U (Pin 10)
This pin provides the gate voltage for the VTT_GMCH/CPU
linear regulator upper pass transistor. Connect this pin to the
gate terminal of an external N-Channel MOSFET transistor.
DRIVE2 _L (Pin 13)
This pin provides the gate voltage for the VTT_GMCH/CPU
linear regulator lower pass transistor. Connect this pin to the
gate terminal of an external N-Channel MOSFET transistor.
FB3 (Pin 18)
Connect the output of the lower VGMCH linear regulator to
this pin through a properly sized resistor divider. The voltage
at this pin is regulated to 0.8V. This pin is monitored for
undervoltage events.
DRIVE3 (Pin 19)
This pin provides the gate voltage for the lower VGMCH
linear regulator pass transistor. Connect this pin to the gate
terminal of an external N-Channel MOSFET transistor.
Overview
The ISL6548 provides complete control, drive, protection
and ACPI compliance for a regulator powering DDR memory
systems and the GMCH core and GMCH/CPU termination
rails. It is primarily designed for computer applications
powered from an ATX power supply.
A 250kHz Synchronous Buck Regulator with a precision
0.8V reference provides the proper Core voltage to the
system memory of the computer. An internal LDO regulator
with the ability to both sink and source current tracks the
VDDQ output by 50% and provides the VTT termination
voltage.
A dual stage LDO controller provides the GMCH core
voltage. A third LDO controller is included for the regulation
of the GMCH/CPU termination rail.
ACPI compliance is realized through the SLP_S3 and
SLP_S5 sleep signals and through monitoring of the 12V
ATX bus.
Initialization
Connect the output of the upper VGMCH linear regulator to
this pin. The voltage at this pin is regulated via the RAFADJ4
pin.
The ISL6548 automatically initializes upon receipt of input
power. Special sequencing of the input supplies is not
necessary. The Power-On Reset (POR) function continually
monitors the input bias supply voltages. The POR monitors
the bias voltage at the 5VSBY and P12V pins. The POR
function initiates soft-start operation after the bias supply
voltages exceed their POR thresholds.
DRIVE4 (Pin 21)
ACPI State Transitions
This pin provides the gate voltage for the upper VGMCH
linear regulator pass transistor. Connect this pin to the gate
terminal of an external N-Channel MOSFET transistor.
Figure 1 shows how the individual regulators are controlled
during all state transitions. All references to timing in this
section are in reference to Figure 1.
REFADJ4 (Pin 20)
Cold Start (S4/S5 to S0 Transition)
This pin controls the reference for the upper VGMCH linear
regulator. To insure that both upper and lower pass
transistors dissipate the same power, tie this pin to the
VGMCH output rail.
At the onset of a mechanical start, time t0 in Figure 1, the
ISL6548 receives its bias voltage from the 5V Standby bus
(5VSBY). Once the 5VSBY rail has exceeded the POR
threshold, the ISL6548 will remain in an internal S5 state
until both the SLP_S3 and SLP_S5 signal have transitioned
high and the 12V POR threshold has been exceeded by the
+12V rail from the ATX, which occurs at time t1.
FB4 (Pin 17)
VIDPGD (Pin 12)
The VIDPGD pin is an open-drain logic output that changes
to a logic low if the VTT_GMCH/CPU linear regulator is out of
regulation in S0/S1/S2 state. VIDPGD will always be low in
any state other than S0/S1/S2.
S5# (Pin 23)
This pin accepts the SLP_S5# sleep state signal.
S3# (Pin 2)
This pin accepts the SLP_S3# sleep state signal.
Once all of these conditions are met, the PWM error
amplifier will first be reset by internally shorting the COMP
pin to the FB pin. This reset lasts for three soft-start cycles,
which is typically 24ms (one soft-start cycle is typically
8.2ms). The digital soft-start sequence will then begin. Each
regulator is enabled and soft-started according to a preset
sequence.
At time t2, the 3 soft-start cycle reset has ended and the
VDDQ_DDR rail and the upper VGMCH LDO are digitally softstarted.
7
FN9188.2
January 3, 2006
SLP_S3#
SLP_S5#
8
12V
POR
12V
0V
VDDQ_DDR
0V
VTT_DDR Soft-Start Rise Time Dependent Upon Capacitor On VREF_IN Pin
VDDQ_DDR
VTT_DDR
ISL6548
VTT_DDR FLOATING
0V
VGMCH_UPPER
0V
VGMCH
0V
VTT_GMCH/CPU
0V
VIDPGD
t0
(3 SOFTSTART CYCLES)
t1
t2
t3
t4
t5
t6
t7
t8
(3 SOFTSTART CYCLES)
t10
t9
FN9188.2
January 3, 2006
FIGURE 1. ISL6548 TIMING DIAGRAM
t11
t12
t13
t14
t15
ISL6548
The digital soft-start for the PWM regulator is accomplished by
clamping the error amplifier reference input to a level
proportional to the internal digital soft-start voltage. As the softstart voltage slews up, the PWM comparator generates PHASE
pulses of increasing width that charge the output capacitor(s).
This method provides a rapid and controlled output voltage rise.
The linear regulators, with the exception of the internal
VTT_DDR LDO, are soft-started in a similar manner. The
error amplifier reference is clamped to the internal digital
soft-start voltage. As the soft-start voltage ramps up, the
respective DRIVE pin voltages increase, thus enhancing the
N-MOSFETs and charging the output capacitors in a
controlled manner.
At time t3, the VDDQ_DDR and upper VGMCH LDO output
rails are in regulation and the lower VGMCH LDO is softstarted. At time t4, the VGMCH rail is in regulation and the
VTT_GMCH/CPU linear regulator is soft-started. At time t5,
the VTT_GMCH/CPU rail is in regulation and the VTT_DDR
internal regulator is soft-started.
The VTT_DDR LDO soft-starts in a manner unlike the other
regulators. When the VTT_DDR regulator is disabled, the
reference is internally shorted to the VTT_DDR output. This
allows the termination voltage to float during the S3 sleep
state. When the ISL6548 enables the VTT_DDR regulator or
enters S0 state from a sleep state, this short is released and
the internal divide down resistors which set the VTT_DDR
voltage to 50% of VDDQ_DDR will provide a controlled
voltage rise on the capacitor that is tied to the VREF_IN pin.
The voltage on this capacitor is the reference for the
VTT_DDR regulator and the output will track it as it settles to
50% of the VDDQ voltage. The combination of the internal
resistors and the VREF_IN capacitor will determine the rise
time of the VTT_DDR regulator (see the Functional Pin
Description section for proper sizing of the VREF_IN
capacitor).
At time t6, a full soft-start cycle has passed from the time that
the VTT_DDR regulator was enabled. At this time the
VIDPGD comparator is enabled. Once enabled if the
VTT_GMCH/CPU output is within regulation, the VIDPGD pin
will be forced to a high impedance state.
Active to Sleep (S0 to S3 Transition)
When SLP_S3 goes LOW with SLP_S5 still HIGH, the
ISL6548 will disable all the regulators except for the VDDQ
regulator, which is continually supplied by the 5VDUAL rail.
VIDPGD will also transition LOW. When VTT is disabled, the
internal reference for the VTT regulator is internally shorted
to the VTT rail. This allows the VTT rail to float. When
floating, the voltage on the VTT rail will depend on the
leakage characteristics of the memory and MCH I/O pins. It
is important to note that the VTT rail may not bleed down to
0V. Figure 1 shows how the individual regulators are
affected by the S3 state at time t7.
9
Sleep to Active (S3 to S0 Transition)
When SLP_S3 transitions from LOW to HIGH with SLP_S5
held HIGH and after the 12V rail exceeds POR, the ISL6548
will initiate the soft-start sequence. This sequence is very
similar to the mechanical start soft-start sequencing. The
transition from S3 to S0 is represented in Figure 1 between
times t8 and t14.
At time t8, the SLP_S3 signal transitions HIGH. This enables
the ATX, which brings up the 12V rail. At time t9, the 12V rail
has exceeded the POR threshold and the ISL6548 enters a
reset mode that lasts for 3 soft-start cycles. At time t10, the 3
soft-start cycle reset is ended and the individual regulators
are enabled and soft-started in the same sequence as the
mechanical cold start sequence, with the exception that the
VDDQ regulator is already enabled and in regulation.
Active to Shutdown (S0 to S5 Transition)
When the system transitions from active, S0, state to
shutdown, S4/S5, state, the ISL6548 IC disables all
regulators and forces the VIDPGD pin LOW. This transition
is represented on Figure 1 at time t15.
Fault Protection
The ISL6548 monitors the VDDQ regulator for under and
overvoltage events. The VDDQ regulator also has overcurrent
protection. The internal VTT_DDR LDO regulator is monitored
for under and overvoltage events. All other regulators are
monitored for undervoltage events.
An overvoltage event on either the VDDQ or VTT_DDR
regulator will cause an immediate shutdown of all regulators.
This can only be cleared by toggling the SLP_S5 signal such
that the system enters the S5 sleep state and then
transitions back to the active, S0, state.
If a regulator experiences any other fault condition (an
undervoltage or an overcurrent on VDDQ), then that
regulator, and only that regulator, will be disabled and an
internal fault counter will be incremented by 1. If the disabled
regulator is used as the input for another regulator, then that
cascoded regulator will also experience a fault condition due
to a loss of input. The cascoded regulator will be disabled
and the fault counter incremented by 1.
At every fault occurrence, the internal fault counter is
incremented by 1 and an internal Fault Reset Counter is
cleared to zero. The Fault Reset Counter will increment once
for every clock cycle (1 clock cycle is typically 1/250kHz, or
4s). If the Fault Reset Counter reaches a count of 16384
before another fault occurs, then the Fault Counter is
cleared to 0. If a fault occurs prior to the Fault Reset Counter
reaching a count of 16384, then the Fault Reset Counter is
set back to zero.
The ISL6548 will immediately shut down when the Fault
Counter reaches a count of 4 when the system is restarting
from an S5 state into the active, or S0, state. The ISL6548
FN9188.2
January 3, 2006
ISL6548
will immediately shut down when the Fault Counter reaches
a count of 5 at any other time.
The 16384 counts that are required to reset the Fault Reset
Counter represent 8 soft-start cycles, as one soft-start cycle
is 2048 clock cycles. This allows the ISL6548 to attempt at
least one full soft-start sequence to restart the faulted
regulators.
When attempting to restart a faulted regulator, the ISL6548
will follow the preset start up sequencing. If a regulator is
already in regulation, then it will not be affected by the start
up sequencing.
VDDQ Overcurrent Protection
The overcurrent function protects the switching converter from
a shorted output by using the upper MOSFET on-resistance,
rDS(ON), to monitor the current. This method enhances the
converter’s efficiency and reduces cost by eliminating a
current sensing resistor.
The overcurrent function cycles the soft-start function in a
hiccup mode to provide fault protection. A resistor (ROCSET)
programs the overcurrent trip level (see Typical Application
diagrams on page 3). An internal 20A (typical) current sink
develops a voltage across ROCSET that is referenced to the
converter input voltage. When the voltage across the upper
MOSFET (also referenced to the converter input voltage)
exceeds the voltage across ROCSET, the overcurrent function
initiates a soft-start sequence. The initiation of soft-start may
affect other regulators. The VTT_DDR regulator is directly
affected as it receives it’s reference and input from VDDQ.
The overcurrent function will trip at a peak inductor current
(IPEAK) determined by:
I OCSET x R OCSET
I PEAK = ---------------------------------------------------r DS  ON 
Thermal Protection (S0/S3 State)
If the ISL6548 IC junction temperature reaches a nominal
temperature of 140°C, all regulators will be disabled. The
ISL6548 will not re-enable the outputs until the junction
temperature drops below 110°C and either the bias voltage is
toggled in order to initiate a POR or the SLP_S5 signal is
forced LOW and then back to HIGH.
Shoot-Through Protection
A shoot-through condition occurs when both the upper and
lower MOSFETs are turned on simultaneously, effectively
shorting the input voltage to ground. To protect from a shootthrough condition, the ISL6548 incorporates specialized
circuitry on the VDDQ regulator which insures that
complementary MOSFETs are not ON simultaneously.
The adaptive shoot-through protection utilized by the VDDQ
regulator looks at the lower gate drive pin, LGATE, and the
upper gate drive pin, UGATE, to determine whether a
MOSFET is ON or OFF. If the voltage from UGATE or from
LGATE to GND is less than 0.8V, then the respective
MOSFET is defined as being OFF and the other MOSFET is
allowed to turned ON. This method allows the VDDQ
regulator to both source and sink current.
Since the voltage of the MOSFET gates are being measured
to determine the state of the MOSFET, the designer is
encouraged to consider the repercussions of introducing
external components between the gate drivers and their
respective MOSFET gates before actually implementing
such measures. Doing so may interfere with the shootthrough protection.
Application Guidelines
Layout Considerations
where IOCSET is the internal OCSET current source (20A
typical). The OC trip point varies mainly due to the MOSFET
rDS(ON) variations. To avoid overcurrent tripping in the
normal operating load range, find the ROCSET resistor from
the equation above with:
1. The maximum rDS(ON) at the highest junction
temperature.
2. The minimum IOCSET from the specification table.
 I 
3. Determine IPEAK for I PEAK > I OUT  MAX  + ---------- ,
2
whereI is the output inductor ripple current.
For an equation for the ripple current see the section under
component guidelines titled ‘Output Inductor Selection’.
A small ceramic capacitor should be placed in parallel with
ROCSET to smooth the voltage across ROCSET in the
presence of switching noise on the input voltage.
Layout is very important in high frequency switching
converter design. With power devices switching efficiently at
250kHz, the resulting current transitions from one device to
another cause voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit,
and lead to device overvoltage stress. Careful component
layout and printed circuit board design minimizes these
voltage spikes.
As an example, consider the turn-off transition of the control
MOSFET. Prior to turn-off, the MOSFET is carrying the full
load current. During turn-off, current stops flowing in the
MOSFET and is picked up by the lower MOSFET. Any
parasitic inductance in the switched current path generates a
large voltage spike during the switching interval. Careful
component selection, tight layout of the critical components,
and short, wide traces minimizes the magnitude of voltage
spikes.
There are two sets of critical components in the ISL6548
switching converter. The switching components are the most
10
FN9188.2
January 3, 2006
ISL6548
A multi-layer printed circuit board is recommended. Figure 2
shows the connections of the critical components in the
converter. Note that capacitors CIN and COUT could each
represent numerous physical capacitors. Dedicate one solid
layer, usually a middle layer of the PC board, for a ground
plane and make all critical component ground connections
with vias to this layer. Dedicate another solid layer as a
power plane and break this plane into smaller islands of
common voltage levels. Keep the metal runs from the
PHASE terminals to the output inductor short. The power
plane should support the input power and output power
nodes. Use copper filled polygons on the top and bottom
circuit layers for the phase nodes. Use the remaining printed
circuit layers for small signal wiring. The wiring traces from
the GATE pins to the MOSFET gates should be kept short
and wide enough to easily handle the 1A of drive current.
12VATX
P12V
GNDP
CBP
5VDUAL
5VSBY
5VSBY
CIN
CBP
ISL6548
UGATE
Q1
LGATE
COMP
Q2
C2
R1
FB
C3 R3
R4
VDDQ
VDDQ(2)
VTT(2)
VTT
LOAD
COUT2
Q3
DRIVE4
FB4
COUT3
Q4
DRIVE3
VGMCH
R5
FB3
COUT4
LOAD
R6
The critical small signal components include any bypass
capacitors, feedback components, and compensation
components. Place the PWM converter compensation
components close to the FB and COMP pins. The feedback
resistors should be located as close as possible to the FB
pin with vias tied straight to the ground plane as required.
REFADJ4
Q5
DRIVE2_U
VTT_GMCH/CPU
R7
FB2
DRIVE2_L
Q6
COUT4
LOAD
R8
Feedback Compensation - PWM Buck Converter
11
COUT1
C1
R2
The switching components should be placed close to the
ISL6548 first. Minimize the length of the connections
between the input capacitors, CIN, and the power switches
by placing them nearby. Position both the ceramic and bulk
input capacitors as close to the upper MOSFET drain as
possible. Position the output inductor and output capacitors
between the upper and lower MOSFETs and the load.
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR . The DC Gain of
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage VOSC .
VDDQ
PHASE
In order to dissipate heat generated by the internal VTT
LDO, the ground pad, pin 29, should be connected to the
internal ground plane through at least four vias. This allows
the heat to move away from the IC and also ties the pad to
the ground plane through a low impedance path.
Figure 3 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The error
amplifier output (VE/A) is compared with the oscillator (OSC)
triangular wave to provide a pulse-width modulated (PWM)
wave with an amplitude of VIN at the PHASE node. The
PWM wave is smoothed by the output filter (LO and CO).
L1
LOAD
critical because they switch large amounts of energy, and
therefore tend to generate large amounts of noise. Next are
the small signal components which connect to sensitive
nodes or supply critical bypass current and signal coupling.
GND PAD
KEY
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT AND/OR POWER PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 2. PRINTED CIRCUIT BOARD POWER PLANES
AND ISLANDS
FN9188.2
January 3, 2006
ISL6548
PWM
COMPARATOR
LO
-
VOSC
Compensation Break Frequency Equations
VIN
DRIVER
OSC
DRIVER
+
PHASE
VDDQ
CO
ESR
(PARASITIC)
ZFB
VE/A
-
ZIN
+
REFERENCE
ERROR
AMP
DETAILED COMPENSATION COMPONENTS
ZFB
C1
C2
VDDQ
ZIN
C3
R2
R3
R1
COMP
FB
+
R4
ISL6548
REFERENCE
R 

V DDQ = 0.8   1 + ------1-
R 4

FIGURE 3. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN AND OUTPUT
VOLTAGE SELECTION
1
F Z1 = -----------------------------------2 x R 2 x C 1
1
F P1 = -------------------------------------------------------- C 1 x C 2
2 x R 2 x  ----------------------
 C1 + C2 
1
F Z2 = ------------------------------------------------------2 x  R 1 + R 3  x C 3
1
F P2 = -----------------------------------2 x R 3 x C 3
Figure 4 shows an asymptotic plot of the DC/DC converter’s
gain vs. frequency. The actual Modulator Gain has a high
gain peak due to the high Q factor of the output filter and is
not shown in Figure 4. Using the above guidelines should
give a Compensation Gain similar to the curve plotted. The
open loop error amplifier gain bounds the compensation
gain. Check the compensation gain at FP2 with the
capabilities of the error amplifier. The Closed Loop Gain is
constructed on the graph of Figure 4 by adding the
Modulator Gain (in dB) to the Compensation Gain (in dB).
This is equivalent to multiplying the modulator transfer
function to the compensation transfer function and plotting
the gain.
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW) overall
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
100
FZ1 FZ2
FP1
FP2
80
Modulator Break Frequency Equations
1
F ESR = -------------------------------------------2 x ESR x C O
The compensation network consists of the error amplifier
(internal to the ISL6548) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R1 , R2 ,
R3 , C1 , C2 , and C3) in Figure 5. Use these guidelines for
locating the poles and zeros of the compensation network:
1. Pick Gain (R2/R1) for desired converter bandwidth.
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).
3. Place 2ND Zero at Filter’s Double Pole.
4. Place 1ST Pole at the ESR Zero.
5. Place 2ND Pole at Half the Switching Frequency.
6. Check Gain against Error Amplifier’s Open-Loop Gain.
7. Estimate Phase Margin - Repeat if Necessary.
12
GAIN (dB)
1
F LC = ------------------------------------------2 x L O x C O
OPEN LOOP
ERROR AMP GAIN
60
40
20
20LOG
(R2/R1)
20LOG
(VIN/VOSC)
0
COMPENSATION
GAIN
MODULATOR
GAIN
-20
-40
-60
CLOSED LOOP
GAIN
FLC
10
100
1K
FESR
10K
100K
1M
10M
FREQUENCY (Hz)
FIGURE 4. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Output Voltage Selection
The output voltage of the all the external voltage regulators
converter can be programmed to any level between their
individual input voltage and the internal reference, 0.8V. An
external resistor divider is used to scale the output voltage
relative to the reference voltage and feed it back to the
inverting input of the error amplifier, refer to the Typical
Application on page 3.
FN9188.2
January 3, 2006
ISL6548
The output voltage programming resistor will depend on the
value chosen for the feedback resistor and the desired
output voltage of the particular regulator.
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
R1  0.8V
R4 = ----------------------------------V DDQ – 0.8V
Output Capacitor Selection - LDO Regulators
The output capacitors used in LDO regulators are used to
provide dynamic load current. The amount of capacitance
and type of capacitor should be chosen with this criteria in
mind.
R5  0.8V
R6 = ---------------------------------------V GMCH – 0.8V
Output Inductor Selection
R7  0.8V
R8 = ------------------------------------------------------------V TT_GMCH/CPU – 0.8V
If the output voltage desired is 0.8V, simply route the output
voltage back to the respective FB pin through the feedback
resistor and do not populate the output voltage programming
resistor.
The output voltage for the internal VTT_DDR linear regulator
is set internal to the ISL6548 to track the VDDQ voltage by
50%. There is no need for external programming resistors.
Component Selection Guidelines
Output Capacitor Selection - PWM Buck Converter
An output capacitor is required to filter the inductor current
and supply the load transient current. The filtering
requirements are a function of the switching frequency and
the ripple current. The load transient requirements are a
function of the slew rate (di/dt) and the magnitude of the
transient load current. These requirements are generally met
with a mix of capacitors and careful layout.
DDR memory systems are capable of producing transient
load rates above 1A/ns. High frequency capacitors initially
supply the transient and slow the current load rate seen by the
bulk capacitors. The bulk filter capacitor values are generally
determined by the ESR (Effective Series Resistance) and
voltage rating requirements rather than actual capacitance
requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
13
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
I =
VIN - VOUT
Fs x L
x
VOUT
VIN
VOUT = I x ESR
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6548 will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
VOUT
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Input Capacitor Selection - PWM Buck Converter
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time the upper MOSFET
FN9188.2
January 3, 2006
ISL6548
turns on. Place the small ceramic capacitors physically close
to the MOSFETs and between the drain of upper MOSFET
and the source of lower MOSFET.
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. Their voltage rating should be
at least 1.25 times greater than the maximum input voltage,
while a voltage rating of 1.5 times is a conservative
guideline. For most cases, the RMS current rating
requirement for the input capacitor of a buck regulator is
approximately 1/2 the DC load current.
The maximum RMS current required by the regulator may be
closely approximated through the following equation:
I RMS
MAX
=
V IN – V OUT V OUT 2
V OUT 
2
1
--------------  I OUT
+ ------   -----------------------------  -------------- 

V IN
V IN  
12  L  f s
MAX
For a through hole design, several electrolytic capacitors
may be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge-current at
power-up. Some capacitor series available from reputable
manufacturers are surge current tested.
MOSFET Selection - PWM Buck Converter
The ISL6548 requires 2 N-Channel power MOSFETs for
switching power. These should be selected based upon
rDS(ON) , gate supply requirements, and thermal
management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss
components; conduction loss and switching loss. The
conduction losses are the largest component of power
dissipation for both the upper and the lower MOSFETs.
These losses are distributed between the two MOSFETs
according to duty factor. The switching losses seen when
sourcing current will be different from the switching losses
seen when sinking current. When sourcing current, the
upper MOSFET realizes most of the switching losses. The
lower switch realizes most of the switching losses when the
converter is sinking current (see the equations below).
These equations assume linear voltage-current transitions
14
and do not adequately model power loss due the reverserecovery of the upper and lower MOSFET’s body diode. The
gate-charge losses are dissipated in part by the ISL6548 and
do not significantly heat the MOSFETs. However, large gatecharge increases the switching interval, tSW which increases
the MOSFET switching losses. Ensure that both MOSFETs
are within their maximum junction temperature at high
ambient temperature by calculating the temperature rise
according to package thermal-resistance specifications. A
separate heatsink may be necessary depending upon
MOSFET power, package type, ambient temperature and air
flow.
Approximate Losses while Sourcing current
2
1
P UPPER = Io  r DS  ON   D + ---  Io  V IN  t SW  f s
2
PLOWER = Io2 x rDS(ON) x (1 - D)
Approximate Losses while Sinking current
PUPPER = Io2 x rDS(ON) x D
2
1
P LOWER = Io  r DS  ON    1 – D  + ---  Io  V IN  t SW  f s
2
Where: D is the duty cycle = VOUT / VIN ,
tSW is the combined switch ON and OFF time, and
fs is the switching frequency.
MOSFET Selection - LDO
The main criteria for selection of the linear regulator pass
transistor is package selection for efficient removal of heat.
Select a package and heatsink that maintains the junction
temperature below the rating with a maximum expected
ambient temperature.
The power dissipated in the linear regulator is:
P LINEAR  I O   V IN – V OUT 
where IO is the maximum output current and VOUT is the
nominal output voltage of the linear regulator.
FN9188.2
January 3, 2006
ISL6548
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L28.6x6
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VJJC ISSUE C)
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
9
A3
b
0.20 REF
0.23
D
0.35
5, 8
6.00 BSC
D1
D2
0.28
9
-
5.75 BSC
3.95
4.10
9
4.25
7, 8
E
6.00 BSC
-
E1
5.75 BSC
9
E2
3.95
e
4.10
4.25
7, 8
0.65 BSC
-
k
0.25
-
-
-
L
0.35
0.60
0.75
8
L1
-
-
0.15
10
N
28
2
Nd
7
3
Ne
7
3
P
-
-
0.60
9

-
-
12
9
Rev. 1 10/02
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P &  are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9001 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
15
FN9188.2
January 3, 2006
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