DATASHEET

Triple, 180° Out-of-Phase, Synchronous Step-Down
PWM Controller
ISL9443
Features
The ISL9443 is a triple-output synchronous buck controller that
integrates three PWM controllers which are fully featured and
designed to provide multi-rail power for use in products such as
cable and satellite set-top boxes, VoIP gateways, cable modems,
and other home connectivity products as well as a variety of
industrial and general purpose applications. Each output is
adjustable down to 0.7V. The PWMs are synchronized at 180°
out-of-phase, thus reducing the input RMS current and ripple
voltage.
• Three Integrated Synchronous Buck PWM Controllers
- Internal Bootstrap Diodes
- Independent Programmable Output Voltage
- Independent Soft-Starting and Tracking
• Power-Good Indicator
• Light Load Efficiency Enhancement
- Low Ripple Diode Emulation Mode with Pulse Skipping
• Supports Pre-Biased Output
The ISL9443 offers programmable soft-start and tracking
functions for ease of supply rail sequencing and integrated
UV/OV/OC/OT protections in a space conscious 5mmx5mm QFN
package.
• Programmable Frequency: 200kHz to 1200kHz
• Adaptive Shoot-Through Protection
• Out-of-Phase Switching (0°/180°/0°)
Switching frequency can be set between 200kHz and 1200kHz
using a resistor. The ISL9443 can be synchronized to an external
clock to reduce beat frequencies.
• No External Current Sense Resistor
- Uses Lower MOSFET’s rDS(ON)
• Complete Protection
- Overcurrent, Overvoltage, Over-Temperature
The ISL9443 utilizes internal loop compensation to keep
minimum peripheral components for a compact design and a
low total solution cost. The controller is implemented with
current mode control with feed forward to cover various
applications even with fixed internal compensation.
• Wide Input Voltage Range: 4.5V to 28V
• Pb-Free (RoHS Compliant)
Applications
Related Literature
• VoX Gateway Devices
• Technical Brief TB389 “PCB Land Pattern Design and Surface
Mount Guidelines for QFN (MLFP) Packages”
• NAS/SAN Devices
• ATX power supplies
+12V
+
Q1
Q2
CIN2
0.1µF
C1
4.7µF
R5
100kΩ
R6
200kΩ
LGATE2
PHASE2
R7
200kΩ
LGATE3
FB3
TK/SS2,3
MODE/SYNC
OCSET2
OCSET1
CSS
10nF
SGND
EN/SS1
OCSET2
FB1
RT
ISEN3
CO2
100µF
Q3
BOOT3
PHASE3
R9
11.5kΩ
R8
3.09kΩ
FB2
UGATE3
EN23
RT
49.9kΩ
ISEN2
+12V
ISL9443
R4
15.8kΩ
R3
31.6kΩ
BOOT2
VIN
VCC_5V
UGATE2
ISEN1
+12V
PGND
CO1
100µF
VOUT2
+3.3V,6A
+
RESN2
1.3kΩ
CB2
CB1
BOOT1
1.0µH
L2
2.2µH
0.1µF
UGATE1
+
RESN1
1.3kΩ
PHASE1
L1
LGATE1
VOUT1
+1.0V, 6A
PGOOD
CIN1
CB3
0.1µF
L3
3.3µH
VOUT3
+5.0V,6A
+
RESN3
1.3kΩ
CO3
100µF
R4
10.7kΩ
R3
1.74kΩ
FIGURE 1. TYPICAL APPLICATION
February 24, 2012
FN7663.1
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2011, 2012. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL9443
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
ISL9443IRZ
PART
MARKING
TEMP. RANGE
(°C)
ISL9443 IRZ
PACKAGE
(Pb-Free)
-40 to +85
32 Ld 5x5 QFN
PKG.
DWG. #
L32.5X5B
NOTES:
1. Add “-T*” for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL9443. For more information on MSL please see techbrief TB363.
Pin Configuration
PHASE1
BOOT1
UGATE1
LGATE1
LGATE2
UGATE2
BOOT2
PHASE2
ISL9443
(32 LD 5x5 QFN)
TOP VIEW
32
31
30
29
28
27
26
25
PGND
VIN
3
22
LGATE3
EN/SS1
4
21
UGATE3
FB1
5
20
BOOT3
OCSET1
6
19
PHASE3
RT
7
18
ISEN3
8
17
MODE/SYNC
9
10
11
12
13
14
15
16
TK/SS3
PGOOD
FB3
23
TK/SS2
2
OCSET3
VCC_5V
FB2
ISEN2
OCSET2
24
SGND
1
EN23
ISEN1
Pin Descriptions
PIN
NAME
FUNCTION
1
ISEN1
2
VCC_5V
Output of the internal 5V linear regulator. This output supplies bias for the IC, the low side gate drivers, and the external
boot circuitry for the high-side gate drivers. The VCC_5V pin must be always decoupled to power ground with a minimum
of 4.7µF ceramic capacitor, placed very close to the pin. Do not allow the voltage at VCC_5V to exceed VIN at any time.
3
VIN
This pin should be tied to the input rail. It provides power to the internal linear drive circuitry and is also used by the feedforward controller to adjust the amplitude of each PWM sawtooth. Decouple this pin with a small ceramic capacitor
(0.1µF to 1µF) to ground.
4
EN/SS1
This pin provides an enable/disable function and soft-starting for PWM1 output. The output is disabled when the pin is
pulled to GND. During start-up, a regulated 1.55µA soft-start current charges an external capacitor connected at this pin.
When the voltage on the EN/SS1 pin reaches 1.3V, the PWM1 output becomes active. From 1.3V to 2.0V, the reference
voltage of the PWM1 is clamped to the voltage at EN/SS1 minus 1.3V. The capacitance of the soft-start capacitors sets
the soft-starting time and enable delay time. Setting the soft-starting time too short might create undesirable overshoot
at the output during start-up. VCC_5V UVLO discharges the EN/SS1 via an internal MOSFET.
Current signal input for PWM1. This pin is used to monitor the voltage drop across the lower MOSFET for current loop
feedback and overcurrent protection.
2
FN7663.1
February 24, 2012
ISL9443
Pin Descriptions
(Continued)
PIN
NAME
FUNCTION
5
FB1
PWM1 feedback input. Connect FB1 to a resistive voltage divider from the output of PWM1 to GND to adjust the output
voltage.
6
OCSET1
7
RT
A resistor from this pin to ground adjusts the overcurrent threshold for PWM1.
A resistor from this pin to ground adjusts the switching frequency from 200kHz to 1.2MHz.
R T = ( 23.36 × ( 1.5 × t SW – 0.36 ) ) ⋅ kΩ
(EQ. 1)
Where tSW is the switching period in µs.
8
PGOOD
Open drain logic output used to indicate the status of the PWM output voltages. This pin is pulled LOW when any of the
outputs is not within ±11% of the nominal voltage.
9
EN23
Enable/Disable input for PWM2 and PWM3. The outputs of PWM2 and PWM3 are enabled when this pin is pulled HIGH,
and disabled when this pin is pulled LOW. Do not float this pin.
10
SGND
This is the small-signal ground common to all 3 controllers. It is suggested to route this separately from the high current
ground (PGND). SGND and PGND can be tied together if there is one solid ground plane with no noisy currents around
the chip. All voltage levels are measured with respect to this pin.
11
OCSET2
12
FB2
PWM2 feedback input. Connect FB2 to a resistive voltage divider from the output of PWM2 to GND to adjust the output
voltage.
13
TK/SS2
Dual function pin. The reference voltage of PWM2 is clamped to the voltage at TK/SS2 during start-up. When this pin is
used for tracking, another channel is configured as the master and the output voltage of the master channel is applied
to this pin via a resistor divider.
When used for soft-starting control, a soft-start capacitor is connected from this pin to GND. A regulated 1.55µA soft-starting
current charges up the soft-start capacitor. Value of the soft-start capacitor sets the PWM2 output voltage ramp.
14
OCSET3
A resistor from this pin to ground adjusts the overcurrent threshold for PWM3.
15
FB3
PWM3 feedback input. Connect FB3 to a resistive voltage divider from the output of PWM3 to GND to adjust the output
voltage.
16
TK/SS3
Dual function pin. The reference voltage of PWM3 is clamped to the voltage at TK/SS3 during start-up. When this pin is
used for tracking, another channel is configured as the master and the output voltage of the master channel is applied
to this pin via a resistor divider.
When used for soft-starting control, a soft-start capacitor is connected from this pin to GND. A regulated 1.55µA soft-starting
current charges up the soft-start capacitor. Value of the soft-start capacitor sets the PWM3 output voltage ramp.
17
MODE/SYNC
Dual function pin. Tie this pin to ground or VCC_5V for DEM or CCM operation mode selection. Connect this pin to ground
to select Diode Emulation Mode with pulse skipping at light load. While connected to VCC_5V, the controllers operate in
PWM Mode at light load.
Connect this pin to an external clock for synchronization. The controller operates in PWM mode at light load when
synchronized with an external clock.
18
ISEN3
19
PHASE3
Phase node connection for PWM3. This pin is connected to the junction of the upper MOSFET’s source, output filter inductor,
and lower MOSFET’s drain. PHASE3 is the internal lower supply rail for UGATE3.
20
BOOT3
Bootstrap pin to provide bias for PWM3 high-side driver. The positive terminal of the bootstrap capacitor connects to this
pin. The bootstrap diodes are integrated to help reduce total cost and reduce layout complexity.
21
UGATE3
High-side MOSFET gate driver output for PWM3.
22
LGATE3
Low-side MOSFET gate driver output for PWM3.
23
PGND
Power ground connection for all three PWM channels. This pin should be connected to the sources of the lower MOSFETs
and the (-) terminals of the external input capacitors
24
ISEN2
Current signal input for PWM2. This pin is used to monitor the voltage drop across the lower MOSFET for current loop
feedback and overcurrent protection.
25
PHASE2
A resistor from this pin to ground adjusts the overcurrent threshold for PWM2.
Current signal input for PWM3. This pin is used to monitor the voltage drop across the lower MOSFET for current loop
feedback and overcurrent protection.
Phase node connection for PWM2. This pin is connected to the junction of the upper MOSFET’s source, output filter inductor,
and lower MOSFET’s drain. PHASE2 is the internal lower supply rail for UGATE2.
3
FN7663.1
February 24, 2012
ISL9443
Pin Descriptions
(Continued)
PIN
NAME
FUNCTION
26
BOOT2
Bootstrap pin to provide bias for PWM2 high-side driver. The positive terminal of the bootstrap capacitor connects to this
pin. The bootstrap diodes are integrated to help reduce total cost and reduce layout complexity.
27
UGATE2
High-side MOSFET gate driver output for PWM2.
28
LGATE2
Low-side MOSFET gate driver output for PWM2.
29
LGATE1
Low-side MOSFET gate driver output for PWM1.
30
UGATE1
High-side MOSFET gate driver output for PWM1.
31
BOOT1
Bootstrap pin to provide bias for PWM1 high-side driver. The positive terminal of the bootstrap capacitor connects to this
pin. The bootstrap diodes are integrated to help reduce total cost and reduce layout complexity.
32
PHASE1
Phase node connection for PWM1. This pin is connected to the junction of the upper MOSFET’s source, output filter inductor,
and lower MOSFET’s drain. PHASE1 is the internal lower supply rail for UGATE1.
-
EPAD
EPAD at ground potential. Solder it directly to GND plane for better thermal performance.
4
FN7663.1
February 24, 2012
Block Diagram
PGOOD
BOOT1
VIN VCC_5V
EN23
BOOT2
VCC_5V
VCC_5V
UGATE1
UGATE2
PHASE1
PHASE2
ADAPTIVE DEAD-TIME
ADAPTIVE DEAD-TIME
V/I SAMPLE TIMING
V/I SAMPLE TIMING
VCC_5V
VCC_5V
LGATE1
LGATE2
5
PGND
POR
PGND
ENABLE
PGND
BOOT3
BIAS SUPPLIES
EN/SS1
VCC_5V
REFERENCE
UGATE3
FAULT LATCH
PHASE3
EN23
SOFT-START
ADAPTIVE DEAD-TIME
V/I SAMPLE TIMING
(Note 6)
FB1
180kΩ
1000kΩ
VCC_5V
LGATE3
15pF
OCP
+
+ 0.7V
REF
EN/SS1
ERROR AMP 1
_
PWM1
OC1 OC2 OC3
UV/OV
+
PWM3
FB1 FB2 FB3
0.7V REF
TK/SS3
ERROR AMP 3
OC3
EN/SS1
1.3V
VIN
ISEN1
FB3
_
+
1.55µA
+
PGND
16kΩ
VCC_5V
ISEN3
MINIMUM
SOFT-START
OCSET3
_
CURRENT
SAMPLE
+
DUTY CYCLE RAMP GENERATOR
CURRENT
SAMPLE
CHANNEL 3
PWM CHANNEL PHASE CONTROL
OCSET1
FB2
PWM2
+
1.75V REFERENCE
TK/SS2
CHANNEL 1
SAME STATE FOR
2 CLOCK CYCLES
REQUIRED TO LATCH
OVERCURRENT FAULT
ISEN2
OC2
RT
FN7663.1
February 24, 2012
+
OC1
MODE/SYNC
-
CHANNEL 2
SGND
OCSET2
ISL9443
_
ISL9443
Typical Application
+12V
+
C1
1µF
C2
4.7µF
3
VIN
C3
CIN1
100µF
C4
10µF
2
VCC_5V
10µF
31
C5
0.1µF
VOUT1
+3.3V, 6A
CO1
47µF
R1
115k
L1
R3
1.5µH
1.3k
BOOT1
30
UGATE1
32
PHASE1
1
29
BOOT2
UGATE2
PHASE2
ISEN1
ISEN2
LGATE1
LGATE2
26
C6
0.1µF
27
25
24
R4
L2
1.1k
1.5µH
28
5
FB1
FB2
C8
1000pF
12
V VOUT1
R2
30.9k
R13
100k
+12V
8
PGOOD
9
BOOT3
EN23
UGATE3
R7
100k
6
R8
100k
11
R9
100k
14
CSS1
10nF
VOUT2
R14
V
25.5k
6
PHASE3
OCSET1
ISEN3
OCSET2
CSS2
10nF 13
R15
49.9k
17
21
C11
0.1µF
19
VOUT3
L3
R10
1.0µH
1.3K
OCSET3
LGATE3
Q3
IRF7907
22
R11
15.8k
EN/SS1
FB3
16
C10
10µF
20
18
R5
16.5k
R6
10.5k
ISL9443
PGOOD
CO2
47µF
Q2
IRF7907
Q1
IRF7907
C7
47pF
VOUT2
+1.8V, 6A
15
+1.05V, 6A
CO3
100µF
C12
470pF
TK/SS3
R12
31.6k
TK/SS2
RT
MODE/SYNC
7
RT
49.9k
6
PGND
SGND
23
10
FN7663.1
February 24, 2012
ISL9443
Typical Application
+12V
C10
0.1µF
+ CIN3 + CIN2 +
150µF
150µF
CIN1
150µF
C2
4.7µF
CIN5
10µF
CIN4
10µF
3
VIN
31
30
BSC057N03LS
32
C5
C6
L1
+
BOOT2
BOOT1
Q2
C3
0.22µF
VOUT1
+0.9V, 25A
2
R4
230nH
Q9
Q2
1
UGATE2
UGATE1
PHASE2
ISEN2
29
3x100µF 3x270µF
C4
0.22µF
Q3
BSC057N03LS
27
25
24
LGATE2
LGATE1
28
VOUT2
+ 1.0V, 25A
L2
R5
2k
2k
230nH
Q8
CFF1
2200pF
CFF2
BSZ019N03LS
FB2
5
R2
35.7k
+ C8
BSZ019N03LS
Q4
2200pF
R1
10.2k
10µF
26
PHASE1
ISEN1
CIN6
CIN7
10µF
VCC_5V
C7
3x270µF 3x100µF
R8
10.7k
12
CP2
ISL9443
5600pF
FB1
R11
24.9k
+12V
CP1
5600pF
17
V +12V
R14
64.9k
MODE/SYNC
BOOT3
21 TK/SS3
UGATE3
R15
10k
VOUT1
9
PHASE3
EN23
ISEN3
13
CSS2
47nF
4
CSS1
47nF
21
LGATE3
19
18
22
VOUT3
+0.9V,25A
L3
R8
230nH
Q6
BSZ019N03LS
EN/SS1
C9
0.22µF
Q5
909
TK/SS2
CIN8
10µF
CIN9
10µF
20
BSC057N03LS
+ C10
C11
3x270µF 3x100µF
Q10
CFF3
2200pF
R3
10.2k
FB3 15
CP3
5600pF
R6
35.7k
VCC_5V
V
R11
100k
6
R12
100k
11
R13
100k
14 OCSET3
RPG
100k
OCSET1
OCSET2
PGOOD
PGND SGND
23
10
8
PGOOD
RT
7
RT
78.7k (Fsw = 400kHz)
7
FN7663.1
February 24, 2012
ISL9443
Table of Contents
Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Recommended Operating Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Typical Performance Curves. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Functional Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
General Description. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Input Voltage Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Internal 5V Linear Regulator (VCC_5V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Enable Signals and Soft-Start Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Output Voltage Programming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Tracking Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Light Load Efficiency Enhancement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Pre-biased Power-Up . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16
Frequency Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Frequency Synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16
Out-of-Phase Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16
Gate Control Logic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Gate Drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Adaptive Dead-Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Internal Bootstrap Diode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Power-Good Indicator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Protection Circuits. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Undervoltage Lockout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Overcurrent Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Overvoltage Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Over-Temperature Protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
17
18
18
18
Feedback Loop Compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Layout Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Layout Considerations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
General PowerPAD Design Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Component Selection Guideline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
MOSFET Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output Inductor Selection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output Capacitor Selection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input Capacitor Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
20
20
20
21
Revision History. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
8
FN7663.1
February 24, 2012
ISL9443
Absolute Maximum Ratings
Thermal Information
VCC_5V to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6.2V
VIN to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +30V
BOOT1,2,3/UGATE1,2,3 to PHASE1,2,3 . . . . . . . . . -0.3V to VCC_5V+0.3V
PHASE1,2,3 and
ISEN1,2,3, to GND . . . . . . . . . . . -5V (<100ns, 10µJ)/-0.3V (DC) to +30V
EN/SS1,EN23, FB1, FB2, FB3, to GND . . . . . . . . . . -0.3V to VCC_5V+0.3V
OCSET1, OCSET2, OCSET3, TKSS2, TKSS3
LGATE1, LGATE2, LGATE3, to GND . . . . . . . . . . . . -0.3V to VCC_5V+0.3V
RT, MODE/SYNC to GND . . . . . . . . . . . . . . . . . . . . . . -0.3V to VCC_5V+0.3V
PGOOD to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +VCC_5V + 0.3V
VCC_5V Short Circuit to GND Duration. . . . . . . . . . . . . . . . . . . . . . . . . . . . .1s
ESD Rating
Human Body Model (Tested per JESD22-A114F) . . . . . . . . . . . . . . 3000V
Machine Model (Tested per JESD22-115-C) . . . . . . . . . . . . . . . . . . . 200V
Charge Device Model (Tested per JESD22-C110D) . . . . . . . . . . . . 2000V
Latch Up (Tested per JESD78C; Class II, Level A, +85°C) . . . . . . . . 100mA
Thermal Resistance (Typical)
θJA (°C/W) θJC (°C/W)
32 Ld QFN Package (Notes 4, 5) . . . . . . . .
31
2.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . .-55°C to +150°C
Maximum Operating Temperature . . . . . . . . . . . . . . . . . . . . -40°C to +85°C
Maximum Storage Temperature. . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5V to 28V
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTE:
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application
Schematic. VIN = 5.0V to 28V, or VCC_5V = 5V ±10%, C_VCC_5V = 4.7µF, TA = -40°C to +85°C (Note 9), Typical values are at TA = +25°C, unless otherwise
specified. Boldface limits apply over the operating temperature range, -40°C to +85°C
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
(Note 9)
TYP
MAX
(Note 9)
UNITS
4.5
12.0
28.0
V
VIN SUPPLY
VIN
Input Voltage Range
VIN SUPPLY CURRENT
IVINQ
Shutdown Current (Note 7)
EN/SS1 = EN23 = 0V,
PGOOD is floating
30
38
µA
IVINOP
Operating Current (Note 8)
EN/SS1, EN23, PGOOD are floating
5
6
mA
5.7
V
VCC_5V SUPPLY (Note 6)
VCC
IVCC_MAX
Operation Voltage
VIN = 12V, IL = 0mA
5.1
5.4
Internal LDO Output Voltage
VIN = 4.5V, IL = 30mA
4.05
4.35
V
Internal LDO Output Voltage
VIN > 5.6V, IL = 75mA
4.5
5.4
V
Maximum Supply Current of Internal LDO
VVCC_5V = 0V, VIN = 12V
150
250
mA
UNDERVOLTAGE LOCKOUT
VUVLOTHR
Undervoltage Lockout, Rising
VCC_5V Voltage
3.4
3.95
4.45
V
VUVLOTHF
Undervoltage Lockout, Falling
VCC_5V Voltage
3.05
3.60
4.15
V
EN/SS1, EN23 THRESHOLD
VENSS_TH
EN/SS1 Threshold
1.1
1.3
1.5
V
VEN_THR
EN23 Logic Threshold, Rising
1.4
1.7
2.0
V
VEN_THF
EN23 Logic Threshold, Falling
1.1
1.25
1.4
V
1.05
1.55
2.05
µA
SOFT-START CURRENT
ISS
EN/SS1, TK/SSx Soft-Start Charge Current
9
VEN/SS1 = VTK/SSx = 0V
FN7663.1
February 24, 2012
ISL9443
Electrical Specifications
Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application
Schematic. VIN = 5.0V to 28V, or VCC_5V = 5V ±10%, C_VCC_5V = 4.7µF, TA = -40°C to +85°C (Note 9), Typical values are at TA = +25°C, unless otherwise
specified. Boldface limits apply over the operating temperature range, -40°C to +85°C (Continued)
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
(Note 9)
TYP
MAX
(Note 9)
UNITS
1.3
2.1
2.9
ms
DEFAULT INTERNAL MINIMUM SOFT-STARTING FOR PWM2 AND PWM3
tSS_MIN
Default Internal Output Ramping Time
POWER-GOOD MONITORS
VPGOV
PGOOD Upper Threshold, PWM 1, 2 and 3
105.5
111
115.5
%
VPGUV
PGOOD Lower Threshold, PWM 1, 2 and 3
85
89
94
%
VPGLOW
PGOOD Low Level Voltage
I_SINK = 2mA
0.3
V
IPGLKG
PGOOD Leakage Current
PGOOD = 5V
150
nA
PGOOD Rise Time
RPULLUP = 10k to 3.3V
0.05
µs
PGOOD Fall Time
RPULLUP = 10k to 3.3V
0.05
µs
1
PGOOD TIMING
tPGR
VOUT Rising Threshold to PGOOD Rising
0.7
1.1
1.5
ms
tPGF
VOUT Falling Threshold to PGOOD Falling
40
75
110
µs
REFERENCE SECTION
VREF
IFBLKG
Internal Reference Voltage
Across specified temperature range
Reference Voltage Accuracy
TA = 0°C to +85°C
-1.0
0.7
+1.0
%
TA = -40°C to +85°C
-1.15
+1.0
%
100
nA
FB Bias Current (Note 10)
V
PWM CONTROLLER ERROR AMPLIFIERS
DC Gain (Note 10)
88
dB
GBW
Gain-BW Product (Note 10)
15
MHz
SR
Slew Rate (Note 10)
2.0
V/µs
PWM REGULATOR
tOFF_MIN
Minimum Off Time
RFS = 169kΩ
95
ΔVRAMP
Peak-to-Peak Saw-tooth Amplitude (Note 9)
VIN = 12V
1.2
V
VIN = 5.0V
0.55
V
1
V
Ramp Offset
125
155
ns
SWITCHING FREQUENCY (Note 10)
FSW
VRT
Switching Frequency
RT = 20.5kΩ
1080
1200
1320
kHz
Switching Frequency
RT = 169kΩ
168
198
228
kHz
Switching Frequency
RT = 49.9kΩ
540
600
660
kHz
RT Voltage
RT = 49.9kΩ
485
500
515
mV
RT = 49.9kΩ
1020
1380
kHz
SYNCHRONIZATION
FSYNC
SYNC Synchronization Range
LIGHT LOAD EFFICIENCY MODE
VMODETHH
MODE/SYNC Threshold High
1.3
1.6
1.9
V
VMODETHL
MODE/SYNC Threshold Low
1.1
1.4
1.7
V
VCROSS
Diode Emulation Phase Threshold (Note 11)
10
VIN = 12V
-3
mV
FN7663.1
February 24, 2012
ISL9443
Electrical Specifications
Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application
Schematic. VIN = 5.0V to 28V, or VCC_5V = 5V ±10%, C_VCC_5V = 4.7µF, TA = -40°C to +85°C (Note 9), Typical values are at TA = +25°C, unless otherwise
specified. Boldface limits apply over the operating temperature range, -40°C to +85°C (Continued)
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
(Note 9)
TYP
MAX
(Note 9)
UNITS
PWM GATE DRIVER (Note 10)
IGSRC
Source Current
800
mA
IGSNK
Sink Current
2000
mA
RUG_UP
Upper Drive Pull-Up
VCC_5V = 5.0V
1.5
3
Ω
RUG_DN
Upper Drive Pull-Down
VCC_5V = 5.0V
1.1
2.5
Ω
RLG_UP
Lower Drive Pull-Up
VCC_5V = 5.0V
1.5
3
Ω
RLG_DN
Lower Drive Pull-Down
VCC_5V = 5.0V
0.6
1.5
Ω
tGR
Rise Time
COUT = 1000pF
8
ns
tGF
Fall Time
COUT = 1000pF
10
ns
OVERVOLTAGE PROTECTION
VOVTH
OV Trip Point
114.5
118.5
123.5
%
OVERCURRENT PROTECTION
IOCSET
Overcurrent Threshold (OCSET_) (Note 11)
ROCSET = 55kΩ
Full Scale Input Current (ISEN_) (Note 11)
VOCSET
Overcurrent Set Voltage (OCSET_)
1.67
32
µA
15
µA
1.74
1.81
V
OVER-TEMPERATURE (Note 9)
TOT-TH
Over-Temperature Shutdown
150
°C
TOT-HYS
Over-Temperature Hysteresis
15
°C
NOTES:
6. In normal operation, where the device is supplied with voltage on the VIN pin, the VCC_5V pin provides a 5V output capable of 75mA (min). When the
VCC_5V pin is connected to external 5V supply, the internal LDO regulator is disabled. The voltage at VCC_5V should not exceed the voltage at VIN
at any time. (Refer to the “Pin Descriptions” on page 2 for more details.)
7. This is the total shutdown current with VIN = 5.6V and 28V.
8. Operating current is the supply current consumed when the device is active but not switching. It does not include gate drive current.
9. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
10. Check Note 6 for VCC_5V and VIN configurations.
11. Threshold voltage at PHASE1, PHASE2, PHASE3 pins for turning off the bottom MOSFET during DEM.
11
FN7663.1
February 24, 2012
ISL9443
Typical Performance Curves
Oscilloscope plots are taken using the ISL9443EVAL1Z Evaluation Board,
VIN = 12V, VOUT1 = 0.9V, VOUT2 = 1.0V, VOUT3 = 0.9V unless otherwise noted.
5.3
OPERATING CURRENT (mA)
SHUTDOWN CURRENT (µA)
33.0
32.5
32.0
31.5
31.0
30.5
30.0
-40
-20
0
20
40
60
TEMPERATURE (°C)
80
3
2
1
150
200
250
VCC5V LOAD CURRENT (mA)
NORMALIZED OUTPUT VOLTAGE (%)
4.5
-20
0
20
40
60
80
100
1.6
CHANNELS 1 AND 3
1.2
CHANNEL 2
0.8
0.4
0
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
SOFT-START PIN VOLTAGE (V)
FIGURE 4. VCC5V LOAD REGULATION
120
SOFT-START CHARGING CURRENT (µA)
VCC5 VOLTAGE (V)
4
100
VIN = 4.5V
FIGURE 3. QUIESCENT CURRENT vs TEMPERATURE
5
50
4.7
TEMPERATURE (°C)
6
0
4.9
4.3
-40
100
FIGURE 2. SHUTDOWN CURRENT vs TEMPERATURE
0
VIN = 28V
5.1
FIGURE 5. SOFT-START PIN CHARGING CURRENT vs VOLTAGE ON
SOFT-START PIN
CHANNELS 2 AND 3
PWM1
100
80
PWM2
60
CHANNEL 1
PWM3
40
20
TIME AT 1µs/DIV
0
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
SOFT-START PIN VOLTAGE (V)
FIGURE 6. NORMALIZED OUTPUT VOLTAGE VS VOLTAGE ON
SOFT-START PIN
12
FIGURE 7. PHASE NODE WAVEFORMS
FN7663.1
February 24, 2012
ISL9443
Typical Performance Curves
Oscilloscope plots are taken using the ISL9443EVAL1Z Evaluation Board,
VIN = 12V, VOUT1 = 0.9V, VOUT2 = 1.0V, VOUT3 = 0.9V unless otherwise noted. (Continued)
720
REFERENCE VOLTAGE (mV)
SWITCHING FREQUENCY (kHz)
650
630
610
590
570
550
-40
-20
0
20
40
60
TEMPERATURE (°C)
80
VOUT1
0.898
40
30
20
0.894
10
0
0
5
10
15
20
0.890
25
INPUT CURRENT (A)
EFFICIENCY (%)
0.902
PWM1 OUTPUT VOLTAGE (V)
0.906
70
50
-20
0
20
40
60
TEMPERATURE (°C)
80
100
10
0.910
80
60
690
FIGURE 9. REFERENCE VOLTAGE vs TEMPERATURE
EFFICIENCY @ CCM (%)
90
700
680
-40
100
FIGURE 8. SWITCHING FREQUENCY vs TEMPERATURE
(RT = 49.9 kΩ)
100
710
1
CCM
DEM
0.1
0.01
0.001
0.01
0.1
1
10
LOAD CURRENT (A)
LOAD CURRENT (A)
FIGURE 10. PWM1 EFFICIENCY AND LOAD REGULATION
VOUT1 AT 500mV/DIV
FIGURE 11. PWM1 INPUT CURRENT COMPARISON WITH
MODE = CCM/DEM
VOUT2 AT 1V/DIV
TKSS2 AT 2V/DIV
ENSS1 AT 1V/DIV
VOUT2 AT 1V/DIV
TKSS3 AT 2V/DIV
TIME AT 10ms/DIV
FIGURE 12. PWM1 START-UP WAVEFORM
13
TIME AT 10ms/DIV
FIGURE 13. PWM2 AND PWM3 START-UP WAVEFORMS
FN7663.1
February 24, 2012
ISL9443
Typical Performance Curves
Oscilloscope plots are taken using the ISL9443EVAL1Z Evaluation Board,
VIN = 12V, VOUT1 = 0.9V, VOUT2 = 1.0V, VOUT3 = 0.9V unless otherwise noted. (Continued)
VOUT1 AT 500mV/DIV
VOUT1 AT 500mV/DIV
VOUT2 AT 500mV/DIV
VOUT2 AT 500mV/DIV
VOUT3 AT 500mV/DIV
VOUT3 AT 500mV/DIV
PGOOD AT 5V/DIV
TIME AT 10ms/DIV
FIGURE 14. PGOOD RISING WAVEFORM
PGOOD AT 5V/DIV
TIME AT 10ms/DIV
FIGURE 15. PRE-BIASED START-UP WAVEFORM
VOUT1 AT 20mV/DIV, LOAD = 0mA
VOUT1 AT 20mV/DIV, LOAD = 0mA
TIME AT 10ms/DIV
TIME AT 2µs/DIV
VOUT1 AT 20mV/DIV, LOAD = 100mA
VOUT1 AT 20mV/DIV, LOAD = 100mA
TIME AT 10µs/DIV
TIME AT 2µs/DIV
VOUT1 AT 20mV/DIV, LOAD = 10A
VOUT1 AT 20mV/DIV, LOAD = 10A
TIME AT 2µs/DIV
TIME AT 2µs/DIV
FIGURE 16. PWM1 OUTPUT RIPPLE. MODE = 0V (DEM)
VOUT1 AT 100mV/DIV
FIGURE 17. PWM1 OUTPUT RIPPLE. MODE = 5V (CCM)
VOUT1 AT 1V/DIV
VOUT2 AT 100mV/DIV
ENSS1 AT 5V/DIV
VOUT3 AT 100mV/DIV
OUTPUT CURRENT AT 10A/DIV
15A
5A
5A
TIME AT 20µs/DIV
FIGURE 18. PWM1 LOAD TRANSIENT RESPONSE
14
PGOOD AT 5V/DIV
TIME AT 50ms/DIV
FIGURE 19. PWM1 OCP RESPONSE, OUTPUT SHORT CIRCUITED TO
GROUND
FN7663.1
February 24, 2012
ISL9443
Functional Description
The internal LDO has an overcurrent limit of typically 150mA.
For better efficiency, connect VCC_5V to VIN for 5V ±10% input
applications.
General Description
The ISL9443 integrates control circuits for three synchronous
buck converters. The three synchronous bucks operate
out-of-phase to substantially reduce the input ripple and thus
reduce the input filter requirements.
Each part has separate enable/disable control lines (EN/SS1,
EN23), which provide flexible power-up sequencing. The
soft-start time is programmable individually by adjusting the
soft-start capacitors connected from EN/SS1, TK/SS2 and
TK/SS3 respectively.
The valley current mode control scheme with input voltage
feed-forward ramp simplifies loop compensation and provides
excellent rejection to input voltage variation.
Input Voltage Range
The ISL9443 is designed to operate from input supplies
ranging from 4.5V to 28V.
The input voltage range can be effectively limited by the
available minimum PWM off time.
V OUT + V d1
⎛
⎞
V IN ( min ) = ⎜ -----------------------------------------------------------------------⎟ + V d2 – V d1
1
–
t
⎝
OFF ( min ) × Frequency⎠
(EQ. 2)
Where,
Vd1 = sum of the parasitic voltage drops in the inductor
discharge path, including the lower FET, inductor and PC board.
Vd2 = sum of the voltage drops in the charging path, including
the upper FET, inductor and PC board resistances.
The maximum input voltage and minimum output voltage is
limited by the minimum on-time tON(min).
V OUT
⎛
⎞
V IN ( max ) ≤ V OUT x ⎜ ------------------------------------------------------------⎟
t
×
Frequency
⎝ ON ( min )
⎠
(EQ. 3)
Where tON(min) = 100ns
Internal 5V Linear Regulator (VCC_5V)
All ISL9443 functions are internally powered from an on-chip,
low dropout 5V regulator. Bypass the linear regulator’s output
(VCC_5V) with a 4.7µF capacitor to the power ground. The
ISL9443 also employs an undervoltage lockout circuit that
disables all regulators when VCC_5V falls below 3.6V.
The internal LDO can source over 75mA to supply the IC, power
the low side gate drivers and charge the boot capacitors. When
driving large FETs at high switching frequency, little or no
regulator current may be available for external loads.
For example, a single large FET with 15nC total gate charge
requires 15nC x 300kHz = 4.5mA (15nC x 600kHz = 9mA).
Also, at higher input voltages with larger FETs, the power
dissipation across the internal 5V will increase. Excessive
dissipation across this regulator must be avoided to prevent
junction temperature rise. Thermal protection may be
triggered if die temperature increases above +150°C due to
excessive power dissipation.
Enable Signals and Soft-Start Operation
Typical applications for the ISL9443 use programmable
analog soft-start or the TK/SSx pins for tracking. The soft-start
time can be set by the value of the soft-start capacitors
connected from the EN/SS1 for PWM1 to ground and from
TK/SSx pins to ground for PWM2 and PWM3. Inrush current
during start-up can be alleviated by adjusting the soft-start
time.
After the VCC_5V pin reaches the UVLO threshold, the ISL9443
PWM1 soft-start circuitry becomes active. The internal 1.55µA
charge current begins charging up the soft-start capacitor
connected from the EN/SS1 pin to GND. The PWM1 output
remains inactive until voltage on the EN/SS1 pin reaches 1.3V.
As the voltage on the EN/SS1 pin rises from 1.3V to 2V, the
PWM1 reference voltage is clamped to the voltage on the
EN/SS1 pin minus 1.3V. PWM1 output voltage thus rises from
0V to regulation as EN/SS1 rises from 1.3V to 2V. Charging of
the soft-start capacitor continues until the voltage on the
EN/SS1 pin reaches 3.5V.
Power sequencing can be achieved by using the EN23 and
TK/SSx pins. When the EN23 pin is pulled high, the internal
1.55µA charge current begins charging up the soft-start
capacitor connected from the TK/SSx pin to GND. The
respective reference voltage is clamped to the voltage on the
TK/SSx pin. Thus, PWM2 and PWM3 output voltages ramp
from 0V to regulation as voltage on TK/SS2 and TK/SS3 goes
up from 0V to 0.7V. Charging of the soft-start capacitors
continues until the voltage on the TK/SSx reaches 3.5V.
The typical soft-start time is set according to Equation 4:
C SSx
t SSx = 0.7V ⎛ --------------------⎞
⎝ 1.55μA⎠
(EQ. 4)
For PWM2 and PWM3, when the soft-start time set by external
CSS or tracking is less than 2ms, an internal soft-start circuit of
2ms takes over the soft-start. There is no internal soft-start for
PWM1.
PGOOD will toggle to high when all the outputs are up and in
regulation.
Pulling the EN23 low disables the PWM2 and PWM3 channels.
The TK/SSx pin will also be discharged to GND by internal
MOSFETs.
Output Voltage Programming
The ISL9443 provides a precision internal reference voltage to
set the output voltage. Based on this internal reference, the
output voltage can thus be set from 0.7V up to a level
determined by the input voltage, the maximum duty cycle, and
the conversion efficiency of the circuit.
A resistive divider from the output to ground sets the output
voltage of any PWM channel. The center point of the divider
shall be connected to the FBx pin. The output voltage value is
15
FN7663.1
February 24, 2012
ISL9443
determined by Equation 5.
R1 + R2
V OUTx = 0.7V ⎛ ----------------------⎞
⎝ R2 ⎠
(EQ. 5)
where R1 is the top resistor of the feedback divider network
and R2 is the bottom resistor connected from FBx to ground.
set by a resistor connected from the RT pin to GND according
to Equation 1.
Frequency setting curve shown in Figure 20 assists in selecting
the correct value for RT.
1250
Tracking Operation
To minimize the impact of the 1.55µA soft-start current on the
tracking function, it is recommended to use resistors of less
than 10kΩ for the tracking resistive dividers.
When overcurrent-protection (OCP) is triggered for the slave
PWM channel, the internal minimum soft-start circuit
determines the OCP soft-start hiccup.
Light Load Efficiency Enhancement
When MODE/SYNC pin is tied to GND, the ISL9443 operates in
high efficiency diode emulation mode and pulse skipping
mode in light load condition. The inductor current is not
allowed to reverse (discontinuous operation). At very light
loads, the converter goes into diode emulation and triggers the
pulse skipping function. Here, the upper MOSFET remains off
until the output voltage drops to the point the error amplifier
output goes above the pulse skipping mode threshold.
The minimum tON in the pulse skipping mode is 80ns, please
select the switching frequency so the PWM tON is greater than
80ns at maximum VIN at no load.
Pre-biased Power-Up
The ISL9443 has the ability to soft-start with a pre-biased
output. The output voltage would not be yanked down during
pre-biased start-up. The PWM is not active until the soft-start
ramp reaches the output voltage times the resistive divider
ratio.
Overvoltage protection is alive during soft-starting.
Frequency Selection
Switching frequency selection is a trade-off between efficiency
and component size. Low switching frequency improves
efficiency by reducing MOSFET switching loss. To meet output
ripple and load transient requirements, operation at a low
switching frequency would require larger inductance and
output capacitance. The switching frequency of the ISL9443 is
16
1000
FREQUENCY (kHz)
The PWM2 and PWM3 of the ISL9443 can be independently
set up to track the output of another PWM or an external
supply. In the following discussion, we refer to the voltage rail
to be tracked as the master rail while we refer to the voltage
rail that follows the master as the slave rail. To implement
tracking, an additional resistive divider is connected between
the master rail and ground. The center point of the divider shall
be connected to the TK/SSx pin of the slave PWM. The
resistive divider ratio sets the ramping ratio between the two
voltage rails. To implement coincident tracking, set the
tracking resistive divider ratio exactly the same as the slave
rail output resistive divider given by Equation 5. Make sure that
the voltage at TK/SSx is greater than 0.7V when the master
rail reaches regulation.
750
500
250
0
0
20
40
60
80
100
RT (kΩ)
120
140
160
180
FIGURE 20. RT vs SWITCHING FREQUENCY
Frequency Synchronization
The MODE/SYNC pin may be used to synchronize the ISL9443
with an external clock.
When the MODE/SYNC pin is connected to an external clock,
the ISL9443 will synchronize to this external clock at half of
the clock frequency. For proper operation, the frequency setting
resistor, RT, should be set according to Equation 1.
When frequency synchronization is in action, the controllers
will enter forced continuous current mode at light load.
Out-of-Phase Operation
To reduce input ripple current, the three PWM channels operate
180° out-of-phase. This reduces the input capacitor ripple current
requirements, reduces power supply-induced noise, and improves
EMI. This effectively helps to lower component cost, save board
space and reduce EMI.
Triple PWMs traditionally operate in-phase and turn on all three
upper FETs at the same time. The input capacitor must then
support the instantaneous current requirements of the three
switching regulators simultaneously, resulting in increased ripple
voltage and current. The higher RMS ripple current lowers the
efficiency due to the power loss associated with the ESR of the
input capacitor. This typically requires more low-ESR capacitors in
parallel to minimize the input voltage ripple and ESR-related
losses, or to meet the required ripple current specification.
With synchronized out-of-phase operation, the high-side MOSFETs
turn off 180° out-of-phase. The instantaneous input current
peaks of both regulators no longer overlap, resulting in reduced
RMS ripple current and input voltage ripple. This reduces the
required input capacitor ripple current rating, allowing fewer or
less expensive capacitors, and reducing the shielding
requirements for EMI. The typical operating curves show the
synchronized 180° out-of-phase operation.
FN7663.1
February 24, 2012
ISL9443
Gate Control Logic
Adaptive Dead-Time
The gate control logic translates generated PWM signals into
gate drive signals providing amplification, level shifting and
shoot-through protection. The gate drivers have circuitry that
helps optimize the IC performance over a wide range of
operational conditions. As MOSFET switching times can vary
dramatically from type to type and with input voltage, the gate
control logic provides adaptive dead-time by monitoring real gate
waveforms of both the upper and lower MOSFETs. Shoot-through
control logic provides a 16ns dead-time to ensure that both the
upper and lower MOSFETs will not turn on simultaneously
causing a shoot-through condition.
The ISL9443 incorporates an adaptive dead-time algorithm on
the synchronous buck PWM controllers that optimizes operation
with varying MOSFET conditions. This algorithm provides
approximately 16ns of dead-time between switching the upper
and lower MOSFET’s. This dead-time is adaptive and allows
operation with different MOSFET’s without having to externally
adjust the dead-time using a resistor or capacitor. During turn-off
of the lower MOSFET, the LGATE voltage is monitored until it
reaches a threshold of 1V, at which time the UGATE is released to
rise. Adaptive dead-time circuitry monitors the upper MOSFET
gate voltage during UGATE turn-off. Once the upper MOSFET
gate-to-source voltage has dropped below a threshold of 1V, the
LGATE is allowed to rise.
Gate Drivers
The low-side gate drivers are supplied from VCC_5V and provide
a peak sink current of 2A and source current of 800mA for each
PWM channel. The high-side gate drivers are also capable of
delivering the same currents as the low-side gate drivers.
Gate-drive voltage for the upper N-Channel MOSFETs are
generated by flying capacitor boot circuits. A boot capacitor
connected from the BOOT pin to the PHASE node provides power
to the high-side MOSFET driver. To limit the peak current in the IC,
an external resistor may be placed between the BOOT pin and the
boot capacitor. This small series resistor also damps any
oscillations caused by the resonant tank of the parasitic
inductances in the traces of the board and the FET’s input
capacitance.
At start-up, the low-side MOSFET turns on first and forces PHASE
to ground in order to charge the BOOT capacitor to 5V. After the
low-side MOSFET turns off, the high-side MOSFET is turned on by
closing an internal switch between BOOT and UGATE. This
provides the necessary gate-to-source voltage to turn on the
upper MOSFET, an action that boosts the 5V gate drive signal
above VIN. The current required to drive the upper MOSFET is
drawn from the internal 5V regulator.
For optimal EMI performance or reducing phase node ringing, a
small resistor might be placed between these pins to the positive
terminal of the bootstrap capacitors.
VCC_5V
BOOT
OPTIONAL
EXTERNAL
SCHOTTKY
VIN
The ISL9443 has integrated bootstrap diodes to help reduce total
cost and reduce layout complexity. Simply adding an external
capacitor across the BOOT and PHASE pins completes the
bootstrap circuit. The bootstrap capacitor must have a maximum
voltage rating above the maximum input voltage plus 5V. The
bootstrap capacitor can be chosen from Equation 6.
Q GATE
C BOOT ≥ --------------------ΔV BOOT
(EQ. 6)
Where QGATE is the amount of gate charge required to fully
charge the gate of the upper MOSFET. The ΔVBOOT term is defined
as the allowable droop in the rail of the upper drive.
As an example, suppose an upper MOSFET has a gate charge
(QGATE) of 25nC at 5V and also assume the droop in the drive
voltage over a PWM cycle is 200mV. One will find that a
bootstrap capacitance of at least 0.125µF is required. The next
larger standard value capacitance of 0.22µF should be used. A
good quality ceramic capacitor is recommended.
The internal bootstrap Schottky diodes have a resistance of 1.5Ω
(typ) at 800mA. Combined with the resistance RBOOT, this could
lead to the boot capacitor charging insufficiently in cases where
the bottom MOSFET is turned on for a very short time. If such
circumstances are expected, an additional external Schottky
diode may be added from VCC_5V to the positive of the boot
capacitor. RBOOT may still be necessary to lower EMI due to fast
turn-on of the upper MOSFET.
Power-Good Indicator
RBOOT
CB
UGATE
Internal Bootstrap Diode
PHASE
The PGOOD pin can be used to indicate the status of the outputs.
PGOOD will be true (open drain) when all three FB pins are within
±11% of the internal voltage reference.
Protection Circuits
The converter outputs are monitored and protected against
overload, short circuit and undervoltage conditions.
ISL9443
FIGURE 21. UPPER GATE DRIVER CIRCUIT
Undervoltage Lockout
The ISL9443 includes UVLO protection which keeps the device in
a reset condition until a proper operating voltage is applied. It
also shuts down the ISL9443 if the operating voltage drops
below a pre-defined value. All controllers are disabled when UVLO
17
FN7663.1
February 24, 2012
ISL9443
is asserted. When UVLO is asserted, PGOOD is valid and will be
de-asserted.
Overcurrent Protection
All the PWM controllers use the lower MOSFET's on-resistance,
rDS(ON) , to monitor the current in the converter. The sensed
voltage drop is compared with a threshold set by a resistor
connected from the OCSETx pin to ground.
( 7 ) ( R CS )
R OCSET = --------------------------------------( I OC ) ( r DS ( ON ) )
(EQ. 7)
Where IOC is the desired overcurrent protection threshold, and
RCS is a value of the current sense resistor connected to the
ISENx pin.
If an overcurrent is detected for 2 consecutive clock cycles, the IC
enters a hiccup mode by turning off the gate drivers and entering
soft-start. The IC will cycle 5 times through soft-start before trying
to restart. The IC will continue to cycle through soft-start until the
overcurrent condition is removed. Hiccup mode is active during
soft-start so care must be taken to ensure that the peak inductor
current does not exceed the overcurrent threshold during
soft-start.
Because of the nature of this current sensing technique, and to
accommodate a wide range of rDS(ON) variations, the value of the
overcurrent threshold should represent an overload current about
150% to 180% of the maximum operating current. If more
accurate current protection is desired, place a current sense
resistor in series with the lower MOSFET source.
When OCP is triggered the EN/SS1 or TK/SSx pins are pulled to
ground by internal MOSFET. For PWM rails configured to track
another voltage rail the TK/SSx pin rises up much faster than the
internal minimum soft-start ramp. The voltage reference will
then be clamped to the internal minimum soft-start ramp. Thus,
smooth soft-start hiccup is achieved even with the tracking
function.
Overvoltage Protection
All switching controllers within the ISL9443 have fixed
overvoltage set points. The overvoltage set point is set at 118%
of the nominal output voltage, the output voltage set by the
feedback resistors. In the case of an overvoltage event, the IC will
attempt to bring the output voltage back into regulation by
keeping the upper MOSFET turned off and modulating the lower
MOSFET for 2 consecutive PWM cycles. If the overvoltage
condition has not been corrected in 2 cycles and the output
voltage is above 118% of the nominal output voltage, the
ISL9443 will turn off both the upper MOSFET and the lower
MOSFET. The ISL9443 will enter hiccup mode until the output
voltage return to 110% of the nominal output voltage.
Over-Temperature Protection
The IC incorporates an over-temperature protection circuit that
shuts the IC down when a die temperature of +150°C is
reached. Normal operation resumes when the die temperatures
drops below +130°C through the initiation of a full soft-start
cycle. When all the three channels are disabled, thermal
protection is inactive. This helps achieve a very low shutdown
current of 33µA.
18
Feedback Loop Compensation
To reduce the number of external components and to simplify the
process of determining compensation components, all PWM
controllers have internally compensated error amplifiers. To make
internal compensation possible, several design measures were
taken.
Firstly, the ramp signal applied to the PWM comparator is
proportional to the input voltage provided at the VIN pin. This
keeps the modulator gain constant with varying input voltages.
Secondly, the load current proportional signal is derived from the
voltage drop across the lower MOSFET during the PWM time
interval and is subtracted from the amplified error signal on the
comparator input. This creates an internal current control loop.
The resistor connected to the ISEN pin sets the gain in the current
feedback loop. The following expression estimates the required
value of the current sense resistor depending on the maximum
operating load current and the value of the MOSFET’s rDS(ON).
( I MAX ) ( r DS ( ON ) )
R CS ≥ -------------------------------------------30μA
(EQ. 8)
Choosing RCS to provide 30µA of current to the current sample
and hold circuitry is recommended but values down to 2µA and
up to 100µA can be used. A higher sampling current will help to
stabilize the loop.
Due to the current loop feedback, the modulator has a single
pole response with -20dB slope at a frequency determined by the
load.
1
F PO = -----------------------------2π ⋅ R O ⋅ C O
(EQ. 9)
Where RO is load resistance and CO is load capacitance. For this
type of modulator, a Type 2 compensation circuit is usually
sufficient.
Figure 22 shows a Type 2 amplifier and its response along with
the responses of the current mode modulator and the converter.
The Type 2 amplifier, in addition to the pole at origin, has a
zero-pole pair that causes a flat gain region at frequencies
between the zero and the pole.
1
F Z = ------------------------------ = 10kHz
2π ⋅ R 2 ⋅ C 1
(EQ. 10)
1
F P = ------------------------------ = 600kHz
2π ⋅ R 1 ⋅ C 2
(EQ. 11)
Zero frequency, amplifier high-frequency gain and modulator
gain are chosen to satisfy most typical applications. The
crossover frequency will appear at the point where the modulator
attenuation equals the amplifier high frequency gain. The only
task that the system designer has to complete is to specify the
output filter capacitors to position the load main pole
somewhere within one decade lower than the amplifier zero
frequency. With this type of compensation, plenty of phase
margin is easily achieved due to zero-pole pair phase ‘boost’.
FN7663.1
February 24, 2012
ISL9443
Layout Considerations
C2
R2
1. The input capacitors, upper FET, lower FET, inductor and
output capacitor should be placed first. Isolate these power
components on the topside of the board with their ground
terminals adjacent to one another. Place the input high
frequency decoupling ceramic capacitors very close to the
MOSFETs.
C1
CONVERTER
R1
EA
TYPE 2 EA
GM = 17.5dB
GEA = 18dB
MODULATOR
FZ
FPO
FP
FC
2. Use separate ground planes for power ground and small
signal ground. Connect the SGND and PGND together close to
the IC. Do not connect them together anywhere else.
3. The loop formed by the input capacitor, the top FET and the
bottom FET must be kept as small as possible.
4. Ensure the current paths from the input capacitor to the
MOSFET, to the output inductor and output capacitor are as
short as possible with maximum allowable trace widths.
FIGURE 22. FEEDBACK LOOP COMPENSATION
Conditional stability may occur only when the main load pole is
positioned too much to the left side on the frequency axis due to
excessive output filter capacitance. In this case, the ESR zero
placed within the 1.2kHz to 30kHz range gives some additional
phase ‘boost’. Some phase boost can also be achieved by
connecting capacitor CZ in parallel with the upper resistor R1 of
the divider that sets the output voltage value. Please refer to
“Input Capacitor Selection” on page 21 for further details.
Layout Guidelines
Careful attention to layout requirements is necessary for
successful implementation of an ISL9443 based DC/DC
converter. The ISL9443s switch at a very high frequency and
therefore the switching times are very short. At these switching
frequencies, even the shortest trace has significant impedance.
Also, the peak gate drive current rises significantly in an
extremely short time. Transition speed of the current from one
device to another causes voltage spikes across the
interconnecting impedances and parasitic circuit elements.
These voltage spikes can degrade efficiency, generate EMI,
increase device overvoltage stress and ringing. Careful
component selection and proper PC board layout minimizes the
magnitude of these voltage spikes.
There are three sets of critical components in a DC/DC converter
using the ISL9443: The controller, the switching power
components and the small signal components. The switching
power components are the most critical from a layout point of
view because they switch a large amount of energy, so they tend
to generate a large amount of noise. The critical small signal
components are those connected to sensitive nodes or those
supplying critical bias currents. A multi-layer printed circuit board
is recommended.
19
5. Place The PWM controller IC close to the lower FET. The LGATE
connection should be short and wide. The IC can be best
placed over a quiet ground area. Avoid switching ground loop
currents in this area.
6. Place the VCC_5V bypass capacitor very close to the VCC_5V
pin of the IC and connect its ground to the PGND plane.
7. Place the gate drive components - optional BOOT diode and
BOOT capacitors - together near the controller IC.
8. The output capacitors should be placed as close to the load as
possible. Use short wide copper regions to connect output
capacitors to load to avoid inductance and resistances.
9. Use copper filled polygons or wide but short trace to connect
the junction of the upper FET, Lower FET and output inductor.
Also, keep the PHASE node connection to the IC short. Do not
unnecessarily oversize the copper islands for PHASE node.
Since the phase nodes are subjected to very high dv/dt
voltages, the stray capacitor formed between these islands
and the surrounding circuitry will tend to couple switching
noise.
10. Route all high speed switching nodes away from the control
circuitry.
11. Create a separate small analog ground plane near the IC.
Connect the SGND pin to this plane. All small signal grounding
paths including feedback resistors, current limit setting
resistors, soft-starting capacitors and ENx pull-down resistors
should be connected to this SGND plane.
12. Separate current sensing traces from PHASE node
connections.
13. Ensure the feedback connection to the output capacitor is
short and direct.
FN7663.1
February 24, 2012
ISL9443
General PowerPAD Design Considerations
Output Inductor Selection
The following is an example of how to use vias to remove heat
form the IC.
The PWM converters require output inductors. The output
inductor is selected to meet the output voltage ripple
requirements. The inductor value determines the converter’s
ripple current and the ripple voltage is a function of the ripple
current and the output capacitor(s) ESR. The ripple voltage
expression is given in the capacitor selection section and the
ripple current is approximated by Equation 14:
( V IN – V OUT ) ( V OUT )
ΔI L = --------------------------------------------------( f S ) ( L ) ( V IN )
(EQ. 14)
Output Capacitor Selection
FIGURE 23. PCB VIA PATTERN
It is recommended to fill the thermal pad area with vias. A typical
via array fills the thermal pad foot print such that their centers
are 3x the radius apart from each other. Keep the vias small but
not so small that their inside diameter prevents solder wicking
through during reflow.
Connect all vias to the ground plane. It is important the vias have
a low thermal resistance for efficient heat transfer. It is
important to have a complete connection of the plated-through
hole to each plane.
Component Selection Guideline
MOSFET Considerations
The logic level MOSFETs are chosen for optimum efficiency given
the potentially wide input voltage range and output power
requirements. Two N-Channel MOSFETs are used in each of the
synchronous-rectified buck converters for the 3 PWM outputs.
These MOSFETs should be selected based upon rDS(ON), gate
supply requirements, and thermal management considerations.
The power dissipation includes two loss components; conduction
loss and switching loss. These losses are distributed between the
upper and lower MOSFETs according to duty cycle (see the
following equations). The conduction losses are the main
component of power dissipation for the lower MOSFETs. Only the
upper MOSFET has significant switching losses, since the lower
device turns on and off into near zero voltage. The equations
assume linear voltage-current transitions and do not model
power loss due to the reverse-recovery of the lower MOSFET’s
body diode.
2
( I O ) ( r DS ( ON ) ) ( V OUT ) ( I O ) ( V IN ) ( t SW ) ( F SW )
P UPPER = ---------------------------------------------------------- + -------------------------------------------------------V IN
2
(EQ. 12)
2
( I O ) ( r DS ( ON ) ) ( V IN – V OUT )
P LOWER = -----------------------------------------------------------------------V IN
(EQ. 13)
A large gate-charge increases the switching time, tSW, which
increases the upper MOSFETs’ switching losses. Ensure that both
MOSFETs are within their maximum junction temperature at high
ambient temperature by calculating the temperature rise
according to package thermal-resistance specifications.
20
The output capacitors for each output have unique requirements.
In general, the output capacitors should be selected to meet the
dynamic regulation requirements including ripple voltage and
load transients. Selection of output capacitors is also dependent
on the output inductor, so some inductor analysis is required to
select the output capacitors.
One of the parameters limiting the converter’s response to a load
transient is the time required for the inductor current to slew to
its new level. The ISL9443 will provide either 0% or maximum
duty cycle in response to a load transient.
The response time is the time interval required to slew the
inductor current from an initial current value to the load current
level. During this interval the difference between the inductor
current and the transient current level must be supplied by the
output capacitor(s). Minimizing the response time can minimize
the output capacitance required. Also, if the load transient rise
time is slower than the inductor response time, as in a hard drive
or CD drive, it reduces the requirement on the output capacitor.
The maximum capacitor value required to provide the full, rising
step, transient load current during the response time of the
inductor is:
2
( L O ) ( I TRAN )
C OUT = ----------------------------------------------------2 ( V IN – V O ) ( DV OUT )
(EQ. 15)
Where COUT is the output capacitor(s) required, LO is the output
inductor, ITRAN is the transient load current step, VIN is the input
voltage, VO is output voltage, and DVOUT is the drop in output
voltage allowed during the load transient.
High frequency capacitors initially supply the transient current
and slow the load rate-of-change seen by the bulk capacitors. The
bulk filter capacitor values are generally determined by the ESR
(Equivalent Series Resistance) and voltage rating requirements
as well as actual capacitance requirements.
The output voltage ripple is due to the inductor ripple current and
the ESR of the output capacitors as defined by:
V RIPPLE = ΔI L ( ESR )
(EQ. 16)
Where IL is calculated in the “Output Inductor Selection” on
page 20. High frequency decoupling capacitors should be placed
as close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load circuitry
for specific decoupling requirements.
FN7663.1
February 24, 2012
ISL9443
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. In most
cases, multiple small-case electrolytic capacitors perform better
than a single large-case capacitor.
1
C OUT = ---------------------------------2π ( ESR ) ( f Z )
(EQ. 17)
1. They must have sufficient bulk capacitance to sustain the
output voltage during a load transient while the output
inductor current is slewing to the value of the load transient.
2. The ESR must be sufficiently low to meet the desired output
voltage ripple due to the output inductor current.
3. The ESR zero should be placed, in a rather large range, to
provide additional phase margin.
The recommended output capacitor value for the ISL9443 is
between 100µF to 680µF, to meet stability criteria with external
compensation. Use of aluminum electrolytic (POSCAP) or
tantalum type capacitors is recommended. Use of low ESR
ceramic capacitors is possible but would take more rigorous loop
analysis to ensure stability.
Input Capacitor Selection
The important parameters for the bulk input capacitor(s) are the
voltage rating and the RMS current rating. For reliable operation,
select bulk input capacitors with voltage and current ratings
above the maximum input voltage and largest RMS current
required by the circuit. The capacitor voltage rating should be at
least 1.25 times greater than the maximum input voltage and
1.5 times is a conservative guideline. The AC RMS Input current
varies with the load. The total RMS current supplied by the input
capacitance is:
2
4.0
IN PHASE
3.5
3.0
2.5
OUT-OF-PHASE
2.0
1.5
5V
3.3V
1.0
0.5
In conclusion, the output capacitors must meet three criteria:
I RMS =
4.5
INPUT RMS CURRENT
The stability requirement on the selection of the output capacitor
is that the ‘ESR zero’ (f Z) be between 2kHz and 60kHz. This range
is set by an internal, single compensation zero at 8.8kHz. The
ESR zero can be a factor of five on either side of the internal zero
and still contribute to increased phase margin of the control loop.
Therefore:
5.0
2
I RMS1 + I RMS2
(EQ. 18)
0
0
1
2
3
3.3V AND 5V LOAD CURRENT
4
5
FIGURE 24. INPUT RMS CURRENT vs LOAD
Depending on the specifics of the input power and its
impedance, most (or all) of this current is supplied by the input
capacitor(s). Figure 24 shows the advantage of having the PWM
converters operating out-of-phase. If the converters were
operating in-phase, the combined RMS current would be the
algebraic sum, which is a much larger value as shown. The
combined out-of-phase current is the square root of the sum of
the square of the individual reflected currents and is significantly
less than the combined in-phase current.
Use a mix of input bypass capacitors to control the voltage ripple
across the MOSFETs. Use ceramic capacitors for the high
frequency decoupling and bulk capacitors to supply the RMS
current. Small ceramic capacitors can be placed very close to the
upper MOSFET to suppress the voltage induced in the parasitic
circuit impedances.
For board designs that allow through-hole components, the
Sanyo OS-CON™ series offers low ESR and good temperature
performance. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These capacitors
must be capable of handling the surge-current at power-up. The
TPS series available from AVX is surge current tested.
Where DC is duty cycle of the respective PWM.
I RMSx =
2
DC – DC ⋅ I O
(EQ. 19)
21
FN7663.1
February 24, 2012
ISL9443
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make
sure you have the latest revision.
DATE
February 13, 2012
June 20, 2011
REVISION
CHANGE
FN7663.1 Changed Input Voltage Range from “4.5V to 26V” to “4.5V to 28V” throughout datasheet.
Page 3 - Pin Description table, Pin 17/MODE/SYNC: Update 2nd sentence from “Tie this pin to ground or VCC_5V for
light load operation mode selection.” to “Tie this pin to ground or VCC_5V for DEM or CCM operation mode selection.”
FN7663.0 Initial Release
Products
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address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks.
Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a
complete list of Intersil product families.
*For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page
on intersil.com: ISL9443
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time
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22
FN7663.1
February 24, 2012
ISL9443
Package Outline Drawing
L32.5x5B
32 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 3, 5/10
4X 3.5
5.00
28X 0.50
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
32
25
1
5.00
24
3 .30 ± 0 . 15
17
(4X)
8
0.15
9
16
TOP VIEW
0.10 M C A B
+ 0.07
32X 0.40 ± 0.10
4 32X 0.23 - 0.05
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 90 ± 0.1
C
BASE PLANE
SEATING PLANE
0.08 C
( 4. 80 TYP )
(
( 28X 0 . 5 )
SIDE VIEW
3. 30 )
(32X 0 . 23 )
C
0 . 2 REF
5
( 32X 0 . 60)
0 . 00 MIN.
0 . 05 MAX.
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
23
FN7663.1
February 24, 2012