DATASHEET

ISL6341, ISL6341A, ISL6341B, ISL6341C
®
Data Sheet
December 2, 2008
5V or 12V Single Synchronous Buck
Pulse-Width Modulation (PWM) Controller
The ISL6341, ISL6341A, ISL6341B, ISL6341C makes
simple work out of implementing a complete control and
protection scheme for a DC/DC stepdown converter driving
N-Channel MOSFETs in a synchronous buck topology. Since
it can work with either 5V or 12V supplies, this one family of
IC’s can be used in a wide variety of applications within a
system. It integrates the control, gate drivers, output
adjustment, monitoring and protection functions into a single
10 Ld Thin DFN package.
The ISL6341, ISL6341A, ISL6341B, ISL6341C (hereafter
referred to as “ISL6341x”, except as needed) provides single
feedback loop, voltage-mode control with fast transient
response. The output voltage can be precisely regulated to as
low as 0.8V, with a maximum tolerance of ±0.8%
over-temperature and line voltage variations. A fixed frequency
oscillator and wide duty cycle range reduces design complexity,
while balancing typical application cost and efficiency. The
frequency, duty cycle and OCP response are the only
differences among the ISL6341x versions. See Table 1.
Protection from overcurrent conditions is provided by
monitoring the rDS(ON) of the lower MOSFET to inhibit PWM
operation appropriately (see “Overcurrent Protection (OCP)”
on page 8 for details). This approach simplifies the
implementation and improves efficiency by eliminating the
need for a current sense resistor. The output voltage is also
monitored for undervoltage and overvoltage protection, in
addition to monitoring for a PGOOD output.
FN6538.2
Features
• Operates from +4.5V to 14.4V Supply Voltage (for Bias)
- 1.5V to 12V VIN Input Range (Up to 20V is Possible with
Restrictions; see “Input Voltage Considerations” on
page 12)
- 0.8V to ~VIN Output Range (Duty Cycle Limited)
- Integrated Gate Drivers; LGATE Uses VCC (5V to 12V);
UGATE Uses External Boot Diode to (5V to 12V)
- 0.8V Internal Reference; ±0.8% Tolerance
• Simple Single-Loop Control Design
- Traditional Dual Edge Modulator
- Voltage-Mode PWM Control
- Drives N-Channel MOSFETs
• Fast Transient Response
- High-Bandwidth Error Amplifier
- 0% to 85% Max Duty Cycle for ISL6341, ISL6341C
- 0% to 75% Max Duty Cycle for ISL6341A, ISL6341B
• Lossless, Programmable Overcurrent Protection
- Uses Lower MOSFET’s rDS(ON)
- Latch off mode (ISL6341, ISL6341B)
- Infinite Retry (Hiccup) Mode (ISL6341A)
- Infinite Retry (Hiccup) Mode; no UVP (ISL6341C)
• Output Voltage Monitoring
- Undervoltage and Overvoltage Shutdown
- PGOOD Output
ISL6341ACRZ*
41AC
0 to +70
10 Ld 3x3 TDFN L10.3x3B
• Small Converter Size in 10 Ld 3x3 Thin DFN
- 300kHz Fixed Oscillator (ISL6341, ISL6341C)
- 600kHz Fixed Oscillator (ISL6341A, ISL6341B)
- Fixed Internal Soft-Start, Capable into a Pre-biased
Load
- Enable/Shutdown Function on COMP/EN Pin
ISL6341BCRZ*
41BC
0 to +70
10 Ld 3x3 TDFN L10.3x3B
• Pb-Free (RoHS Compliant)
ISL6341CCRZ*
41CC
0 to +70
10 Ld 3x3 TDFN L10.3x3B
ISL6341CRZ*
341C
0 to +70
10 Ld 3x3 TDFN L10.3x3B
ISL6341AIRZ*
41AI
-40 to +85
10 Ld 3x3 TDFN L10.3x3B
ISL6341BIRZ*
41BI
-40 to +85
10 Ld 3x3 TDFN L10.3x3B
Ordering Information
PART NUMBER
PART
TEMP.
(Note)
MARKING RANGE (°C)
PACKAGE
(Pb-free)
PKG.
DWG. #
ISL6341CIRZ*
41CI
-40 to +85
10 Ld 3x3 TDFN L10.3x3B
ISL6341IRZ*
6341
-40 to +85
10 Ld 3x3 TDFN L10.3x3B
ISL6341EVAL1Z Evaluation Board
*Add “-T” suffix for tape and reel. Please refer to TB347 for details on reel
specifications.
NOTE: These Intersil Pb-free plastic packaged products employ special
Pb-free material sets, molding compounds/die attach materials, and
100% matte tin plate plus anneal (e3 termination finish, which is RoHS
compliant and compatible with both SnPb and Pb-free soldering
operations). Intersil Pb-free products are MSL classified at Pb-free peak
reflow temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
1
Applications
• Power Supplies for Microprocessors or Peripherals
- PCs, Servers, Memory Supplies
- DSP and Core Communications Processor Supplies
• Subsystem Power Supplies
- PCI, AGP; Graphics Cards; Digital TV
- SSTL-2 and DDR/DDR2/DDR3 SDRAM Bus
Termination Supply
• Cable Modems, Set-Top Boxes, and DSL Modems
• Industrial Power Supplies; General Purpose Supplies
• 5V or 12V-Input DC/DC Regulators
• Low-Voltage Distributed Power Supplies; Point of Load
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2007, 2008. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6341, ISL6341A, ISL6341B, ISL6341C
Pinout
ISL6341, ISL6341A, ISL6341B, ISL6341C
(10 LD 3x3 TDFN)
TOP VIEW
BOOT
1
10 PGOOD
PHASE
2
9 VOS
UGATE
3
8 FB
LGATE/OCSET
4
7 COMP/EN
GND
5
6 VCC
Block Diagram
VCC
+
-
SAMPLE
AND
HOLD
POR AND
SOFT-START
OC
COMPARATOR
INTERNAL
REGULATOR
BOOT
UGATE
5V INT.
10µA
TO
LGATE/OCSET
ERROR
AMP
0.8V
+
-
20kΩ
PWM
COMPARATOR
+
-
FB
INHIBIT
GATE
CONTROL
PWM LOGIC
VCC
EN
+
+25% -
5V INT.
COMP/EN
PHASE
20µA
0.7V
+
-25% +
-
EN
LGATE/OCSET
OV1
GND
UV1
OSCILLATOR
OC
300kHz OR 600kHz
+
+10% -
OV2
PGOOD
VOS
-10% +
-
2
UV2
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
Typical Application
VCC
5V TO 12V
VIN
1.5V TO 12V
VGD
5V TO 12V
CDCPL
CHF
CBULK
VCC
BOOT
PGOOD 10
1
CBOOT
TYPE II
COMPENSATION
PHASE
2
SHOWN
COMP/EN
ISL6341x
7
UGATE
3
RF
CI FB 8
LGATE/OCSET
4
9
5
CF
VOS
GND
R
6
LOUT
+VO
COUT
OCSET
ROFFSET
RS
RVOS1
RVOS2
3
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
Absolute Maximum Ratings
Thermal Information
Supply Voltage, (VCC) . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 15V
BOOT Voltage (VBOOT-GND). . . . . . . . . . . . . . . . GND - 0.3V to 36V
BOOT to PHASE Voltage (VBOOT - VPHASE) . . . GND - 0.3V to 15V
-0.3V to 16V (<10ns, 10µJ)
UGATE Voltage (VUGATE) . . . . . . . VPHASE - 0.3V to VBOOT + 0.3V
VPHASE - 3.5V (<100ns Pulse Width, 2µJ) to VBOOT +0.3V
LGATE/OCSET Voltage (VLGATE) . . . . . . GND - 0.3V to VCC + 0.3V
GND - 5V (<100ns Pulse Width, 2µJ) to VCC + 0.3V
PHASE Voltage (VPHASE) . . . . . . . . . .GND - 0.3V to VBOOT + 0.3V
GND - 8V (<400ns, 20µJ) to 30V (<200ns, VBOOT-GND <36V)
FB, VOS, COMP/EN Voltage . . . . . . . . . . . . . . . . . GND - 0.3V to 6V
PGOOD Voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 7V
Thermal Resistance (Typical)
θJA (°C/W)
θJC (°C/W)
10 Ld TDFN Package (Notes 1, 2). . . .
44
3.5
Maximum Power Dissipation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.0W
Maximum Junction Temperature (Plastic Package) . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . -65°C to +150°C
Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Operating Conditions
Supply Voltage Range, VCC . . . . . . . . . . . . . . . . . . . +4.5V to 14.4V
Ambient Temperature Range
ISL6341xCRZ (Commercial) . . . . . . . . . . . . . . . . . . 0°C to +70°C
ISL6341xIRZ (Industrial) . . . . . . . . . . . . . . . . . . . .-40°C to +85°C
Junction Temperature Range. . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air, with “direct attach” features. See
Tech Brief TB379 for details.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Test Conditions: VCC = 12V, TJ = 0°C to +85°C, Unless Otherwise Noted. Parameters with MIN and/or MAX
limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
VCC SUPPLY CURRENT
Input Bias Supply Current
IVCC_dis
VCC = 12V; disabled
Input Bias Supply Current
IVCC_en
VCC = 12V; enabled (but not switching)
7.0
6.4
8.9
mA
10.4
mA
POWER-ON RESET
Rising VCC POR Threshold
VPOR
VCC POR Threshold Hysteresis
3.8
4.2
4.47
V
0.15
0.48
0.85
V
270
300
330
kHz
OSCILLATOR
Switching Frequency
fOSC
ISL6341C, ISL6341CC
ISL6341I, ISL6341CI
240
300
330
kHz
ISL6341AC, ISL6341BC
540
600
660
kHz
ISL6341AI, ISL6341BI
510
600
660
ΔVOSC
Ramp Amplitude (Note 3)
kHz
1.5
VP-P
REFERENCE
Reference Voltage Tolerance
VREF
ISL6341C, ISL6341AC, ISL6341BC,
ISL6341CC
0.7936
0.8000
0.8064
V
ISL6341I, ISL6341AI, ISL6341BI, ISL6341CI
0.7920
0.8000
0.8080
V
ERROR AMPLIFIER
DC Gain (Note 3)
GAIN
96
dB
Gain-Bandwidth Product (Note 3)
GBWP
20
MHz
SR
8
V/µs
Slew Rate (Note 3)
GATE DRIVERS
Upper Gate Source Impedance
RUG-SRCh VCC = 12V; I = 50mA
2.1
Ω
Upper Gate Sink Impedance
RUG-SNKh VCC = 12V; I = 50mA
1.6
Ω
Lower Gate Source Impedance
RLG-SRCh VCC = 12V; I = 50mA
1.4
Ω
Lower Gate Sink Impedance
RLG-SNKh VCC = 12V; I = 50mA
1.0
Ω
Upper Gate Source Impedance
RUG-SRCl VCC = 5V; I = 50mA
2.4
Ω
4
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
Electrical Specifications
Test Conditions: VCC = 12V, TJ = 0°C to +85°C, Unless Otherwise Noted. Parameters with MIN and/or MAX
limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
Upper Gate Sink Impedance
RUG-SNKl VCC = 5V; I = 50mA
1.7
Ω
Lower Gate Source Impedance
RLG-SRCl
VCC = 5V; I = 50mA
1.5
Ω
Lower Gate Sink Impedance
RLG-SNKl
VCC = 5V; I = 50mA
1.1
Ω
IOCSET
LGATE/OCSET = 0V
PROTECTION/DISABLE
OCSET Current Source
9
10
11
µA
0.683
0.700
0.717
V
0.868
0.880
0.888
V
0.708
0.720
VOS Rising Threshold (OV; +25%)
0.980
1.000
1.020
V
VOS Falling Threshold (UV; -25%)
(Note 5)
0.580
0.600
0.620
V
Enable Threshold (COMP/EN pin)
VENABLE
VOS Rising Trip (PGOOD OV; +10%)
VOS Rising Trip (PGOOD OV) Hysteresis
16
VOS Falling Trip (PGOOD UV; -10%)
VOS Falling Trip (PGOOD UV) Hysteresis
mV
0.732
16
VOS Threshold (OV; 50% of VOUT)
VOS Bias Current
VOS = 0.25V
PGOOD
IPGOOD = 4mA
V
mV
0.380
0.400
0.410
V
-1500
-250
-100
nA
0.10
0.18
0.30
V
NOTES:
3. Limits should be considered typical and are not production tested.
4. Limits established by characterization and are not production tested.
5. The UVP is disabled on the ISL6341C; no trip point is measured.
Functional Pin Description
VCC (Pin 6)
This pin provides the bias supply for the ISL6341x, as well
as the lower MOSFET’s gate. An internal regulator will
supply bias as VCC rises above 5V, but the LGATE/OCSET
will still be sourced by VCC. Connect a well-decoupled 5V to
12V supply to this pin.
FB (Pin 8)
This pin is the inverting input of the internal error amplifier. Use
FB, in combination with the COMP/EN pin, to compensate the
voltage-control feedback loop of the converter. A resistor divider
from VOUT to FB to GND is used to set the regulation voltage.
VOS (Pin 9)
This input pin monitors the regulator output for OV and UV
protection, and PGOOD (OV and UV). Connect a resistor
divider from VOUT to VOS to GND, with the same ratio as
the FB resistor divider. It is not recommended to share one
divider for both FB and VOS; the response to a fault may not
be as quick or robust. There is a small pull-up bias current
on the pin; if the VOS pin is not connected, the OV protection
would be tripped to protect the load.
GND (Pin 5)
This pin represents the signal and power ground for the IC.
This pin is the high current connection, and should be tied to
the ground island/plane through the lowest impedance
5
connection available. The metal pad under the package
should also be connected to the GND plane for thermal
conductivity, but does not conduct any current.
PHASE (Pin 2)
Connect this pin to the source of the upper MOSFET, and
the drain of the lower MOSFET. It is used as the sink for the
UGATE driver, and to monitor the voltage drop across the
lower MOSFET for overcurrent protection. This pin is also
monitored by the adaptive shoot-through protection circuitry
to determine when the upper MOSFET has turned off.
UGATE (Pin 3)
Connect this pin to the gate of upper MOSFET; it provides
the PWM-controlled gate drive. It is also monitored by the
adaptive shoot-through protection circuitry to determine
when the upper MOSFET has turned off.
BOOT (Pin 1)
This pin provides ground referenced bias voltage to the upper
MOSFET driver. A bootstrap circuit is used to create a voltage
suitable to drive an N-Channel MOSFET (equal to VGD minus
the BOOT diode voltage drop), with respect to PHASE.
COMP/EN (Pin 7)
This is a multiplexed pin. During soft-start and normal converter
operation, this pin represents the output of the error amplifier.
Use COMP/EN, in combination with the FB pin, to compensate
the voltage-control feedback loop of the converter.
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
Pulling COMP/EN low (VENABLE = 0.7V nominal) will
disable the controller, which causes the oscillator to stop, the
LGATE and UGATE outputs to be held low, and the soft-start
circuitry to re-arm. The external pull-down device will initially
need to overcome up to 5mA of COMP/EN output current.
However, once the IC is disabled, the COMP output will also
be disabled, so only a 20µA current source will continue to
draw current.
When the pull-down device is released, the COMP/EN pin will
start to rise, at a rate determined by the 20µA charging up the
capacitance on the COMP/EN pin. When the COMP/EN pin
rises above the VENABLE trip point, the ISL6341x will begin a
new initialization and soft-start cycle.
Functional Description
TABLE 1. SUMMARY OF FEATURE DIFFERENCES
fSW
(kHz)
MAX
DUTY
CYCLE
(%)
ISL6341
300
85
Latch off; toggle POR or COMP/EN
to restart
ISL6341A
600
75
“Hiccup” mode (infinite retries)
ISL6341B
600
75
Latch off; toggle POR or COMP/EN
to restart
ISL6341C
300
85
“Hiccup” mode (infinite retries); UVP
is disabled
PART
NUMBER
LGATE/OCSET (Pin 4)
Connect this pin to the gate of the lower MOSFET; it provides
the PWM-controlled gate drive (from VCC). This pin is also
monitored by the adaptive shoot-through protection circuitry to
determine when the lower MOSFET has turned off.
OCP
(OVERCURRENT PROTECTION)
Initialization (POR and OCP Sampling)
VCC (2V/DIV)
During a short period of time following Power-On Reset
(POR) or shut-down release, this pin is also used to
determine the overcurrent threshold of the converter.
Connect a resistor (ROCSET) from this pin to GND. See
“Overcurrent Protection (OCP)” on page 8 for equations. See
Table 1 for summary of overcurrent responses. Some of the
text describing the LGATE function may leave off the
OCSET part of the name when it is not relevant to the
discussion.
~4.3V POR
VOUT (1V/DIV)
COMP/EN (1V/DIV)
GND>
PGOOD (Pin 10)
This output is an open-drain pull-down device that reflects
the state of the PGOOD comparators. An external pull-up
resistor should be connected to a supply ≤6V. The output will
be held low through the soft-start ramp, and is allowed to go
high at the end of soft-start, if the VOS voltage is within its
window. The PGOOD window is tighter than the OV or UV
protection window, to give an early warning of a problem.
The PGOOD does respond directly to an OCP condition, but
may also go low if VOUT drops low enough before an OCP
trip.
Figure 1 shows a simplified timing diagram. The
Power-On-Reset (POR) function continually monitors the
bias voltage at the VCC pin. Once the rising POR threshold
is exceeded (VPOR = 4.3V nominal), the POR function
initiates the Overcurrent Protection (OCP) sample and hold
operation (while COMP/EN is ~1V). When the sampling is
complete, VOUT begins the soft-start ramp.
6
FIGURE 1. POR AND SOFT-START OPERATION
If the COMP/EN pin is held low during power-up, that will just
delay the initialization until it is released and the COMP/EN
voltage is above the VENABLE trip point.
Figure 2 shows a typical power-up sequence in more detail.
The initialization starts at t0, when either VCC rises above
VPOR, or the COMP/EN pin is released (after POR). The
COMP/EN will be pulled up by an internal 20µA current
source, but the timing will not begin until the COMP/EN
exceeds the VENABLE trip point (at t1). The external
capacitance of the disabling device, as well as the
compensation capacitors, will determine how quickly the
20µA current source will charge the COMP/EN pin. With
typical values, it should add a small delay compared to the
soft-start times. The COMP/EN will continue to ramp to ~1V.
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
.
LGATE
STARTS
SWITCHING
VOUT OVERCHARGED
COMP/EN (0.25V/DIV)
0.7V
VOUT
(0.25V/DIV)
VOUT PRE-BIASED
PGOOD (2V/DIV)
LGATE/OCSET
0.25V/DIV
GND>
GND>
t0
t1
t2
4.0ms
t3
0.8ms
t4
4.0ms
VOUT NORMAL
t0
t1
t2
FIGURE 3. SOFT-START WITH PRE-BIAS
FIGURE 2. LGATE/OCSET AND SOFT-START OPERATION
From t1, there is a nominal 4ms delay, which allows the VCC
pin to rise. At the same time, the LGATE/OCSET pin is
initialized by disabling the LGATE driver and drawing IOCSET
(nominal 10µA) through ROCSET. This sets up a voltage that
will represent the OCSET trip point for the OCP sample and
hold operation. The sample and hold uses a digital counter and
DAC (to save the voltage so the stored value does not degrade)
for as long as the VCC is above VPOR. See “Overcurrent
Protection (OCP)” on page 8 for more details on the equations
and variables. Upon the completion of sample and hold at t2,
the soft-start operation is initiated (around 0.8ms delay to t3),
and then around 4ms for the output voltage to ramp up (0% to
100%) between t3 and t4. The PGOOD output is allowed to go
high at t4 if VOS (and thus VOUT) is within the PGOOD
window.
Soft-Start and Pre-Biased Outputs
Functionally, the soft-start internally ramps the reference on the
non-inverting terminal of the error amp from zero to 0.8V in a
nominal 4ms. The output voltage will thus follow the ramp, from
zero to final value, in the same 4ms. The ramp is created
digitally, so there will be small discrete steps. There is no simple
way to change this ramp rate externally, as it is fixed by the
300kHz (or 600kHz) switching frequency (and the ramp and
delay time is the same for both frequencies).
After an initialization period (t2 to t3), the error amplifier
(COMP/EN pin) is enabled, and begins to regulate the
converter’s output voltage during soft-start. The oscillator’s
triangular waveform is compared to the ramping error amplifier
voltage. This generates PHASE pulses of increasing width that
charge the output capacitors. When the internally generated
soft-start voltage exceeds the reference voltage (0.8V), the
soft-start is complete, and the output should be in regulation at
the expected voltage. This method provides a rapid and
controlled output voltage rise; there is no large in-rush current
charging the output capacitors. The entire start-up sequence
from POR typically takes 9ms; 5ms for the delay and OCP
sample, and 4ms for the soft-start ramp.
7
Figure 3 shows the normal VOUT curve in blue; initialization
begins at t0, and the output ramps between t1 and t2. If the
output is pre-biased to a voltage less than the expected
value (as shown by the magenta curve), the ISL6341x will
detect that condition. Neither MOSFET will turn-on until the
soft-start ramp voltage exceeds the output; VOUT starts
seamlessly ramping from there.
There is a restriction for the pre-bias case; if the pre-biased
VOUT is greater than VGD, then the boot cap may get
discharged, and will not be able to restart. For example, if
VIN = 12V, VOUT = 8V and prebiased to 6V, and VGD is only
5V, then the voltage left on the boot cap (to UGATE) will not
be able to turn on the upper FET. The simple solution here is
to use the 12V for VGD. The guideline is to make VGD diode - Vth upper FET > VOUT to prevent this condition.
If the output is pre-biased to a voltage above the expected
value (as in the red curve), neither MOSFET will turn-on until
the end of the soft-start, at which time it will pull the output
voltage quickly down to the final value. Any resistive load
connected to the output will help pull-down the voltage (at
the RC rate of the R of the load and the C of the output
capacitance).
One exception to the overcharged case is if the pre-bias is
high enough to trip OV protection (>1V on VOS); then
LGATE will pulse to try to pull VOUT lower. The IC will remain
latched in this mode until VCC power is toggled.
If the VIN to the upper MOSFET drain (or the VGD voltage to
the boot diode) is from a different supply that comes up after
VCC, the soft-start would start its cycle, but with no output
voltage ramp. Once the undervoltage protection is enabled
(at the end of the soft-start ramp), the output will latch off.
Therefore, for normal operation, VIN (and VGD) must be high
enough before or with VCC. If this is not possible, then the
alternative is add sequencing logic to the COMP/EN pin to
delay the soft-start until the VIN (and VGD) supply is ready
(see “Input Voltage Considerations” on page 12).
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
If the IC is disabled after soft-start (by pulling COMP/EN pin
low), and then enabled (by releasing the COMP/EN pin),
then the full initialization (including a new OCP sample) will
take place.
If the output is shorted to GND during soft-start, the OCP will
handle it, as described in the next section.
Overcurrent Protection (OCP)
The overcurrent function protects the converter from a shorted
output by using the lower MOSFET’s ON-resistance, rDS(ON),
to monitor the current. A resistor (ROCSET) programs the
overcurrent trip level (see “Typical Application” on page 3).
This method enhances the converter’s efficiency and reduces
cost by eliminating a current sensing resistor.
Following POR and release of COMP/EN, the ISL6341x
initiates the Overcurrent Protection sample and hold
operation. The LGATE driver is disabled to allow an internal
10µA current source to develop a voltage across ROCSET.
The ISL6341x samples this voltage (which is referenced to
the GND pin) at the LGATE/OCSET pin, and holds it in a
counter and DAC combination. This sampled voltage is held
internally as the Overcurrent Set Point, for as long as power
is applied, or until a new sample is taken after coming out of
a COMP/EN shut-down.
The actual monitoring of the lower MOSFET’s ON-resistance
starts 200ns (nominal) after the edge of the internal PWM logic
signal (that creates the rising external LGATE signal). This is
done to allow the gate transition noise and ringing on the
PHASE pin to settle out before monitoring. The monitoring
ends when the internal PWM edge (and thus LGATE) goes low.
The OCP can be detected anywhere within the above window.
To allow sufficient time to detect OCP, the regulator will limit
the maximum UGATE duty cycle to ~85% at 300kHz (~75%
at 600kHz); there will always be an LGATE pulse of at least
300ns. This minimum width will also act as a boot-refresh
function. If the boot capacitor loses any charge while UGATE
is high, it will be refreshed each cycle while LGATE is high.
The ISL6341x share most of the detection circuitry; the main
difference among them is what happens after detection.
ISL6341, ISL6341B
When overcurrent is detected (while LGATE is high), the logic
will disable UGATE, and leave LGATE high until the current
drops to 1/2 of its programmed OCP value. This may take
several clock cycles, and it keeps the current from building up
too high. Once the current is low enough, UGATE will go high
on the next PWM cycle, and OCP will be monitored when
LGATE goes high. If OCP trips a 2nd time, it will again wait
until the current drops. If it trips for the 3rd time, it will latch off
the output (LGATE and UGATE low). If there is no OCP trip on
one of the retries, then the trip-counter resets to zero, and
three new consecutive cycles are required to latch off.
8
IINDUCTOR (10A/DIV)
OC
1/2 OC
0A>
LGATE (12V/DIV)
GND>
UGATE (24V/DIV)
GND>
FIGURE 4. OCP TIMING (ISL6341, ISL6341B)
Figure 4 shows a typical waveform for the ISL6341,
ISL6341B, where the normal inductor current is around 10A,
and the OCP trip is 16A. This is just an illustration; the actual
shape of the waveforms depends on the component values,
as well as the characteristics of the load and the short. On the
third trip, the gate drivers stop switching, and the current goes
to zero. To recover from this latched off condition, the user
must toggle VCC (power-down and power-up) for a new POR,
or toggle COMP/EN pin to restart (either includes initialization
and soft-start).
As the output inductor current rises and falls, the output
voltage is also affected. Note that in extreme cases during
the three consecutive trips, the UV may actually trip before
the OCP. The IC provides protection in either case, but
perhaps not quite at the programmed current. An OCP trip
can be reset by toggling either POR or COMP/EN, but a UV
trip is only reset by toggling POR. See Table 2 for the
protection summary.
Starting up into a shorted load will be handled the same way;
but the waveforms may look different, since the output is not
yet at its final value. OCP is always enabled during soft-start
(UV is not); it will need the three consecutive trips to latch off.
ISL6341A, ISL6341C
Figure 5 shows the same conditions for the ISL6341A,
ISL6341C. For this version, when overcurrent is first
detected (while LGATE is high), the logic will shut off the
output (LGATE and UGATE both go low), and the current
goes to zero.
It will then go into a “hiccup” mode of infinite retries. After two
dummy soft-start time-outs, a real soft-start will begin. If the
short is still there, it will trip during the soft-start ramp, and
will start another retry cycle. Once the short is removed, the
next real soft-start will be successful, and normal operation
can continue.
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
.
IINDUCTOR (10A/DIV)
INTERNAL SOFT-START RAMP DELAYS
OC
VOUT
(0.5V/DIV)
0A>
LGATE (12V/DIV)
GND>
UGATE (24V/DIV)
GND>
4.8ms
GND>
FIGURE 5. OCP TIMING (ISL6341A, ISL6341C)
Figure 6 shows the ISL6341A, ISL6341C output response
during a retry of an output shorted to GND. At time t0, the
output has been turned off, due to sensing an overcurrent
condition. There are two internal soft-start delay cycles (t1
and t2) to allow the MOSFETs to cool down, to keep the
average power dissipation in retry at an acceptable level. At
time t2, the output starts a normal soft-start cycle, and the
output tries to ramp. If the short is still applied, and the
current reaches the OCSET trip point any time during
soft-start ramp period, the output will shut off and return to
time t0 for another delay cycle. The retry period is thus two
dummy soft-start cycles plus one variable one, which
depends on how long it takes to trip the sensor each time.
Figure 6 shows an example where the output gets about
half-way up before shutting down; therefore, the retry (or
hiccup) time will be around 12ms. The minimum should be
nominally 9.6ms and the maximum 14.4ms. If the short
condition is finally removed, the output should ramp up
normally on the next t2 cycle.
Starting up into a shorted load looks the same as a retry into
that same shorted load. In both cases, OCP is always
enabled during soft-start; once it trips, it will go into retry
(hiccup) mode. The retry cycle will always have two dummy
time-outs, plus whatever fraction of the real soft-start time
passes before the detection and shut-off; at that point, the
logic immediately starts a new two dummy cycle time-out.
Both OCP and UVP protect against shorts to GND, but the
responses (and recovery from) are different, as shown in
Table 2. For some combinations of output components and
shorting method, it may be difficult to predict which
protection will trip first (output voltage going too low, or
current going too high). The ISL6341C removes that
uncertainty by disabling the UVP, and relying only on the
OCP. Note that for the other 3 versions, if OCP trips first, it
locks out the UVP from also tripping, so that only the OCP
response (and recovery) are active.
9
t0
4.8ms
t1
0ms TO 4.8ms
t2
4.8ms
t0
FIGURE 6. OCP RETRY OPERATION (ISL6341A, ISL6341C)
OVERCURRENT EQUATIONS
For all the ISL6341x, versions, the overcurrent function will
trip at a peak inductor current (IPEAK) determined by
Equation 1:
I OCSET xR OCSET
I PEAK = ------------------------------------------------r DS ( ON )
(EQ. 1)
where IOCSET is the internal OCSET current source (10µA
typical). The OC trip point varies in a system mainly due to
the MOSFET’s rDS(ON) variations (over process, current and
temperature). To avoid overcurrent tripping in the normal
operating load range, find the ROCSET resistor from
Equation 1 with:
1. The maximum rDS(ON) at the highest junction
temperature.
2. The minimum IOCSET from the “Electrical Specification
Table” on page 5.
( ΔI )
3. Determine IPEAK for I PEAK > I OUT ( MAX ) + ---------- ,
2
where ΔI is the output inductor ripple current.
For an equation for the ripple current see “Output Inductor
Selection” on page 15.
The range of allowable voltages detected (IOCSET*ROCSET)
is 0mV to 550mV; but the practical range for typical
MOSFETs is smaller. If the voltage drop across ROCSET is
set too low (< ~20mV), that can cause almost continuous
OCP tripping. It would also be very sensitive to system noise
and in-rush current spikes, so it should be avoided. The
maximum setting is 550mV, but most of the recommended
MOSFETs for the ISL6341x are not expected to handle the
power of the maximum trip point.
There is no way to disable the OCP, but setting it above the
maximum value (>600mV) will come close; for most cases, it
should be high enough (compared to the normal expected
range) to appear disabled. No resistor at all could give the
clamped maximum value (unless the loading on the LGATE
prevents charging the node fully). But there is no low-voltage
clamp on LGATE, so it could rise to over 3V and turn-on for
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
4ms during the sampling; that could discharge a pre-biased
output. Therefore, to avoid that case, but still come close to
disabling OCP, a resistor (>60kΩ) is recommended.
Note that conditions during power-up may look different than
normal operation. For example, during power-up in a 12V
system, the IC starts operation just above 4V; if the supply
ramp is slow, the soft-start ramp might be over well before
12V is reached. So with lower gate drive voltages, the
rDS(ON) of the MOSFETs will be higher during power-up,
effectively lowering the OCP trip. In addition, the ripple
current will likely be different at lower input voltage.
Another factor is the digital nature of the soft-start ramp. On
each discrete voltage step, there is in effect a small load
transient and a current spike to charge the output capacitors.
The height of the current spike is not controlled; it is affected
by the step size of the output, the value of the output
capacitors, as well as the IC error amp compensation. So it
is possible to trip the overcurrent with in-rush current, in
addition to the normal load and ripple considerations.
OCP is always enabled during soft-start, so there is
protection starting up into a shorted load.
Undervoltage Protection
The output is protected against undervoltage conditions by
monitoring the VOS pin. An external resistor divider (similar
ratio to the one on the FB pin) makes the voltage equal the
0.8V internal reference under normal operation. If the output
goes too low (25% below 0.8V = 0.6V nominal on VOS), the
output will latch off, with UGATE and LGATE both forced low.
This requires toggling VCC (power-down and up) to restart
(toggling COMP/EN will NOT restart it). The UV protection is
not enabled until the end of the soft-start ramp (as shown in
Figure 2).
Figure 7 shows a case where VOUT (and thus VOS) is pulled
down to the 75% point; both gate drivers stop switching, and
the VOUT is pulled low by the disturbance, as well as the
load, at a rate determine by the conditions, and the output
components.
The ISL6341C version does not have UVP; it relies on the
OCP for shorted loads. The PGOOD UV comparator is
separate, and is still active.
VOUT (0.25V/DIV)
75%
GND>
LGATE (12V/DIV)
GND>
UGATE (24V/DIV)
GND>
FIGURE 7. UNDERVOLTAGE PROTECTION
Overvoltage Protection
The output is protected against overvoltage conditions by
monitoring the VOS pin, similar to undervoltage. If the output
goes too high (25% above 0.8V = 1.0V nominal on VOS), the
output will latch off. As shown in Figure 8, UGATE will be
forced low, but LGATE will be forced high (to try to pull-down
the output) until the output drops to 1/2 of the normal voltage
(50% of 0.8V = 0.4V nominal on VOS). The LGATE will then
shut off, but will keep turning back on whenever the output
goes too high again.
Overvoltage latch-off requires toggling VCC (power-down and
up) to restart (toggling COMP/EN will NOT restart it). The OV
protection is not enabled until the rising VCC POR trip point is
exceeded. The OV protection is active during soft-start at the
fixed 25% above the final expected voltage. The OVP is not
gated off by tripping OCP (but the UVP is gated off if OCP
trips first).
If the VOS pin is disconnected, a small bias current on-chip
will force an overvoltage condition.
VOUT (0.5V/DIV)
125%
50%
GND>
LGATE (12V/DIV)
GND>
UGATE (24V/DIV)
GND>
FIGURE 8. OVERVOLTAGE PROTECTION
10
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
PGOOD
The PGOOD function output monitors the output voltage
using the same VOS pin and resistor divider of the
undervoltage and overvoltage protection, but with separate
comparators for each. The rising OV trip point (10% above
0.8V = 0.88V nominal on VOS) and the falling UV trip point
(10% below 0.8V = 0.72V nominal on VOS) will trip sooner
than the protection, in order to give an early warning to a
possible problem. The response time of the comparators
should be less than 1µs; the separate VOS input is not slowed
down by the compensation on the FB pin. It is NOT
recommended to connect the VOS pin to the FB pin, in order
to share the resistor divider. If the VOS pin is accidentally
disconnected, a small bias current on-chip will force an
overvoltage condition.
Figure 9 shows how the PGOOD output responds to a ramp
that trips in each direction (without reaching either protection
trip point at ±25%); PGOOD is valid (high) as long as VOUT
(and thus VOS) is within the ±10% window.
but since the PGOOD UV and OV windows are tighter, the
PGOOD output should already be low by the time either
protection is tripped.
TABLE 2. PROTECTION SUMMARY
PROTECTION
ACTION TAKEN
ENABLED
AFTER
RESET
BY
OCP
ISL6341
ISL6341B
VOUT latches off;
LGATE and UGATE low.
OCP
ISL6341A
ISL6341C
POR or
Not
Infinite retries; wait ~10ms,
and try a new Soft-Start ramp. COMP/EN Applicable
ISL6341C has UVP disabled
UVP
(-25%)
VOUT latches off;
after SS
ramp
LGATE and UGATE low.
ISL6341C has UVP disabled
POR
OVP
(+25%)
POR
VOUT latches off;
UGATE low;
LGATE goes low and high to
keep VOUT within 50% and
125% of nominal.
VOS pin open will trigger OV.
POR
PGOOD
(UV; -10%)
PGOOD goes low if VOS is
10% too low.
after SS
ramp
POR or
COMP/EN
PGOOD
(OV; +10%)
PGOOD goes low if VOS is
10% too high.
after SS
ramp
POR or
COMP/EN
PGOOD
(OCP)
PGOOD goes low if OCP trips after SS
ramp
POR or
COMP/EN
or good
SS ramp
POR or
POR or
COMP/EN COMP/EN
110%
90%
VOUT (0.25V/DIV)
GND>
PGOOD (2V/DIV)
Switching Frequency
GND>
FIGURE 9. PGOOD UNDERVOLTAGE AND OVERVOLTAGE
The PGOOD output is an open-drain pull-down NMOS
device; it can deliver 4.0mA of sink current at 0.3V when
power is NOT GOOD. A pull-up resistor to an external supply
voltage sets the high level voltage when power is GOOD. The
supply should be ≤6.0V, and is usually the one that powers
the logic monitoring the PGOOD output. If PGOOD function is
not used, the PGOOD pin can be left floating.
The PGOOD pin will be held low once VCC is above the rising
POR trip point, and during soft-start (but if the PGOOD supply
is up before or with VCC, it may be pulled high initially until the
logic has enough voltage to turn on the output). Once the
soft-start ramp is done (VOUT, VOS and FB should each be at
100% of their final value), the PGOOD pin will be allowed to
go high, if the output voltage is within the expected window.
There is no additional delay after soft-start is done.
Note that the overcurrent protection does directly affect the
PGOOD output, before the output voltage monitoring would
sense when VOUT drops 10%. The overvoltage and
undervoltage protection circuits don’t directly effect PGOOD,
11
The switching frequency is a fixed 300kHz for the ISL6341,
ISL6341C and 600kHz for the ISL6341A, ISL6341B. It
cannot be adjusted externally, and the various soft-start
delays and ramps are fixed at the same times for either
frequency.
Output Voltage Selection
The output voltage can be programmed to any level between
the 0.8V internal reference, up to the VIN supply, with the
85% duty cycle restriction for the ISL6341, ISL6341C (75%
for the ISL6341A, ISL6341B). Additional duty cycle margin
due to the rDS(ON) drop across the upper FET at maximum
load needs to be factored in as well.
An external resistor divider is used to scale the output
voltage relative to the internal reference voltage, and feed it
back to the inverting input of the error amp. See the “Typical
Application” schematic on page 3 for more detail; RS is the
upper resistor; ROFFSET (shortened to RO below) is the
lower one. The recommended value for RS is 1kΩ to 5kΩ
(±1% for accuracy) and then ROFFSET is chosen according
to Equation 2. Since RS is part of the compensation circuit
(see “Feedback Compensation” on page 13), it is often
easier to change ROFFSET to change the output voltage;
that way the compensation calculations do not need to be
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
repeated. If VOUT = 0.8V, then ROFFSET can be left open.
Output voltages less than 0.8V are not available.
( RS + RO )
V OUT = 0.8V • --------------------------RO
(EQ. 2)
R S • 0.8V
R O = ---------------------------------V OUT – 0.8V
The VOS pin is expected to see the same ratio for its resistor
divider; RVOS1 should also be chosen in the 1kΩ to 5kΩ
(±1% for accuracy) range. To simplify the BOM, RVOS1
should match RS, and RVOS2 should match ROFFSET.
If margining (or similar programmability) is added externally
(using a switch to change the effective lower resistor value),
the same method may be needed on the VOS pin resistor
divider. If the new VOUT (FB) is shifted too much compared
to the VOS trip, then PGOOD or UV/OV will be more likely to
trip in one direction (and less likely in the other).
Input Voltage Considerations
The “Typical Application” diagram on page 3 shows a
standard configuration where VCC is 5V to 12V, which
includes the standard 5V (±10%) or 12V (±20%) power
supply ranges. The gate drivers use the VCC voltage for
LGATE, and VGD (also 5V to 12V) for BOOT/UGATE. There
is an internal 5V regulator for bias.
The VIN to the upper MOSFET can share the same supply
as VCC, but can also run off a separate supply or other
sources, such as outputs of other regulators. If VCC powers
up first, and the VIN or VGD are not present by the time the
initialization is done, then undervoltage will trip at the end of
soft-start (and will not recover without toggling VCC; toggling
COMP/EN will not restart it). Therefore, either the supplies
must be turned on in the proper order (together, or VCC last),
or the COMP/EN pin should be used to disable VOUT until all
supplies are ready.
Figure 10 shows a simple sequencer for this situation. If VCC
powers up first, Q1 will be off and R3 pulling to VCC will turn
Q2 on, keeping the ISL6341x in shut-down. When VIN turns
on, the resistor divider R1 and R2 determines when Q1 turns
on, which will turn off Q2, and release the shut-down.
VIN
If VIN powers up first, Q1 will be on, turning Q2 off; so the
ISL6341x will start-up as soon as VCC comes up. The
VENABLE trip point is 0.7V nominal, so a wide variety of
NFET’s or NPN’s or even some logic IC’s can be used as Q1
or Q2. But Q2 should pull down hard when on, and must be
low leakage when off (open-drain or open-collector) so as
not to interfere with the COMP output. The Vth (or Vbe) of Q2
should be reviewed over process and temperature variations
to insure that it will work properly under all conditions. Q2
should be placed near the COMP/EN pin.
The VIN range can be as low as ~1.5V (for VOUT as low as
the 0.8V reference). It can be as high as 20V (for VOUT just
below VIN, limited by the maximum duty cycle). There are
some restrictions for running high VIN voltage.
The first consideration for high VIN is the maximum BOOT
voltage of 36V. The VIN (as seen on PHASE) plus VGD (boot
voltage - minus the diode drop), plus any ringing (or other
transients) on the BOOT pin must be less than 36V. If VIN is
20V, that limits VGD plus ringing to 16V.
The second consideration is the maximum voltage ratings
for VCC and BOOT-PHASE (for VGD); both are set at 15V. If
VIN is above the maximum operating range for VCC of
14.4V, then both VCC and VGD need to be supplied
separately. They can be derived from VIN (using a linear
regulator or equivalent), or they can be independent. In
either case, they must satisfy the power supply sequencing
requirements noted earlier (either power-up in the proper
order, or use a sequencer to disable the output until they are
all ready).
The third consideration for high VIN is duty cycle. Very low
duty cycles (such as 20V in to 1.0V out, for 5% duty cycle)
require component selection compatible with that choice
(such as low rDS(ON) lower MOSFET, a good LC output
filter, and compensation values to match). At the other
extreme (for example, 20V in to 12V out), the upper
MOSFET needs to be lower rDS(ON). There is also the
maximum duty cycle restriction. In all cases, the input and
output capacitors and both MOSFETs must be rated for the
voltages present.
Application Guidelines
VCC
Layout Considerations
R1
R2
R3
TO COMP/EN
Q2
Q1
FIGURE 10. SEQUENCER CIRCUIT
12
As in any high frequency switching converter, layout is very
important. Switching current from one power device to another
can generate voltage transients across the impedances of the
interconnecting bond wires and circuit traces. These
interconnecting impedances should be minimized by using
wide, short printed circuit traces. The critical components
should be located as close together as possible, using ground
plane construction or single point grounding.
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
Feedback Compensation
.
UGATE
Q1
LO
PHASE
LGATE/OCSET
VOUT
CIN
Q2
This section highlights the design consideration for a
voltage-mode controller requiring external compensation. To
address a broad range of applications, a type-3 feedback
network is recommended, as shown in the top part of
Figure 13.
LOAD
ISL6341x
VIN
CO
RETURN
FIGURE 11. PRINTED CIRCUIT BOARD POWER AND
GROUND PLANES OR ISLANDS
Figure 11 shows the critical power components of the
converter. To minimize the voltage overshoot, the
interconnecting wires indicated by heavy lines should be part of
a ground or power plane in a printed circuit board. The
components shown should be located as close together as
possible. Please note that the capacitors CIN and CO may each
represent numerous physical capacitors. For best results,
locate the ISL6341x within 1 inch of the MOSFETs, Q1 and Q2 .
The circuit traces for the MOSFET gate and source
connections from the ISL6341x must be sized to handle up to
2A peak current.
ISL6341x
BOOT
+VIN
CBOOT
Q1 LO
(EQ. 3)
VOUT
PHASE
+VCC
LGATE/OCSET
VCC
1
F CE = -----------------------2π ⋅ C ⋅ E
C2
Q2
CO
LOAD
VOS
COMP/EN
FB
The modulator transfer function is the small-signal transfer
function of VOUT /VCOMP. This function is dominated by a DC
gain, given by dMAXVIN /VOSC , and shaped by the output
filter, with a double pole break frequency at FLC and a zero at
FCE . For the purpose of this analysis, L and D represent the
channel inductance and its DCR, while C and E represent the
total output capacitance and its equivalent series resistance.
1
F LC = --------------------------2π ⋅ L ⋅ C
+VGD
VOUT
Figure 13 also highlights the voltage-mode control loop for a
synchronous-rectified buck converter, applicable to the
ISL6341x circuit. The output voltage (VOUT) is regulated to
the reference voltage, VREF. The error amplifier output
(COMP pin voltage) is compared with the oscillator (OSC)
modified sawtooth wave to provide a pulse-width modulated
wave with an amplitude of VIN at the PHASE node. The
PWM wave is smoothed by the output filter (L and C). The
output filter capacitor bank’s equivalent series resistance is
represented by the series resistor E.
COMP
C3
R3
R2
C1
-
CVCC
GND
ROCSET
R1
FB
E/A
+
Ro
VREF
FIGURE 12. PRINTED CIRCUIT BOARD SMALL SIGNAL
LAYOUT GUIDELINES
Figure 12 shows the circuit traces that require additional
layout consideration. Use single point and ground plane
construction for the circuits shown. Provide local VCC
decoupling between VCC and GND pins. Locate the
capacitor, CBOOT as close as practical to the BOOT and
PHASE pins. Locate the resistor, ROSCET close to the
LGATE/OCSET pin because the internal current source is
only 10µA. Minimize any leakage current paths on the
COMP/EN pin. All components used for feedback
compensation and VOS resistor divider (inside the dotted
box) should be located as close to the IC as practical. Near
the load, pick a point VOUT that will be the regulation center;
run a single unloaded narrow trace from there to the
compensation components. The same trace can also be
used for VOS divider.
13
VOUT
OSCILLATOR
VIN
PWM
CIRCUIT
VOSC
UGATE
HALF-BRIDGE
DRIVE
L
D
PHASE
C
E
LGATE
ISL6341x
EXTERNAL CIRCUIT
FIGURE 13. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
4. Select a value for R1 (1kΩ to 5kΩ, typically). Calculate the
value for R2 for desired converter bandwidth (F0). If
setting the output voltage via an offset resistor connected
to the FB pin (Ro in Figure 13), the design procedure can
be followed as presented in Equation 4.
V OSC ⋅ R 1 ⋅ F 0
R 2 = --------------------------------------------d MAX ⋅ V IN ⋅ F LC
(EQ. 4)
5. Calculate C1 such that FZ1 is placed at a fraction of the FLC,
at 0.1 to 0.75 of FLC (to adjust, change the 0.5 factor to
desired number). The higher the quality factor of the output
filter and/or the higher the ratio FCE/FLC, the lower the FZ1
frequency (to maximize phase boost at FLC).
1
C 1 = ----------------------------------------------2π ⋅ R 2 ⋅ 0.5 ⋅ F LC
d MAX ⋅ V IN
1 + s(f) ⋅ E ⋅ C
G MOD ( f ) = ------------------------------ ⋅ ---------------------------------------------------------------------------------------2
V OSC
1 + s(f) ⋅ (E + D) ⋅ C + s (f) ⋅ L ⋅ C
R
G CL ( f ) = G MOD ( f ) ⋅ G FB ( f )
where, s ( f ) = 2π ⋅ f ⋅ j
(EQ. 8)
COMPENSATION BREAK FREQUENCY EQUATIONS
1
F P1 = -------------------------------------------C
C
R
1⋅ 2
2π ⋅ 2 ⋅ -------------------C
C
1+ 2
1
F Z1 = -----------------------------R
C
2π ⋅ 2 ⋅ 1
1
F Z2 = -----------------------------------------------R
R
C
2π ⋅ ( 1 + 3 ) ⋅ 3
FZ1 FZ2
(EQ. 9)
1
F P2 = -----------------------------R
C
2π ⋅ 3 ⋅ 3
FP1
FP2
MODULATOR GAIN
COMPENSATION GAIN
CLOSED LOOP GAIN
OPEN LOOP E/A GAIN
(EQ. 5)
6. Calculate C2 such that FP1 is placed at FCE.
⎛R ⎞
2
20 log ⎜ --------⎟
⎜R ⎟
⎝ 1⎠
(EQ. 6)
d MAX ⋅ V
IN
20 log --------------------------------V OSC
0
7. Calculate R3 such that FZ2 is placed at FLC. Calculate C3
such that FP2 is placed below fSW (typically, 0.5 to 1.0
times fSW). fSW represents the switching frequency.
Change the numerical factor to reflect desired placement
of this pole. Placement of FP2 lower in frequency helps
reduce the gain of the compensation network at high
frequency, in turn reducing the HF ripple component at
the COMP pin and minimizing resultant duty cycle jitter.
1
C 3 = ----------------------------------------------2π ⋅ R 3 ⋅ 0.7 ⋅ f SW
(EQ. 7)
It is recommended that a mathematical model be used to
plot the loop response. Check the loop gain against the error
amplifier’s open-loop gain. Verify phase margin results and
adjust as necessary. Equations 8 and 9 describe the
frequency response of the modulator (GMOD), feedback
compensation (GFB) and closed-loop response (GCL):
GFB
GCL
LOG
C1
C 2 = -------------------------------------------------------2π ⋅ R 2 ⋅ C 1 ⋅ F CE – 1
R1
R 3 = -------------------f SW
----------- – 1
F LC
C
1 + s(f) ⋅ 2 ⋅ 1
G FB ( f ) = --------------------------------------------------- ⋅
R
C
C
s(f) ⋅ 1 ⋅ ( 1 + 2)
R
R
C
1 + s(f) ⋅ ( 1 + 3) ⋅ 3
⋅ -----------------------------------------------------------------------------------------------------------------------⎛
⎛ C1 ⋅ C2 ⎞ ⎞
R
C
R
( 1 + s ( f ) ⋅ 3 ⋅ 3 ) ⋅ ⎜ 1 + s ( f ) ⋅ 2 ⋅ ⎜ --------------------⎟ ⎟
⎝
⎝ C1 + C2⎠ ⎠
GAIN
The compensation network consists of the error amplifier
(internal to the ISL6341x) and the external R1 to R3, C1 to C3
components. The goal of the compensation network is to
provide a closed loop transfer function with high 0dB crossing
frequency (F0; typically 0.1 to 0.3 of fSW) and adequate phase
margin (better than 45°). Phase margin is the difference
between the closed loop phase at F0dB and 180°. The
equations that follow relate the compensation network’s poles,
zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and
C3) in Figure 13. Use the following guidelines for locating the
poles and zeros of the compensation network:
GMOD
LOG
FLC
FCE
F0
FREQUENCY
FIGURE 14. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Figure 14 shows an asymptotic plot of the DC/DC converter’s
gain vs frequency. The actual Modulator Gain has a high gain
peak dependent on the quality factor (Q) of the output filter,
which is not shown. Using the previous guidelines should yield
a compensation gain similar to the curve plotted. The open loop
error amplifier gain bounds the compensation gain. Check the
compensation gain at FP2 against the capabilities of the error
amplifier. The closed loop gain, GCL, is constructed on the loglog graph of Figure 14 by adding the modulator gain, GMOD (in
dB), to the feedback compensation gain, GFB (in dB). This is
equivalent to multiplying the modulator transfer function and the
compensation transfer function and then plotting the resulting
gain.
A stable control loop has a gain crossing with close to a
-20dB/decade slope and a phase margin greater than 45°.
Include worst case component variations when determining
phase margin. The mathematical model presented makes a
number of approximations and is generally not accurate at
frequencies approaching or exceeding half the switching
14
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
frequency. When designing compensation networks, select
target crossover frequencies in the range of 10% to 30% of
the switching frequency, fSW.
This is just one method to calculate compensation components;
there are variations of the compensation break frequency
equations. The error amp is similar to that on other Intersil
regulators, so existing tools can be used here as well. Special
consideration is needed if the size of a ceramic output
capacitance in parallel with bulk capacitors gets too large; the
calculation needs to model them both separately (attempting to
combine two different capacitors types into one composite
component model may not work properly; a special tool may be
needed; contact your local Intersil person for assistance).
Component Selection Guidelines
Output Capacitor Selection
An output capacitor is required to filter the output and supply
the load transient current. The filtering requirements are a
function of the switching frequency and the ripple current.
The load transient requirements are a function of the slew
rate (di/dt) and the magnitude of the transient load current.
These requirements are generally met with a mix of
capacitors and careful layout.
Modern components and loads are capable of producing
transient load rates above 1A/ns. High frequency capacitors
initially supply the transient and slow the current load rate
seen by the bulk capacitors. The bulk filter capacitor values
are generally determined by the ESR (Effective Series
Resistance) and voltage rating requirements rather than
actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
15
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by Equation 10:
ΔI =
VIN - VOUT
Fsw x L
x
VOUT
ΔVOUT = ΔI x ESR
VIN
(EQ. 10)
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6341x will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval, the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. Equation 11
gives the approximate response time interval for application
and removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
(EQ. 11)
VOUT
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check Equation 11 at the minimum and
maximum output levels for the worst case response time.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time Q1 turns on. Place the
small ceramic capacitors physically close to the MOSFETs
and between the drain of Q1 and the source of Q2 .
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25x greater than the maximum input
voltage and a voltage rating of 1.5x is a conservative
guideline. The RMS current rating requirement for the input
capacitor of a buck regulator is approximately 1/2 the DC
load current.
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
MOSFET Selection/Considerations
The ISL6341x requires 2 N-Channel power MOSFETs. These
should be selected based upon rDS(ON) , gate supply
requirements, and thermal management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor. The switching
losses seen when sourcing current will be different from the
switching losses seen when sinking current. When sourcing
current, the upper MOSFET realizes most of the switching
losses. The lower switch realizes most of the switching losses
when the converter is sinking current (see Equation 12).
Equation 12 assumes linear voltage-current transitions and
does not adequately model power loss due to the reverserecovery of the upper and lower MOSFET’s body diode. The
gate-charge losses are dissipated by the ISL6341x and don't
heat the MOSFETs. However, large gate-charge increases the
switching interval, tSW which increases the MOSFET switching
losses. Ensure that both MOSFETs are within their maximum
junction temperature at high ambient temperature by
calculating the temperature rise according to package
thermal-resistance specifications. A separate heatsink may be
necessary depending upon MOSFET power, package type,
ambient temperature and air flow.
For 5V only operation, given the reduced available gate bias
voltage (5V), logic-level transistors should be used for both
N-MOSFETs. Look for rDS(ON) ratings at 4.5V. Caution
should be exercised with devices exhibiting very low
VGS(ON) characteristics. The shoot-through protection
present aboard the ISL6341x may be circumvented by these
MOSFETs if they have large parasitic impedences and/or
capacitances that would inhibit the gate of the MOSFET from
being discharged below its threshold level before the
complementary MOSFET is turned on. Also avoid MOSFETs
with excessive switching times; the circuitry is expecting
transitions to occur in under 50ns or so.
BOOTSTRAP Considerations
Figure 15 shows the upper gate drive (BOOT pin) supplied
by a bootstrap circuit from VGD. For convenience, VGD
usually shares the VIN or VCC supply; it can be any voltage
in the 5V to 12V range. The boot capacitor, CBOOT,
develops a floating supply voltage referenced to the PHASE
pin. The supply is refreshed to a voltage of VGD less the
boot diode drop (VD) each time the lower MOSFET, Q2 ,
turns on. Check that the voltage rating of the capacitor is
above the maximum VCC voltage in the system; a 16V rating
should be sufficient for a 12V system. A value of 0.1µF is
typical for many systems driving single MOSFETs.
If VCC is 12V, but VIN is lower (such as 5V), then another
option is to connect the BOOT pin to 12V, and remove the
BOOT cap (although, you may want to add a local capacitor
from BOOT to GND). This will make the UGATE VGS voltage
equal to (12V - 5V = 7V). That should be high enough to
drive most MOSFETs, and low enough to improve the
efficiency slightly. This also saves a boot diode and
capacitor.
+VGD
+VCC
VCC
ISL6341x
Losses while Sourcing Current
+VIN
- VD +
For a through-hole design, several electrolytic capacitors may
be needed. For surface mount designs, solid tantalum
capacitors can also be used, but caution must be exercised
with regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge current at
power-up. Some capacitor series available from reputable
manufacturers are surge current tested.
BOOT
CBOOT
1
P UPPER = Io × r DS ( ON ) × D + --- ⋅ Io × V IN × t SW × F S
2
PLOWER = Io2 x rDS(ON) x (1 - D)
2
UGATE
Q1
PHASE
VG-S ≈ VGD - VD
Losses while Sinking Current
(EQ. 12)
PUPPER = Io2 x rDS(ON) x D
2
1
P LOWER = Io × r DS ( ON ) × ( 1 – D ) + --- ⋅ Io × V IN × t SW × F S
2
Where: D is the duty cycle = VOUT / VIN ,
tSW is the combined switch ON and OFF time, and
fSW is the switching frequency.
When operating with a 12V power supply for VCC (or down
to a minimum supply voltage of 4.5V), a wide variety of
N-MOSFETs can be used. Check the absolute maximum
VGS rating for both MOSFETs; it needs to be above the
highest VCC voltage allowed in the system; that usually
means a 20V VGS rating (which typically correlates with a
30V VDS maximum rating). Low threshold transistors
(around 1V or below) are not recommended, as explained in
the following.
16
VCC
Q2
-
+
LGATE/OCSET
VG-S ≈ VCC
GND
FIGURE 15. UPPER GATE DRIVE BOOTSTRAP
FN6538.2
December 2, 2008
ISL6341, ISL6341A, ISL6341B, ISL6341C
Thin Dual Flat No-Lead Plastic Package (TDFN)
L10.3x3B
2X
10 LEAD THIN DUAL FLAT NO-LEAD PLASTIC PACKAGE
0.15 C A
A
D
MILLIMETERS
2X
0.15 C B
E
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.70
0.75
0.80
-
A1
-
-
0.05
-
0.20 REF
A3
6
INDEX
AREA
b
0.18
D2
B
2.23
A
C
SEATING
PLANE
D2
6
INDEX
AREA
7
0.08 C
8
2
7, 8
-
1.49
1.64
1.74
7, 8
0.50 BSC
-
k
0.20
-
-
L
0.30
0.40
0.50
8
N
10
2
Nd
5
3
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
NX k
3. Nd refers to the number of terminals on D.
4. All dimensions are in millimeters. Angles are in degrees.
E2
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
E2/2
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
NX L
N
N-1
NX b
e
(Nd-1)Xe
REF.
BOTTOM VIEW
5
0.10 M C A B
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land
Pattern Design efforts, see Intersil Technical Brief TB389.
9. COMPLIANT TO JEDEC MO-229-WEED-3 except for
dimensions E2 & D2.
CL
NX (b)
2.48
NOTES:
(DATUM A)
8
2.38
Rev. 0 2/06
D2/2
1
E2
e
A3
SIDE VIEW
(DATUM B)
0.10 C
5, 8
3.00 BSC
E
//
0.30
3.00 BSC
D
TOP VIEW
0.25
-
(A1)
9 L
5
e
SECTION "C-C"
C C
TERMINAL TIP
FOR ODD TERMINAL/SIDE
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
17
FN6538.2
December 2, 2008
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