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Data
Sheet
March 20, 2007
T
N
1-888-I
Multiphase PWM Controller with Linear
6-Bit DAC Capable of Precision rDS(ON) or
DCR Differential Current Sensing
The ISL6564A is a Multiphase PWM controller which controls
microprocessor core voltage regulation by driving up to
4 synchronous-rectified buck channels. It features a high
bandwidth control loop to provide optimal response to the load
transients. With switching frequency up to 1.5MHz per phase,
the ISL6564A based voltage regulator requires minimum
components and PCB area in DC/DC converter application.
The ISL6564A senses current by utilizing patented
techniques to measure the voltage across the on resistance,
rDS(ON), of the lower MOSFETs or DCR of the output
inductor during their conduction intervals. Current sensing
provides the needed signals for precision droop, channelcurrent balancing, and overcurrent protection.
A unity gain, differential amplifier is provided for remote
voltage sensing. Any potential difference between remote
and local grounds can be completely eliminated using the
remote-sense amplifier. Eliminating ground differences
improves regulation and protection accuracy. The thresholdsensitive enable input is available to accurately coordinate
the start up of the ISL6564A with any other voltage rail.
Dynamic-VID™ technology allows seamless on-the-fly VID
changes. The offset pin allows accurate voltage offset
settings that are independent of VID setting. The ISL6564A
uses a 5V bias and has a built-in shunt regulator to allow
12V bias using only a small external limiting resistor.
Ordering Information
PART NUMBER
PART
MARKING
ISL6564ACRZ* ISL6564 ACRZ
(Note)
ISL6564AIRZ*
(Note)
ISL6564 AIRZ
TEMP.
(°C)
PACKAGE
PKG.
DWG. #
0 to +70 40 Ld 6x6 QFN L40.6x6
(Pb-free)
ISL6564A
FN6285.1
Features
• Precision Multiphase Core Voltage Regulation
- Differential Remote Voltage Sensing
- 0.5% System Accuracy
- Adjustable Reference-Voltage Offset
• Precision rDS(ON) or DCR Current Sensing
- Accurate Load-Line Programming
- Accurate Channel-Current Balancing
- Differential Current Sense
- Low-Cost, Lossless Current Sensing
• Internal Shunt Regulator for 5V or 12V Biasing
• Microprocessor Voltage Identification Input
- Self Clocked Dynamic VID™ Control Technology
- 6-Bit VID Input
- 0.525V to 1.300V in 12.5mV Steps
• Threshold-Sensitive Enable Function for Power
Sequencing Control
• Overcurrent Protection
• Overvoltage Protection
- No Additional External Components Needed
- OVP Pin to Drive Crowbar Device
• 1, 2, 3, or 4 Phase Operation
• Up to 1.5MHz Per Phase Operation (>6MHz Ripple)
• QFN Package
- Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat
No Leads - Product Outline
- QFN Near Chip Scale Package Footprint; Improves
PCB Efficiency, Thinner in Profile
• Pb-Free Plus Anneal Available (RoHS Compliant)
-40 to +85 40 Ld 6x6 QFN L40.6x6
(Pb-free)
*Add “-T” suffix for tape and reel.
NOTE: Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and 100%
matte tin plate termination finish, which are RoHS compliant and
compatible with both SnPb and Pb-free soldering operations. Intersil
Pb-free products are MSL classified at Pb-free peak reflow
temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2006-2007. All Rights Reserved. Dynamic VID™ is a trademark of Intersil Americas Inc.
All other trademarks mentioned are the property of their respective owners.
ISL6564A
Pinout
2
GND
PGOOD
OVP
GND
FS
DRVEN
ENLL
EN
VCC
PWM4
ISL6564A
(40 LD QFN)
TOP VIEW
40
39
38
37
36
35
34
33
32
31
VID5
1
30 ISEN4+
VID4
2
29 ISEN4-
VID3
3
28 ISEN2-
VID2
4
27 ISEN2+
VID1
5
26 PWM2
GND
9
22 ISEN3-
DAC 10
21 ISEN3+
11
12
13
14
15
16
17
18
19
20
PWM3
IOUT
GND
23 ISEN1-
RGND
8
VSEN
OFS
VDIFF
24 ISEN1+
COMP
7
IDROOP
GND
FB
25 PWM1
GND
6
REF
VID0
FN6285.1
March 20, 2007
ISL6564A
ISL6564A Block Diagram
VDIFF
RGND
OVP
PGOOD
S
x1
OVP
LATCH
VSEN
R
ENLL
VCC
DRVEN
1.29V
POWER-ON
RESET (POR)
EN
Q
THREE-STATE
SOFT-START
AND
FAULT LOGIC
OVP
CLOCK AND
SAWTOOTH
GENERATOR

+200mV
OFS
FS
PWM1
PWM

OFFSET
PWM2
PWM

REF
PWM3
PWM
DAC

VID5
VID4
PWM4
PWM
VID3
DYNAMIC
VID
VID2
D/A
E/A
VID1
CHANNEL
CURRENT
BALANCE
VID0
CHANNEL
DETECT
COMP
ISEN1+
I_TRIP
FB
OC
IDROOP
ISEN1-

SAMPLE
&
HOLD
CHANNEL
ISEN2+
CURRENT
ISEN2-
SENSE
I_TOT
ISEN3+
ISEN3-
IOUT
ISEN4+
ISEN4-
GND
3
FN6285.1
March 20, 2007
ISL6564A
Typical Application for Voltage Regulation without Droop Using rDS(ON) Sensing
+12V
VIN
VCC
BOOT
UGATE
PVCC
ISL6612
DRIVER
PWM
PHASE
LGATE
GND
+5V
FB
COMP REF
IDROOP
VDIFF
VSEN
ENLL
PGOOD
OVP
BOOT
VCC
VIN
VCC
RGND
VIDPGOOD
+12V
DAC
UGATE
EN
ISL6564A
VID5
ISEN1+
ISEN1-
VID4
PWM1
PVCC
PHASE
ISL6612
PWM
DRIVER
LGATE
GND
PWM2
VID3
ISEN2+
VID2
ISEN2-
VID1
PWM3
VID0
ISEN3+
DRVEN
ISEN3OFS
FS
IOUT
+12V
VIN
PWM4
ISEN4+
GND ISEN4-
µP
LOAD
VCC
BOOT
UGATE
PVCC
PHASE
ISL6612
DRIVER
PWM
LGATE
GND
NTC
NETWORK
+12V
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
VCC
VIN
BOOT
UGATE
PVCC
ISL6612
PWM
PHASE
DRIVER
LGATE
GND
4
FN6285.1
March 20, 2007
ISL6564A
Typical Application for Voltage Regulation without Droop Using DCR Sensing
+12V
VIN
VCC
BOOT
UGATE
PVCC
ISL6612
DRIVER
PWM
PHASE
LGATE
GND
+5V
FB
COMP REF
IDROOP
VDIFF
VSEN
ENLL
PGOOD
OVP
VCC
BOOT
VIN
VCC
RGND
VIDPGOOD
+12V
DAC
UGATE
EN
ISL6564A
VID5
ISEN1+
ISEN1-
VID4
PWM1
PVCC
PHASE
ISL6612
PWM
DRIVER
LGATE
GND
PWM2
VID3
ISEN2+
VID2
ISEN2-
VID1
PWM3
VID0
ISEN3+
DRVEN
ISEN3OFS
FS
IOUT
+12V
VIN
PWM4
ISEN4+
GND ISEN4-
µP
LOAD
VCC
BOOT
UGATE
PVCC
ISL6612
PHASE
DRIVER
PWM
LGATE
NTC
NETWORK
GND
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
+12V
VIN
VCC
BOOT
UGATE
PVCC
PHASE
ISL6612
PWM
DRIVER
LGATE
GND
5
FN6285.1
March 20, 2007
ISL6564A
Typical Application for Load Line Regulation Using rDS(ON) Sensing and External NTC
+12V
VIN
VCC
BOOT
UGATE
PVCC
ISL6612
PHASE
DRIVER
PWM
LGATE
FB
IDROOP
COMP REF
VSEN
BOOT
VCC
UGATE
ENLL
PGOOD
PVCC
ISL6564A
OVP
VIN
VCC
RGND
VIDPGOOD
+12V
DAC
VDIFF
NTC
THERMISTOR
GND
+5V
VID4
ISEN1+
ISEN1-
VID3
PWM1
PHASE
ISL6612
DRIVER
PWM
LGATE
GND
PWM2
VID2
ISEN2+
VID1
ISEN2-
VID0
PWM3
VID12.5
ISEN3+
DRVEN
ISEN3OFS
PWM4
FS
ISEN4+
ISEN4GND
IOUT
+12V
VIN
µP
LOAD
VCC
BOOT
UGATE
PVCC
ISL6612
EN
PHASE
DRIVER
PWM
+12V
LGATE
GND
NTC
NETWORK
+12V
VCC
VIN
BOOT
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
UGATE
PVCC
ISL6612
PWM
PHASE
DRIVER
LGATE
GND
6
FN6285.1
March 20, 2007
ISL6564A
Typical Application for Load Line Regulation Using DCR Sensing and External NTC
+12V
VCC
BOOT
VIN
UGATE
PVCC
ISL6612
PHASE
DRIVER
PWM
LGATE
FB
IDROOP
COMP REF
VSEN
VCC
BOOT
UGATE
ENLL
PGOOD
OVP
VIN
VCC
RGND
VIDPGOOD
+12V
DAC
VDIFF
NTC
THERMISTOR
GND
+5V
PVCC
ISL6564A
VID4
ISEN1+
ISEN1-
VID3
PWM1
VID2
PWM2
VID1
ISEN2+
ISEN2-
VID0
PWM
ISEN3+
DRVEN
ISEN3OFS
PWM4
FS
ISEN4+
ISEN4GND
IOUT
DRIVER
LGATE
GND
+12V
PWM3
VID12.5
PHASE
ISL6612
VIN
µP
LOAD
VCC
BOOT
UGATE
PVCC
PHASE
ISL6612
EN
+12V
DRIVER
PWM
LGATE
GND
NTC
NETWORK
+12V
VIN
VCC
BOOT
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
UGATE
PVCC
ISL6612
PWM
PHASE
DRIVER
LGATE
GND
7
FN6285.1
March 20, 2007
ISL6564A
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5.5V
Input and Output Voltage (except OVP). . GND -0.3V to VCC + 0.3V
OVP Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15V
ESD (Human body model) . . . . . . . . . . . . . . . . . . . . . . . . . . . . .>4kV
ESD (Machine model) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .>300V
ESD (Charged device model) . . . . . . . . . . . . . . . . . . . . . . . . . .>2kV
Thermal Resistance
JA (°C/W)
JC (°C/W)
QFN Package (Notes 1, 2). . . . . . . . . .
32
3.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . - 65°C to +150°C
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . +300°C
Operating Conditions
Supply Voltage, VCC (5V bias mode, Note 3) . . . . . . . . . . +5V ±5%
Ambient Temperature (ISL6564ACRZ) . . . . . . . . . . . . 0°C to +70°C
Ambient Temperature (ISL6564AIRZ) . . . . . . . . . . . .-40°C to +85°C
CAUTION: Stress above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational section of this specification is not implied.
NOTES:
1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Operating Conditions: VCC = 5V or ICC < 25mA (Note 3). Unless Otherwise Specified.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
VCC SUPPLY CURRENT
Nominal Supply
VCC = 5VDC; EN = 5VDC; RT = 100k
ISEN1 = ISEN2 = ISEN3 = ISEN4 = -70A
-
14
18
mA
Shutdown Supply
VCC = 5VDC; EN = 0VDC; RT = 100k
-
10
14
mA
VCC rising
4.20
4.31
4.50
V
VCC falling
3.60
3.80
4.00
V
EN rising
1.26
1.29
1.32
V
Hysteresis
110
125
135
mV
Fault reset
1.12
1.16
1.20
V
ENLL Input Logic Low Level
-
-
0.4
V
ENLL input Logic High Level
0.8
-
-
V
-
-
1
A
(Note 4)
-0.5
-
0.5
%VID
System Accuracy (VID = 0.525V-0.9875V) (Note 4)
-0.9
-
0.9
%VID
VID Pull-Up
-55
-45
-35
A
VID Input Low Level
-
-
0.4
V
VID Input High Level
0.8
-
-
V
-200
-
200
A
VID Input Voltage when Floated
1.0
1.15
1.30
V
REF Source/Sink Current
-50
-
50
A
Offset resistor connected to ground
388
400
412
mV
VCC = 5.000V, offset resistor connected to VCC
2.91
3.0
3.09
V
-
-
50
A
POWER-ON RESET AND ENABLE
POR Threshold
ENABLE Threshold
ENLL Leakage Current
ENLL = 5V
REFERENCE VOLTAGE AND DAC
System Accuracy (VID = 1.V-1.3V)
DAC Source/Sink Current
VID = 010100
PIN-ADJUSTABLE OFFSET
Voltage at OFS Pin
Maximum OFS Source and Sink Current
8
FN6285.1
March 20, 2007
ISL6564A
Electrical Specifications
Operating Conditions: VCC = 5V or ICC < 25mA (Note 3). Unless Otherwise Specified. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
260
300
345
kHz
OSCILLATOR
Accuracy
RT = 100k
Adjustment Range
0.08
-
1.5
MHz
Sawtooth Amplitude
-
2
-
V
Max Duty Cycle
-
66.7
-
%
ERROR AMPLIFIER
Open-Loop Gain
RL = 10k to ground
-
80
-
dB
Open-Loop Bandwidth
CL = 100pF, RL = 10k to ground
-
18
-
MHz
Maximum Output Voltage
4.0
4.3
-
V
Output High Voltage @ 2mA
3.7
-
-
V
Output Low Voltage @ 2mA
-
-
1.35
V
-
20
-
MHz
2.48
2.50
2.52
V
REMOTE-SENSE AMPLIFIER
Bandwidth
Output Voltage @ 1mA Load
VSEN - RGND = 2.5V
PWM OUTPUT
PWM Output Voltage LOW
Iload = ±500A
-
-
0.5
V
PWM Output Voltage HIGH
Iload = ±500A
4.0
-
-
V
DRIVER ENABLE OUTPUT
DRVEN Output Voltage LOW
Iload = ±1mA
-
-
0.3
V
DRVEN Output Voltage HIGH
Iload = ±1mA
4.0
-
-
V
ISEN1 = ISEN2 = ISEN3 = ISEN4 = 80A
75
85
93
A
95
110
125
A
2
V
SENSE CURRENT OUTPUT
Sensed Current Accuracy
Overcurrent Trip Level
Maximum Voltage at IDROOP and IOUT
VCC = 4.5V (Note 5)
POWER GOOD AND PROTECTION MONITORS
PGOOD Low Voltage
IPGOOD = 4mA
-
-
0.3
V
Undervoltage Offset From VID
VSEN falling
70
75
80
%VID
Overvoltage Threshold
Voltage above VID, after soft-start (Note 6)
180
200
230
mV
Before enable
1.40
1.45
1.50
V
VCC < POR threshold
1.7
1.8
1.87
V
VCC  POR threshold, VSEN falling
-
0.6
-
V
VCC < POR threshold
-
1.5
-
V
3.0
3.6
5.0
V
Overvoltage Reset Voltage
OVP Drive Voltage
IOVP = -10mA, VCC = 5V
NOTES:
3. When using the internal shunt regulator, VCC is clamped to 6.2V (max). Current must be limited to 25mA or less.
4. These parts are designed and adjusted for accuracy with all errors in the voltage loop included.
5. Guaranteed by design.
6. During soft-start, VDAC rises from 0 to VID. The overvoltage trip level is the higher of 1.5V and VDAC + 0.2V.
9
FN6285.1
March 20, 2007
ISL6564A
Functional Pin Description
FB and COMP
Supplies all the power necessary to operate the chip. The
controller starts to operate when the voltage on this pin
exceeds the rising POR threshold and shuts down when the
voltage on this pin drops below the falling POR threshold.
Connect this pin directly to a +5V supply or through a series
300 resistor to a +12V supply.
Inverting input and output of the error amplifier, respectively.
FB is connected to VDIFF through a resistor. A negative
current, proportional to output current is present on the FB
pin. A properly sized resistor between VDIFF and FB sets
the load line (droop). The droop scale factor is set by the
ratio of the ISEN resistors and the lower MOSFET rDS(ON).
COMP is tied back to FB through an external R-C network
with no DC connection to compensate the regulator.
GND
DAC and REF
Bias and reference ground for the IC.
The DAC output pin is the output of the precision internal
DAC reference. The REF input pin is the positive input of the
Error Amp. In typical applications, a 1k, 1% resistor is used
between DAC and REF to generate a precise offset voltage.
This voltage is proportional to the offset current determined
by the offset resistor from OFS to ground or VCC. A
capacitor is used between REF and ground to smooth the
voltage transition during Dynamic VID™ operations.
VCC
EN
This pin is a threshold-sensitive enable input for the
controller. Connecting the 12V supply to EN through an
appropriate resistor divider provides a means to synchronize
power-up of the controller and the MOSFET driver ICs.
When EN is driven above 1.29V, the ISL6564A is active
depending on status of ENLL, the internal POR, and pending
fault states. Driving EN below 1.16V will clear all fault states
and prime the ISL6564A to soft-start when re-enabled.
ENLL
This pin is a logic-level enable input for the controller. When
asserted to a logic high, the ISL6564A is active depending
on status of EN, the internal POR, VID inputs and pending
fault states. Deasserting ENLL will clear all fault states and
prime the ISL6564A to soft-start when re-enabled.
When floating, ENLL pin will be pulled to high internally with
a typical voltage as 1.15V.
FS
A resistor, RT, placed from FS to ground will set the
switching frequency. There is an inverse relationship
between the value of the resistor and the switching
frequency. See Figure 20 and Equation 30.
VID5, VID4, VID3, VID2, VID1, and VID0
These are the inputs to the internal DAC that provides the
reference voltage for output regulation. Connect these pins
either to open-drain outputs with or without external pull-up
resistors or to active-pull-up outputs. VID5-VID0 have 45µA
internal pull-up current sources that diminish to zero as the
voltage rises above the logic-high level. These inputs can be
pulled up as high as VCC plus 0.3V.
VDIFF, VSEN, and RGND
VSEN and RGND form the precision differential remotesense amplifier. This amplifier converts the differential
voltage of the remote output to a single-ended voltage
referenced to local ground. VDIFF is the amplifier’s output
and the input to the regulation and protection circuitry.
Connect VSEN and RGND to the sense pins of the remote
load.
10
PWM1, PWM2, PWM3, PWM4
Pulse-width modulation outputs. Connect these pins to the
PWM input pins of the Intersil driver IC. The number of
active channels is determined by the state of PWM3 and
PWM4. Leave PWM4 unconnected and tie PWM3 to VCC to
configure for 2-phase operation. Tie PWM4 to VCC to
configure for 3-phase operation. Tie both PWM4 and PWM3
to high for 1-phase operation.
ISEN1+, ISEN1-; ISEN2+, ISEN2-; ISEN3+, ISEN3-;
ISEN4+, ISEN4The ISEN+ and ISEN- pins are current sense inputs to
individual differential amplifiers. The sensed current is used
as a reference for channel balancing, protection, and
regulation. Inactive channels should have their respective
sense inputs left open (for example, for 3-phase operation
open ISEN4+).
For DCR sensing, connect each ISEN- pin to the node
between the RC sense elements. Tie the ISEN+ pin to the
other end of the sense capacitor through a resistor, RISEN.
The voltage across the sense capacitor is proportional to the
inductor current. The sensed current is proportional to the
output current, and scaled by the DCR of the inductor,
divided by RISEN.
When configured for rDS(ON) current sensing, the ISEN1-,
ISEN2-, ISEN3-, and ISEN4- pins are grounded at the lower
MOSFET sources. The ISEN1+, ISEN2+, ISEN3+, and
ISEN4+ pins are then held at a virtual ground, such that a
resistor connected between them, and the drain terminal of
the associated lower MOSFET, will carry a current
proportional to the current flowing through that channel. The
current is determined by the negative voltage developed
across the lower MOSFET’s rDS(ON), which is the channel
current scaled by rDS(ON).
FN6285.1
March 20, 2007
ISL6564A
PGOOD
PGOOD is used as an indication of the end of soft-start per
the microprocessor specification. It is an open-drain logic
output that is low impedance until the soft-start is completed.
It will be pulled low again once the undervoltage point is
reached.
IL1 + IL2 + IL3, 7A/DIV
IL3, 7A/DIV
PWM3, 5V/DIV
OFS
The OFS pin provides a means to program a DC offset
current for generating a DC offset voltage at the REF input.
The offset current is generated via an external resistor and
precision internal voltage references. The polarity of the
offset is selected by connecting the resistor to GND or VCC.
For no offset, the OFS pin should be left unterminated.
IL2, 7A/DIV
PWM2, 5V/DIV
IL1, 7A/DIV
PWM1, 5V/DIV
OVP
1µs/DIV
Overvoltage protection pin. This pin pulls to VCC and is
latched when an overvoltage condition is detected. Connect
this pin to the gate of an SCR or MOSFET tied from VIN or
VOUT to ground to prevent damage to the load. This pin may
be pulled above VCC as high as 15V to ground with an
external resistor. However, it is only capable of pulling low
when VCC is above 2V.
DRVEN
Driver enable pin. This pin can be used to enable the drivers
which have enable pins such as ISL6605 or ISL6608. If
ISL6564A is used with Intersil ISL6612 drivers, it’s not
necessary to use this pin.
IDROOP and IOUT
IDROOP and IOUT are the output pins of sensed average
channel current which is proportional to load current. They
are designed for flexible application purposes.
In the application which does not require loadline, leave
IDROOP pin open. In the application which requires load
line, connect IDROOP pin to FB so that the sensed average
current will flow through the resistor between FB and VDIFF
to create a voltage drop which is proportional to load current.
IOUT is typically used for load current indication.
Operation
Multiphase Power Conversion
Microprocessor load current profiles have changed to the
point that the advantages of multiphase power conversion
are impossible to ignore. The technical challenges
associated with producing a single-phase converter which is
both cost-effective and thermally viable have forced a
change to the cost-saving approach of multiphase. The
ISL6564A controller helps reduce the complexity of
implementation by integrating vital functions and requiring
minimal output components. The block diagrams on pages
3, 4, 5, 6, and 7 provide top level views of multiphase power
conversion using the ISL6564A controller.
11
FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS
FOR 3-PHASE CONVERTER
Interleaving
The switching of each channel in a multiphase converter is
timed to be symmetrically out of phase with each of the other
channels. In a 3-phase converter, each channel switches 1/3
cycle after the previous channel and 1/3 cycle before the
following channel. As a result, the three-phase converter has
a combined ripple frequency three times greater than the
ripple frequency of any one phase. In addition, the peak-topeak amplitude of the combined inductor currents is reduced
in proportion to the number of phases (Equations 1 and 2).
Increased ripple frequency and lower ripple amplitude mean
that the designer can use less per-channel inductance and
lower total output capacitance for any performance
specification.
Figure 1 illustrates the multiplicative effect on output ripple
frequency. The three channel currents (IL1, IL2, and IL3)
combine to form the AC ripple current and the DC load
current. The ripple component has three times the ripple
frequency of each individual channel current. Each PWM
pulse is terminated 1/3 of a cycle after the PWM pulse of the
previous phase. The peak-to-peak current for each phase is
about 7A, and the DC components of the inductor currents
combine to feed the load.
To understand the reduction of ripple current amplitude in the
multiphase circuit, examine the equation representing an
individual channel’s peak-to-peak inductor current.
 V IN – V OUT  V OUT
I PP = ----------------------------------------------------L fS V
(EQ. 1)
IN
In Equation 1, VIN and VOUT are the input and output
voltages respectively, L is the single-channel inductor value,
and fS is the switching frequency.
FN6285.1
March 20, 2007
ISL6564A
PWM Operation
INPUT-CAPACITOR CURRENT, 10A/DIV
CHANNEL 3
INPUT CURRENT
10A/DIV
CHANNEL 2
INPUT CURRENT
10A/DIV
CHANNEL 1
INPUT CURRENT
10A/DIV
1µs/DIV
FIGURE 2. CHANNEL INPUT CURRENTS AND INPUTCAPACITOR RMS CURRENT FOR 3-PHASE
CONVERTER
The output capacitors conduct the ripple component of the
inductor current. In the case of multiphase converters, the
capacitor current is the sum of the ripple currents from each
of the individual channels. Compare Equation 1 to the
expression for the peak-to-peak current after the summation
of N symmetrically phase-shifted inductor currents in
Equation 2. Peak-to-peak ripple current decreases by an
amount proportional to the number of channels. Outputvoltage ripple is a function of capacitance, capacitor
equivalent series resistance (ESR), and inductor ripple
current. Reducing the inductor ripple current allows the
designer to use fewer or less costly output capacitors.
 V IN – N V OUT  V OUT
I C, PP = ----------------------------------------------------------L fS V
(EQ. 2)
IN
Another benefit of interleaving is to reduce input ripple
current. Input capacitance is determined in part by the
maximum input ripple current. Multiphase topologies can
improve overall system cost and size by lowering input ripple
current and allowing the designer to reduce the cost of input
capacitance. The example in Figure 2 illustrates input
currents from a three-phase converter combining to reduce
the total input ripple current.
The converter depicted in Figure 2 delivers 36A to a 1.5V load
from a 12V input. The RMS input capacitor current is 5.9A.
Compare this to a single-phase converter also stepping down
12V to 1.5V at 36A. The single-phase converter has 11.9A
RMS input capacitor current. The single-phase converter
must use an input capacitor bank with twice the RMS current
capacity as the equivalent three-phase converter.
Figures 21, 22 and 23 in the section entitled Input Capacitor
Selection can be used to determine the input-capacitor RMS
current based on load current, duty cycle, and the number of
channels. They are provided as aids in determining the
optimal input capacitor solution. Figure 24 shows the single
phase input-capacitor RMS current for comparison.
12
The timing of each converter leg is set by the number of
active channels. The default channel setting for the
ISL6564A is four. One switching cycle is defined as the time
between PWM1 pulse termination signals. The pulse
termination signal is an internally generated clock signal
which triggers the falling edge of PWM1. The cycle time of
the pulse termination signal is the inverse of the switching
frequency set by the resistor between the FS pin and
ground. Each cycle begins when the clock signal commands
the channel 1 PWM output to go low. The PWM1 transition
signals the channel-1 MOSFET driver to turn off the
channel 1 upper MOSFET and turn on the channel 1
synchronous MOSFET. In the default channel configuration,
the PWM2 pulse terminates 1/4 of a cycle after PWM1. The
PWM3 output follows another 1/4 of a cycle after PWM2.
PWM4 terminates another 1/4 of a cycle after PWM3.
If PWM3 is connected to VCC, two channel operation is
selected and the PWM2 pulse terminates 1/2 of a cycle later.
Connecting PWM4 to VCC selects three channel operation
and the pulse-termination times are spaced in 1/3 cycle
increments. Connecting both PWM3 and PWM4 to VCC
selects single-channel operation.
Once a PWM signal transitions low, it is held low for a
minimum of 1/3 cycle. This forced off time is required to
ensure an accurate current sample. Current sensing is
described in the next section. After the forced off time
expires, the PWM output is enabled. The PWM output state
is driven by the position of the error amplifier output signal,
VCOMP, minus the current correction signal relative to the
sawtooth ramp as illustrated in Figure 7. When the modified
VCOMP voltage crosses the sawtooth ramp, the PWM output
transitions high. The MOSFET driver detects the change in
state of the PWM signal and turns off the synchronous
MOSFET and turns on the upper MOSFET. The PWM signal
will remain high until the pulse termination signal marks the
beginning of the next cycle by triggering the PWM signal low.
Current Sampling
During the forced off-time following a PWM transition low,
the associated channel current sense amplifier uses the
ISEN inputs to reproduce a signal proportional to the
inductor current, IL. No matter the current sense method, the
sense current, ISEN, is simply a scaled version of the
inductor current. Coincident with the falling edge of the PWM
signal, the sample and hold circuitry samples ISEN, as
illustrated in Figure 3. The sample window hold time, tHOLD,
is fixed and equal to 1/3 of the switching period, tSW.
t SW
1
t HOLD = ---------- = -----------------3
3  f SW
(EQ. 3)
Therefore, the sample current, In, is proportional to the
output current and held for one switching cycle. The sample
current is used for current balance, load-line regulation, and
overcurrent protection.
FN6285.1
March 20, 2007
ISL6564A
VIN
I s
L
IL
L
ISL6207
DCR
INDUCTOR
+
+
VC(s)
R
ISEN
PWM(n)
COUT
-
VL
-
PWM
VOUT
C
ISL6564A INTERNAL CIRCUIT
tHOLD
SAMPLE CURRENT, In
RISEN(n)
(PTC)
In
SWITCHING PERIOD
SAMPLE
&
HOLD
TIME
ISEN-(n)
+
FIGURE 3. SAMPLE AND HOLD TIMING
-
ISEN+(n)
Current Sensing
The ISL6564A supports inductor DCR sensing, MOSFET
rDS(ON) sensing, or resistive sensing techniques. The
internal circuitry, shown in Figures 4, 5, and 6, represents
channel n of an N-channel converter. This circuitry is
repeated for each channel in the converter, but may not be
active depending on the status of the PWM3 and PWM4
pins, as described in the PWM Operation section.
INDUCTOR DCR SENSING
An inductor’s winding is characteristic of a distributed
resistance as measured by the DCR (Direct Current
Resistance) parameter. Consider the inductor DCR as a
separate lumped quantity, as shown in Figure 4. The
channel current IL, flowing through the inductor, will also
pass through the DCR. Equation 4 shows the s-domain
equivalent voltage across the inductor VL.
V L = I L   s  L + DCR 
(EQ. 4)
DCR
I SEN = I ----------------LR
ISEN
FIGURE 4. DCR SENSING CONFIGURATION
The capacitor voltage VC, is then replicated across the
sense resistor RISEN. The current through the sense resistor
is proportional to the inductor current. Equation 6 shows the
proportion between the channel current and the sensed
current ISEN, is driven by the value of the sense resistor
chosen and the DCR of the inductor.
DCR
I SEN = I L  -----------------R
ISEN
(EQ. 6)
DCR varies with temperature, so a Positive Temperature
Coefficient (PTC) resistor should be selected for the sense
resistor RISEN.
RESISTIVE SENSING
A simple R-C network across the inductor extracts the DCR
voltage, as shown in Figure 4.
The voltage on the capacitor VC, can be shown to be
proportional to the channel current IL, see Equation 5.
L
 s  ------------+ 1   DCR  I L 
 DCR

V C = -------------------------------------------------------------------- s  RC + 1 
If DCR sensing is not utilized, independent current-sense
resistors in series with each output inductor can serve as the
sense element (see Figure 5). This technique is more
accurate, but reduces overall converter efficiency due to the
addition of a lossy element directly in the output path.
(EQ. 5)
If the R-C network components are selected such that the
RC time constant matches the inductor L/DCR time
constant, then VC is equal to the voltage drop across the
DCR.
13
FN6285.1
March 20, 2007
ISL6564A
I
L
Channel-Current Balance
L
RSENSE VOUT
COUT
ISL6564A INTERNAL CIRCUIT
RISEN(n)
In
SAMPLE
&
HOLD
ISEN-(n)
+
-
ISEN+(n)
I
R SENSE
SEN = I L ------------------------R
ISEN
FIGURE 5. SENSE RESISTOR IN SERIES WITH INDUCTORS
The sampled currents In, from each active channel are
summed together and divided by the number of active
channels. The resulting cycle average current IAVG, provides
a measure of the total load current demand on the converter
during each switching cycle. Channel current balance is
achieved by comparing the sampled current of each channel
to the cycle average current, and making an appropriate
adjustment to each channel pulse width based on the error.
Intersil’s patented current-balance method is illustrated in
Figure 7, with error correction for channel 1 represented. In
the figure, the cycle average current combines with the
channel 1 sample, I1, to create an error signal IER. The
filtered error signal modifies the pulse width commanded by
VCOMP to correct any unbalance and force IER toward zero.
The same method for error signal correction is applied to
each active channel.
MOSFET rDS(ON) SENSING
The controller can also sense the channel load current by
sampling the voltage across the lower MOSFET rDS(ON)
(see Figure 6). The amplifier is ground-reference by
connecting the ISEN- input to the source of the lower
MOSFET. ISEN+ connects to the PHASE node through a
resistor RISEN. The voltage across RISEN is equivalent to
the voltage drop across the rDS(ON) of the lower MOSFET
while it is conducting. The resulting current into the ISEN+
pin is proportional to the channel current IL. The ISEN
current is then sampled and held after sufficient settling time.
The sampled current In, is used for channel-current balance,
load-line regulation, and overcurrent protection. From
Figure 6, Equation 7 for ISEN is derived.
VIN
r
DS  ON 
I SEN = I ------------------------L R
ISEN
In
IL
SAMPLE
&
HOLD
ISEN+(n)
RISEN
(PTC)
-
ISEN-(n)
I r
L DS  ON 
+
+
N-CHANNEL
MOSFETs
ISL6564A INTERNAL CIRCUIT
EXTERNAL CIRCUIT
FIGURE 6. MOSFET rDS(ON) CURRENT-SENSING CIRCUIT
r DS  ON 
I SEN = I L ---------------------R ISEN
(EQ. 7)
where IL is the channel current. Since MOSFET rDS(ON)
increases with temperature, a PTC resistor should be
chosen for RISEN to compensate for this change.
14
VCOMP
+
+
-
FILTER
PWM1
SAWTOOTH SIGNAL
f(j)
I4 *
IER
IAVG
-
+
N

I3 *
I2
I1
NOTE: *Channels 3 and 4 are optional.
FIGURE 7. CHANNEL 1 PWM FUNCTION AND CURRENTBALANCE ADJUSTMENT
Channel current balance is essential in realizing the thermal
advantage of multiphase operation. The heat generated in
down converting is dissipated over multiple devices and a
greater area. The designer avoids the complexity of driving
multiple parallel MOSFETs, and the expense of using heat
sinks and nonstandard magnetic materials.
Voltage Regulation
The integrating compensation network shown in Figure 8
assures that the steady-state error in the output voltage is
limited only to the error in the reference voltage (output of
the DAC) and offset errors in the OFS current source,
remote-sense and error amplifiers. Intersil specifies the
guaranteed tolerance of the ISL6564A to include the
combined tolerances of each of these elements.
The output of the error amplifier, VCOMP, is compared to the
sawtooth waveform to generate the PWM signals. The PWM
signals control the timing of the Intersil MOSFET drivers and
regulate the converter output to the specified reference
voltage. The internal and external circuitry which control
voltage regulation is illustrated in Figure 8.
FN6285.1
March 20, 2007
ISL6564A
EXTERNAL CIRCUIT
RC CC
COMP
RREF
VID5
VID4
VID3
VID2
VID1
VID0
DAC
400
mV
200
mV
100
mV
50
mv
25
mV
12.5
mV
REF
1
1
1
1
1
1
OFF
1
1
1
1
1
0
1.3000V
1
1
1
1
0
1
1.2875V
1
1
1
1
0
0
1.2750V
1
1
1
0
1
1
1.2625V
1
1
1
0
1
0
1.2500V
1
1
1
0
0
1
1.2375V
1
1
1
0
0
0
1.2250V
1
1
0
1
1
1
1.2125V
1
1
0
1
1
0
1.2000V
1
1
0
1
0
1
1.1875V
1
1
0
1
0
0
1.1750V
1
1
0
0
1
1
1.1625V
1
1
0
0
1
0
1.1500V
1
1
0
0
0
1
1.1375V
1
1
0
0
0
0
1.1250v
1
0
1
1
1
1
1.1125V
1
0
1
1
1
0
1.1000V
1
0
1
1
0
1
1.0875V
1
0
1
1
0
0
1.0750V
1
0
1
0
1
1
1.0625V
1
0
1
0
1
0
1.0500V
1
0
1
0
0
1
1.0375V
1
0
1
0
0
0
1.0250V
1
0
0
1
1
1
1.0125V
1
0
0
1
1
0
1.0000V
1
0
0
1
0
1
0.9875V
1
0
0
1
0
0
0.9750V
1
0
0
0
1
1
0.9625V
1
0
0
0
1
0
0.9500V
1
0
0
0
0
1
0.9375V
1
0
0
0
0
0
0.9250V
0
1
1
1
1
1
0.9125V
0
1
1
1
1
0
0.9000V
0
1
1
1
0
1
0.8875V
0
1
1
1
0
0
0.8750V
CREF
+
-
FB
RFB
IDROOP
+
VDROOP
VDIFF
VOUT+
VOUT-
TABLE 1. VOLTAGE IDENTIFICATION (VID) CODES
ISL6564A INTERNAL CIRCUIT
IAVG
VCOMP
ERROR AMPLIFIER
VSEN
+
-
RGND
DIFFERENTIAL
REMOTE-SENSE
AMPLIFIER
FIGURE 8. OUTPUT VOLTAGE AND LOAD-LINE
REGULATION WITH OFFSET ADJUSTMENT
The ISL6564A incorporates an internal differential remotesense amplifier in the feedback path. The amplifier removes
the voltage error encountered when measuring the output
voltage relative to the local controller ground reference point
resulting in a more accurate means of sensing output
voltage. Connect the microprocessor sense pins to the
non-inverting input, VSEN, and inverting input, RGND, of the
remote-sense amplifier. The remote-sense output, VDIFF, is
connected to the inverting input of the error amplifier through
an external resistor.
A digital to analog converter (DAC) generates a reference
voltage based on the state of logic signals at pins VID4
through VID12.5. The DAC decodes the a 6-bit logic signal
(VID) into one of the discrete voltages shown in Table 1.
Each VID input offers a 45A pull-up to an internal 2.5V
source for use with open-drain outputs. The pull-up current
diminishes to zero above the logic threshold to protect
voltage-sensitive output devices. External pull-up resistors
can augment the pull-up current sources if case leakage into
the driving device is greater than 45A.
Load-Line Regulation
Some microprocessor manufacturers require a preciselycontrolled output resistance. This dependence of output
voltage on load current is often termed “droop” or “load line”
regulation. By adding a well controlled output impedance,
the output voltage can effectively be level shifted in a
direction which works to achieve the load-line regulation
required by these manufacturers.
15
VDAC
FN6285.1
March 20, 2007
ISL6564A
TABLE 1. VOLTAGE IDENTIFICATION (VID) CODES (Continued)
VDAC
down so that a larger positive spike can be sustained without
crossing the upper specification limit.
VID5
VID4
VID3
VID2
VID1
VID0
400
mV
200
mV
100
mV
50
mv
25
mV
12.5
mV
0
1
1
0
1
1
0.8625V
0
1
1
0
1
0
0.8500V
0
1
1
0
0
1
0.8375V
As shown in Figure 8, a current proportional to the average
current in all active channels, IAVG, flows from FB through a
load-line regulation resistor, RFB. The resulting voltage drop
across RFB is proportional to the output current, effectively
creating an output voltage droop with a steady-state value
defined as
0
1
1
0
0
0
0.8250V
V DROOP = I AVG R FB
0
1
0
1
1
1
0.8125V
0
1
0
1
1
0
0.8000V
0
1
0
1
0
1
0.7875V
0
1
0
1
0
0
0.7750V
0
1
0
0
1
1
0.7625V
0
1
0
0
1
0
0.7500V
0
1
0
0
0
1
0.7375V
0
1
0
0
0
0
0.7250V
0
0
1
1
1
1
0.7125V
0
0
1
1
1
0
0.7000V
0
0
1
1
0
1
0.6875V
0
0
1
1
0
0
0.6750V
0
0
1
0
1
1
0.6625V
0
0
1
0
1
0
0.6500V
0
0
1
0
0
1
0.6375V
0
0
1
0
0
0
0.6250V
0
0
0
1
1
1
0.6125V
0
0
0
1
1
0
0.6000V
0
0
0
1
0
1
0.5875V
0
0
0
1
0
0
0.5750V
0
0
0
0
1
1
0.5625V
0
0
0
0
1
0
0.5500V
0
0
0
0
0
1
0.5375V
0
0
0
0
0
0
0.525V
In other cases, the designer may determine that a more
cost-effective solution can be achieved by adding droop.
Droop can help to reduce the output-voltage spike that
results from fast load-current demand changes.
The magnitude of the spike is dictated by the ESR and ESL
of the output capacitors selected. By positioning the no-load
voltage level near the upper specification limit, a larger
negative spike can be sustained without crossing the lower
limit. By adding a well controlled output impedance, the
output voltage under load can effectively be level shifted
16
(EQ. 8)
The regulated output voltage is reduced by the droop voltage
VDROOP. The output voltage as a function of load current is
derived by combining Equation 8 with the appropriate
sample current expression defined by the current sense
method employed.
 I OUT R X

- ------------------ R FB
V OUT = V REF – V OFFSET –  ----------- 4 R ISEN

(EQ. 9)
Where VREF is the reference voltage, VOFS is the
programmed offset voltage, IOUT is the total output current
of the converter, RISEN is the sense resistor in the ISEN line,
and RFB is the feedback resistor. RX has a value of DCR,
rDS(ON), or RSENSE depending on the sensing method.
Output-Voltage Offset Programming
The ISL6564A allows the designer to accurately adjust the
offset voltage. When a resistor, ROFS, is connected between
OFS to VCC, the voltage across it is regulated to 2.0V. This
causes a proportional current (IOFS) to flow into OFS. If
ROFS is connected to ground, the voltage across it is
regulated to 0.5V, and IOFS flows out of OFS. A resistor
between DAC and REF, RREF, is selected so that the
product (IOFS x ROFS) is equal to the desired offset voltage.
These functions are shown in Figure 9.
As it may be noticed in Figure 9, the OFSOUT pin must be
connected to the REF pin for this current injection to function
in ISL6564A. The current flow through RREF creates an
offset at the REF pin, which is ultimately duplicated at the
output of the regulator.
Once the desired output offset voltage has been determined,
use the following formulas to set ROFS:
For Positive Offset (connect ROFS to VCC):
2  R REF
R OFS = -------------------------V OFFSET
(EQ. 10)
For Negative Offset (connect ROFS to GND):
0.5  R REF
R OFS = -----------------------------V OFFSET
(EQ. 11)
FN6285.1
March 20, 2007
ISL6564A
Assuming the microprocessor controls the VID change at 1
bit every TVID, the relationship between the time constant of
RREF and CREF network and TVID is given by Equation 12.
FB
DYNAMIC
VID D/A
DAC
RREF
E/A
REF
C REF R REF = k T VID
(EQ. 12)
Where, TVID = 4s, k is the number of the internal VID
change cycle. If Typically RREF is selected to be 1k, the
allowable delay time for VR to respond to new VID code is 5
VID change cycles (totally 20s), the value of CREF should
be 22nF based on Equation 12.
Operation Initialization
VCC
OR
GND
2.0V
-
ROFS
+
+
0.5V
VCC
-
ISL6564A
OFS
GND
FIGURE 9. OUTPUT VOLTAGE OFFSET PROGRAMMING
WITH ISL6564A
Prior to converter initialization, proper conditions must exist
on the enable inputs and VCC. When the conditions are met,
the controller begins soft-start. Once the output voltage is
within the proper window of operation, PGOOD asserts
logic 1.
Enable and Disable
While in shutdown mode, the PWM outputs are held in a
high-impedance state to assure the drivers remain off. The
following input conditions must be met before the ISL6564A
is released from shutdown mode.
ISL6564A INTERNAL CIRCUIT
Dynamic VID
Modern microprocessors need to make changes to their
core voltage as part of normal operation. They direct the
core-voltage regulator to do this by making changes to the
VID inputs during regulator operation. The power
management solution is required to monitor the DAC inputs
and respond to on-the-fly VID changes in a controlled
manner. Supervising the safe output voltage transition within
the DAC range of the processor without discontinuity or
disruption is a necessary function of the core-voltage
regulator.
The ISL6564A checks the VID inputs at the three edges of
16MHz clock. If the VID code is found to have changed, the
controller waits half of a complete cycle before executing a
12.5mV change. If during the half-cycle wait period, the
difference between DAC level and the new VID code
changes sign, no change is made. If the VID code is more
than 1 bit higher or lower than the DAC (not recommended),
the controller will execute step-up and step down VID
change at a speed of 12.5mV every 4s until VID and DAC
are equal.
In order to ensure the smooth transition of output voltage
during VID change, a VID step change smoothing network
composed of RREF and CREF is required for an ISL6564A
based voltage regulator. The selection of RREF is based on
the desired offset as detailed above in Output-Voltage Offset
Programming. The selection of CREF is based on the time
duration for 1 bit VID change and the allowable delay time.
17
EXTERNAL CIRCUIT
+12V
VCC
POR
CIRCUIT
10.7k
ENABLE
COMPARATOR
EN
+
-
1.40k
1.23V
ENLL
SOFT-START
AND
FAULT LOGIC
FIGURE 10. POWER SEQUENCING USING THRESHOLDSENSITIVE ENABLE (EN) FUNCTION
1. The bias voltage applied at VCC must reach the internal
power-on reset (POR) rising threshold. Once this
threshold is reached, proper operation of all aspects of
the ISL6564A is guaranteed. Hysteresis between the
rising and falling thresholds assure that once enabled,
the ISL6564A will not inadvertently turn off unless the
bias voltage drops substantially (see Electrical
Specifications).
FN6285.1
March 20, 2007
ISL6564A
2. The ISL6564A features an enable input (EN) for power
sequencing between the controller bias voltage and
another voltage rail. The enable comparator holds the
ISL6564A in shutdown until the voltage at EN rises above
1.29V. The enable comparator has about 125mV of
hysteresis to prevent bounce. It is important that the
driver ICs reach their POR level before the ISL6564A
becomes enabled. The schematic in Figure 10
demonstrates sequencing the ISL6564A with the
ISL66Xx family of Intersil MOSFET drivers, which require
12V bias.
VOUT, 500mV/DIV
EN, 5V/DIV
3. The voltage on ENLL must be logic high to enable the
controller. This pin is typically connected to the
VID_PGOOD.
4. The VID code must not be 111111. This code signals the
controller that no load is present. The controller will enter
shut-down mode after receiving this code and will
execute soft-start upon receiving any other code. This
code can be used to enable or disable the controller but
it is not recommended. After receiving this code, the
controller executes a 2-cycle delay before changing the
overvoltage trip level to the shut-down level and disabling
PWM. Overvoltage shutdown can not be reset using this
code.
To enable the controller, VCC must be greater than the POR
threshold; the voltage on EN must be greater than 1.29V;
For ISL6564A, ENLL must be logic high; and VID cannot be
equal to 111111. When each of these conditions is true, the
controller immediately begins the soft-start sequence.
500µs/DIV
FIGURE 11. SOFT-START WAVEFORMS WITH AN UN-BIASED
OUTPUT
Fault Monitoring and Protection
The ISL6564A actively monitors output voltage and current
to detect fault conditions. Fault monitors trigger protective
measures to prevent damage to a microprocessor load. One
common power good indicator is provided for linking to
external system monitors. The schematic in Figure 12
outlines the interaction between the fault monitors and the
power good signal.
Power Good Signal
Soft-Start
During soft-start, the DAC voltage ramps linearly from zero
to the programmed VID level as shown in Figure 11. The
PWM signals remain in the high-impedance state until the
controller detects that the ramping DAC level has reached
the pre-bias output-voltage level. This protects the system
against the large, negative inductor currents that would
otherwise occur when starting with a pre-existing charge on
the output as the controller attempted to regulate to zero
volts at the beginning of the soft-start cycle. The soft-start
time, tSS, begins with a delay period equal to 64 switching
cycles followed by a linear ramp with a fixed rate at a speed
of 12.5mV/32µs.
t SS =  2560 VID
The power good pin (PGOOD) is an open-drain logic output
indication that the converter is operating after soft-start.
PGOOD pulls low during shutdown and releases high after a
successful soft-start. PGOOD will only transition low when
an undervoltage condition is detected or the controller is
disabled by a reset from EN, ENLL, POR, or one of the
no-CPU VID codes. After an undervoltage event, PGOOD
will return high unless the controller has been disabled.
PGOOD does not automatically transition low upon
detection of an overvoltage condition.
(EQ. 13)
Equation 13 can be used to calculate the soft-start time. For
example, when VID is set to 1.2V, the soft-start time will be
3.072ms.
A 100mV offset exists on the remote-sense amplifier at the
beginning of soft-start and ramps to zero during the first 640
cycles of soft-start (704 cycles following enable). This
prevents the large inrush current that would otherwise occur
should the output voltage start out with a slight negative
bias.
18
FN6285.1
March 20, 2007
ISL6564A
PGOOD
OC
-
+
UV
SOFT-START, FAULT
AND CONTROL LOGIC
+
VDIFF
110µA
+
I1
REPEAT FOR
EACH CHANNEL
75%
DAC
REFERENCE
-
OC
-
+
110µA
IAVG
OVP
OV
VID + 0.2V
FIGURE 12. POWER GOOD AND PROTECTION CIRCUITRY
Undervoltage Detection
The undervoltage threshold is set at 75% of the VID code.
When the output voltage at VSEN is below the undervoltage
threshold, PGOOD gets pulled low.
Overvoltage Protection
When VCC is above 1.4V, but otherwise not valid as defined
under Power-on Reset in Electrical Specifications, the
overvoltage trip circuit is active using auxiliary circuitry. In
this state, an overvoltage trip occurs if the voltage at VSEN
exceeds 1.8V.
With valid VCC, the overvoltage circuit is sensitive to the
voltage at VDIFF. In this state, the trip level is 1.7V prior to
valid enable conditions being met as described in Enable
and Disable. The only exception to this is when the IC has
been disabled by an overvoltage trip. In that case the
overvoltage trip point is VID plus 200mV. During soft-start,
the overvoltage trip level is the higher of 1.5V or VID plus
200mV. Upon successful soft-start, the overvoltage trip level
is 200mV above VID. Two actions are taken by the
ISL6564A to protect the microprocessor load when an
overvoltage condition occurs.
At the inception of an overvoltage event, all PWM outputs
are commanded low instantly (less than 20ns) until the
voltage at VSEN falls below 0.6V with valid VCC or 1.5V
otherwise. This causes the Intersil drivers to turn on the
lower MOSFETs and pull the output voltage below a level
that might cause damage to the load. The PWM outputs
remain low until VDIFF falls to the programmed DAC level
when they enter a high-impedance state. The Intersil drivers
respond to the high-impedance input by turning off both
upper and lower MOSFETs. If the overvoltage condition
19
reoccurs, the ISL6564A will again command the lower
MOSFETs to turn on. The ISL6564A will continue to protect
the load in this fashion as long as the overvoltage condition
recurs.
Simultaneous to the protective action of the PWM outputs,
the OVP pin pulls to VCC delivering up to 100mA to the gate
of a crowbar MOSFET or SCR placed either on the input rail
or the output rail. Turning on the MOSFET or SCR collapses
the power rail and causes a fuse placed further up stream to
blow. The fuse must be sized such that the MOSFET or SCR
will not overheat before the fuse blows. The OVP pin is
tolerant to 12V (see Absolute Maximum Ratings), so an
external resistor pull-up can be used to augment the driving
capability. If using a pull-up resistor in conjunction with the
internal overvoltage protection function, care must be taken
to avoid nuisance trips that could occur when VCC is below
2V. In that case, the controller is incapable of holding OVP
low.
Once an overvoltage condition is detected, normal PWM
operation ceases until the ISL6564A is reset. Cycling the
voltage on EN or ENLL or VCC below the POR-falling
threshold will reset the controller. Cycling the VID codes will
not reset the controller.
Overcurrent Protection
ISL6564A has two levels of overcurrent protection. Each
phase is protected from a sustained overcurrent condition on
a delayed basis, while the combined phase currents are
protected on an instantaneous basis.
In instantaneous protection mode, the ISL6564A takes
advantage of the proportionality between the load current
and the average current, IAVG, to detect an overcurrent
condition. See the Channel-Current Balance section for
more detail on how the average current is measured. The
average current is continually compared with a constant
110A reference current as shown in Figure 12. Once the
average current exceeds the reference current, a
comparator triggers the converter to shutdown.
In individual overcurrent protection mode, the ISL6564A
continuously compares the current of each channel with the
same 110A reference current. If any channel current
exceeds the reference current continuously for eight
consecutive cycles, the comparator triggers the converter to
shutdown.
FN6285.1
March 20, 2007
ISL6564A
In normal operation, DRVEN remains low until ISL6564A
begins soft-start ramp and then changes to high (Figure 14).
When an overcurrent event occurs, DRVEN is pulled to low
instantly (less than 20ns) to disable the driver so that both
upper and lower FETs be turned off (Figure 15). During an
overvoltage condition, DRVEN remains high to allow the
driver turn on the lower FETs based on the PWM input to
discharge the energy stored in the output inductor. Once the
Output voltage is reduced to 0.6V, DRVEN is pulled to low as
shown in Figure 16.
OUTPUT CURRENT, 50A/DIV
0A
OUTPUT VOLTAGE,
500mV/DIV
DRVEN, 5V/DIV
0V
0V
2ms/DIV
OUTPUT CURRENT, 50A/DIV
FIGURE 13. OVERCURRENT BEHAVIOR IN HICCUP MODE,
FSW = 500kHz
At the beginning of overcurrent shutdown, the controller
places all PWM signals in a high-impedance state within
20ns commanding the Intersil MOSFET driver ICs to turn off
both upper and lower MOSFETs. The system remains in this
state a period of 4096 switching cycles. If the controller is still
enabled at the end of this wait period, it will attempt a
soft-start. If the fault remains, the trip-retry cycles will
continue indefinitely (as shown in Figure 13) until either
controller is disabled or the fault is cleared. Note that the
energy delivered during trip-retry cycling is much less than
during full-load operation, so there, there is no thermal
hazard during this kind of operation.
0A
OUTPUT VOLTAGE,
500mV/DIV
0V
2ms/DIV
FIGURE 15. DRVEN DURING OVERCURRENT OPERATION
DRVEN, 5V/DIV
DRVEN, 5V/DIV
OUTPUT VOLTAGE,
500mV/DIV
VOUT, 1V/DIV
EN, 5V/DIV
2ms/DIV
FIGURE 16. DRVEN DURING OVERCURRENT OPERATION
500µs/DIV
FIGURE 14. DRVEN WAVEFORM AT START-UP
There’s no need to use DRVEN when ISL6564A is used to
work with Intersil’s drivers such as ISL6612 and ISL6605.
Driver Enable Output
Current Sense Output
The ISL6564A has a driver enable output pin DRVEN. The
DRVEN is designed for the application where ISL6564A
needs to work with drivers that can not recognize three-state
PWM input.
The ISL6564A has 2 current sense output pins IDROOP and
IOUT. They are identical. In typical application, IDROOP pin
is connected to FB pin for the application where load line is
required. IOUT pin was designed for load current
measurement. As shown in typical application schematics
20
FN6285.1
March 20, 2007
ISL6564A
on pages 4 to 7, load current information can be obtained by
measuring the voltage between IOUT to ground when a NTC
network from IOUT pin to the ground is placed. The output
current at IOUT pin is proportional to load current as shown
in Figure 17.
current range. If through-hole MOSFETs and inductors can
be used, higher per-phase currents are possible. In cases
where board space is the limiting constraint, current can be
pushed as high as 40A per phase, but these designs require
heat sinks and forced air to cool the MOSFETs, inductors
and heat-dissipating surfaces.
MOSFETs
The choice of MOSFETs depends on the current each
MOSFET will be required to conduct; the switching
frequency; the capability of the MOSFETs to dissipate heat;
and the availability and nature of heat sinking and air flow.
V_IOUT, 200mV/DIV
LOWER MOSFET POWER CALCULATION
0A
50A
100A
FIGURE 17. VOLTAGE AT IOUT PIN WITH A NTC NETWORK
PLACED BETWEEN IOUT TO GROUND WHEN
LOAD CURRENT CHANGES
When selecting the equivalent resistor network components
values, it is important to ensure the voltage at IOUT pin not
exceed 2V.
When ISL6564A is operated at single phase mode (both
PWM3 and PWM4 connected to VCC and PWM2
unconnected). The output current at IOUT and IDROOP is
half of the sensed phase current.
General Design Guide
This design guide is intended to provide a high-level
explanation of the steps necessary to create a multiphase
power converter. It is assumed that the reader is familiar with
many of the basic skills and techniques referenced below. In
addition to this guide, Intersil provides complete reference
designs that include schematics, bills of materials, and
example board layouts for all common microprocessor
applications.
Power Stages
The first step in designing a multiphase converter is to
determine the number of phases. This determination
depends heavily on the cost analysis which in turn depends
on system constraints that differ from one design to the next.
Principally, the designer will be concerned with whether
components can be mounted on both sides of the circuit
board; whether through-hole components are permitted; and
the total board space available for power-supply circuitry.
Generally speaking, the most economical solutions are
those in which each phase handles between 15 and 20A. All
surface-mount designs will tend toward the lower end of this
21
The calculation for heat dissipated in the lower MOSFET is
simple, since virtually all of the heat loss in the lower
MOSFET is due to current conducted through the channel
resistance (rDS(ON)). In Equation 14, IM is the maximum
continuous output current; IPP is the peak-to-peak inductor
current (see Equation 1); d is the duty cycle (VOUT/VIN); and
L is the per-channel inductance.
I L, 2PP  1 – d 
 I M 2
P LOW 1 = r DS  ON   -----  1 – d  + -------------------------------12
 N
(EQ. 14)
An additional term can be added to the lower-MOSFET loss
equation to account for additional loss accrued during the
dead time when inductor current is flowing through the
lower-MOSFET body diode. This term is dependent on the
diode forward voltage at IM, VD(ON); the switching
frequency, fS; and the length of dead times, td1 and td2, at
the beginning and the end of the lower-MOSFET conduction
interval respectively.
I

I M I PP
M I-------P LOW 2 = V D  ON  f S  ----- t d1 +  ----- – PP- t d2
 N- + -------2 
2 
N
(EQ. 15)
Thus the total maximum power dissipated in each lower
MOSFET is approximated by the summation of PLOW,1 and
PLOW,2.
UPPER MOSFET POWER CALCULATION
In addition to rDS(ON) losses, a large portion of the upperMOSFET losses are due to currents conducted across the
input voltage (VIN) during switching. Since a substantially
higher portion of the upper-MOSFET losses are dependent
on switching frequency, the power calculation is more
complex. Upper MOSFET losses can be divided into
separate components involving the upper-MOSFET
switching times; the lower-MOSFET body-diode reverserecovery charge, Qrr; and the upper MOSFET rDS(ON)
conduction loss.
When the upper MOSFET turns off, the lower MOSFET does
not conduct any portion of the inductor current until the
voltage at the phase node falls below ground. Once the
lower MOSFET begins conducting, the current in the upper
MOSFET falls to zero as the current in the lower MOSFET
FN6285.1
March 20, 2007
ISL6564A
ramps up to assume the full inductor current. In Equation 16,
the required time for this commutation is t1 and the
approximated associated power loss is PUP,1.
I M I PP  t 1 
P UP,1  V IN  -----  ----  f
 N- + -------2  2 S
(EQ. 16)
At turn on, the upper MOSFET begins to conduct and this
transition occurs over a time t2. In Equation 17, the
approximate power loss is PUP,2.
 I M I PP  t 2 
P UP, 2  V IN  ----- – ---------  ----  f S
2  2
N
(EQ. 17)
A third component involves the lower MOSFET’s reverserecovery charge, Qrr. Since the inductor current has fully
commutated to the upper MOSFET before the lowerMOSFET’s body diode can draw all of Qrr, it is conducted
through the upper MOSFET across VIN. The power
dissipated as a result is PUP,3 and is approximately
P UP,3 = V IN Q rr f S
(EQ. 18)
Finally, the resistive part of the upper MOSFET’s is given in
Equation 19 as PUP,4.
The total power dissipated by the upper MOSFET at full load
can now be approximated as the summation of the results
from Equations 16, 17, 18 and 19. Since the power
equations depend on MOSFET parameters, choosing the
correct MOSFETs can be an iterative process involving
repetitive solutions to the loss equations for different
MOSFETs and different switching frequencies.
2
 I M
I PP2
P UP,4  r DS  ON   ----- d + ---------12
 N
(EQ. 19)
Current Sensing Resistor
The resistors connected between these pins and the
respective phase nodes determine the gains in the load-line
regulation loop and the channel-current balance loop as well
as setting the overcurrent trip point. Select values for these
resistors based on the room temperature rDS(ON) of the
lower MOSFETs, DCR of inductor or additional resistor; the
full-load operating current, IFL; and the number of phases, N
using Equation 20.
RX
R ISEN = ---------------------70 10 – 6
I FL
-------N
(EQ. 20)
In certain circumstances, it may be necessary to adjust the
value of one or more ISEN resistor. When the components of
one or more channels are inhibited from effectively
dissipating their heat so that the affected channels run hotter
than desired, choose new, smaller values of RISEN for the
affected phases (see the section entitled Channel-Current
Balance). Choose RISEN,2 in proportion to the desired
22
decrease in temperature rise in order to cause proportionally
less current to flow in the hotter phase.
T
R ISEN ,2 = R ISEN ----------2
T 1
(EQ. 21)
In Equation 21, make sure that T2 is the desired temperature
rise above the ambient temperature, and T1 is the measured
temperature rise above the ambient temperature. While a
single adjustment according to Equation 21 is usually
sufficient, it may occasionally be necessary to adjust RISEN
two or more times to achieve optimal thermal balance
between all channels.
Load-Line Regulation Resistor
The load-line regulation resistor is labeled RFB in Figure 8.
Its value depends on the desired full-load droop voltage
(VDROOP in Figure 8). If Equation 20 is used to select each
ISEN resistor, the load-line regulation resistor is as shown in
Equation 22.
V DROOP
R FB = -----------------------–6
70 10
(EQ. 22)
If one or more of the ISEN resistors is adjusted for thermal
balance, as in Equation 21, the load-line regulation resistor
should be selected according to Equation 23 where IFL is the
full-load operating current and RISEN(n) is the ISEN resistor
connected to the nth ISEN pin.
V DROOP
R FB = -------------------------------I FL r DS  ON 
 RISEN  n 
(EQ. 23)
n
Compensation
The two opposing goals of compensating the voltage
regulator are stability and speed. Depending on whether the
regulator employs the optional load-line regulation as
described in Load-Line Regulation, there are two distinct
methods for achieving these goals.
COMPENSATING LOAD-LINE REGULATED
CONVERTER
The load-line regulated converter behaves in a similar
manner to a peak-current mode controller because the two
poles at the output-filter L-C resonant frequency split with
the introduction of current information into the control loop.
The final location of these poles is determined by the system
function, the gain of the current signal, and the value of the
compensation components, RC and CC.
Since the system poles and zero are affected by the values
of the components that are meant to compensate them, the
solution to the system equation becomes fairly complicated.
Fortunately there is a simple approximation that comes very
close to an optimal solution. Treating the system as though it
were a voltage-mode regulator by compensating the L-C
poles and the ESR zero of the voltage-mode approximation
yields a solution that is always stable with very close to ideal
transient performance.
FN6285.1
March 20, 2007
ISL6564A
C2
CC
FB
+
RFB
RC
COMP
IDROOP
VDROOP
CC
COMP
FB
ISL6564A
RC
C1
R1
RFB
IDROOP
ISL6564A
C2 (OPTIONAL)
VDIFF
VDIFF
FIGURE 18. COMPENSATION CONFIGURATION FOR
LOAD-LINE REGULATED ISL6564A CIRCUIT
The feedback resistor, RFB, has already been chosen as
outlined in Load-Line Regulation Resistor. Select a target
bandwidth for the compensated system, f0. The target
bandwidth must be large enough to assure adequate
transient performance, but smaller than 1/3 of the
per-channel switching frequency. The values of the
compensation components depend on the relationships of f0
to the L-C pole frequency and the ESR zero frequency. For
each of the three cases which follow, there is a separate set
of equations for the compensation components.
Case 1:
1
------------------- > f 0
2 LC
2f 0 V pp LC
R C = R FB ----------------------------------0.75V
IN
0.75V IN
C C = ----------------------------------2V PP R FB f 0
Case 2:
2
R C = R FB -------------------------------------------0.75 V IN
(EQ. 24)
0.75V IN
C C = ----------------------------------------------------------- 2  2 f 02 V PP R FB LC
Case 3:
1
f 0 > -----------------------------2C  ESR 
2 f 0 V pp L
R C = R FB ----------------------------------------0.75 V IN  ESR 
0.75V IN  ESR  C
C C = -----------------------------------------------2V PP R FB f 0 L
In Equation 24, L is the per-channel filter inductance divided
by the number of active channels; C is the sum total of all
output capacitors; ESR is the equivalent-series resistance of
23
the bulk output-filter capacitance; and VPP is the peak-topeak sawtooth signal amplitude as described in Figure 7 and
Electrical Specifications.
The optional capacitor C2, is sometimes needed to bypass
noise away from the PWM comparator (see Figure 18). Keep
a position available for C2, and be prepared to install a highfrequency capacitor of between 22pF and 150pF in case any
leading-edge jitter problem is noted.
Once selected, the compensation values in Equation 24
assure a stable converter with reasonable transient
performance. In most cases, transient performance can be
improved by making adjustments to RC. Slowly increase the
value of RC while observing the transient performance on an
oscilloscope until no further improvement is noted. Normally,
CC will not need adjustment. Keep the value of CC from
Equation 24 unless some performance issue is noted.
COMPENSATION WITHOUT LOAD-LINE REGULATION
1
1
-------------------  f 0 < ----------------------------2C  ESR 
2 LC
V PP  2  f 02 LC
FIGURE 19. COMPENSATION CIRCUIT FOR ISL6564A BASED
CONVERTER WITHOUT LOAD-LINE
REGULATION
The non load-line regulated converter is accurately modeled
as a voltage-mode regulator with two poles at the L-C
resonant frequency and a zero at the ESR frequency. A
type III controller, as shown in Figure 19, provides the
necessary compensation.
The first step is to choose the desired bandwidth, f0, of the
compensated system. Choose a frequency high enough to
assure adequate transient performance but not higher than
1/3 of the switching frequency. The type-III compensator has
an extra high-frequency pole, fHF. This pole can be used for
added noise rejection or to assure adequate attenuation at
the error-amplifier high-order pole and zero frequencies. A
good general rule is to choose fHF = 10f0, but it can be
higher if desired. Choosing fHF to be lower than 10f0 can
cause problems with too much phase shift below the system
bandwidth.
In the solutions to the compensation equations, there is a
single degree of freedom. For the solutions presented in
Equation 24, RFB is selected arbitrarily. The remaining
FN6285.1
March 20, 2007
ISL6564A
compensation components are then selected according to
Equation 25.
C  ESR 
R 1 = R FB ----------------------------------------LC – C  ESR 
voltage deviation is less than the allowable maximum.
Neglecting the contribution of inductor current and regulator
response, the output voltage initially deviates by an amount:
di
V   ESL  ----- +  ESR  I
dt
LC – C  ESR 
C 1 = ----------------------------------------R FB
(EQ. 26)
The filter capacitor must have sufficiently low ESL and ESR
so that V < VMAX.
0.75V IN
C 2 = ----------------------------------------------------------------- 2  2 f 0 f HF LCR FB V PP
Most capacitor solutions rely on a mixture of high-frequency
capacitors with relatively low capacitance in combination
with bulk capacitors having high capacitance but limited
high-frequency performance. Minimizing the ESL of the
high-frequency capacitors allows them to support the output
voltage as the current increases. Minimizing the ESR of the
bulk capacitors allows them to supply the increased current
with less output voltage deviation.
2
V PP  2 f 0 f HF LCR FB
 
R C = -------------------------------------------------------------------

0.75 V IN 2f HF LC – 1

0.75V IN 2f
 HF LC – 1
C C = ------------------------------------------------------------------ 2  2 f 0 f HF LCR FB V PP
(EQ. 25)
In Equations 25, L is the per-channel filter inductance
divided by the number of active channels; C is the sum total
of all output capacitors; ESR is the equivalent-series
resistance of the bulk output-filter capacitance; and VPP is
the peak-to-peak sawtooth signal amplitude as described in
Figure 7 and Electrical Specifications.
Output Filter Design
The output inductors and the output capacitor bank together
to form a low-pass filter responsible for smoothing the
pulsating voltage at the phase nodes. The output filter also
must provide the transient energy until the regulator can
respond. Because it has a low bandwidth compared to the
switching frequency, the output filter necessarily limits the
system transient response. The output capacitor must
supply or sink load current while the current in the output
inductors increases or decreases to meet the demand.
In high-speed converters, the output capacitor bank is
usually the most costly (and often the largest) part of the
circuit. Output filter design begins with minimizing the cost of
this part of the circuit. The critical load parameters in
choosing the output capacitors are the maximum size of the
load step, I; the load-current slew rate, di/dt; and the
maximum allowable output-voltage deviation under transient
loading, VMAX. Capacitors are characterized according to
their capacitance, ESR, and ESL (equivalent series
inductance).
At the beginning of the load transient, the output capacitors
supply all of the transient current. The output voltage will
initially deviate by an amount approximated by the voltage
drop across the ESL. As the load current increases, the
voltage drop across the ESR increases linearly until the load
current reaches its final value. The capacitors selected must
have sufficiently low ESL and ESR so that the total output-
24
The ESR of the bulk capacitors also creates the majority of
the output-voltage ripple. As the bulk capacitors sink and
source the inductor ac ripple current (see Interleaving and
Equation 2), a voltage develops across the bulk-capacitor
ESR equal to IC,PP (ESR). Thus, once the output capacitors
are selected, the maximum allowable ripple voltage,
VPP(MAX), determines the lower limit on the inductance.
V – N V

OUT V OUT
 IN
L   ESR  -----------------------------------------------------------f S V IN V PP MAX 
(EQ. 27)
Since the capacitors are supplying a decreasing portion of
the load current while the regulator recovers from the
transient, the capacitor voltage becomes slightly depleted.
The output inductors must be capable of assuming the entire
load current before the output voltage decreases more than
VMAX. This places an upper limit on inductance.
Equation 28 gives the upper limit on L for the cases when
the trailing edge of the current transient causes a greater
output-voltage deviation than the leading edge. Equation 29
addresses the leading edge. Normally, the trailing edge
dictates the selection of L because duty cycles are usually
less than 50%. Nevertheless, both inequalities should be
evaluated, and L should be selected based on the lower of
the two results. In each equation, L is the per-channel
inductance, C is the total output capacitance, and N is the
number of active channels.
2NCVO
L  -------------------- V MAX – I  ESR 
 I  2
 1.25  NC
L  -------------------------- V MAX – I  ESR   V IN – V O


 I  2
(EQ. 28)
(EQ. 29)
FN6285.1
March 20, 2007
ISL6564A
Input Supply Voltage Selection
There are a number of variables to consider when choosing
the switching frequency, as there are considerable effects on
the upper-MOSFET loss calculation. These effects are
outlined in MOSFETs, and they establish the upper limit for
the switching frequency. The lower limit is established by the
requirement for fast transient response and small outputvoltage ripple as outlined in Output Filter Design. Choose the
lowest switching frequency that allows the regulator to meet
the transient-response requirements.
Switching frequency is determined by the selection of the
frequency-setting resistor, RT (see the figures labeled
Typical Application on pages 4, 5, 6 and 7). Figure 20 and
Equation 30 are provided to assist in selecting the correct
value for RT.
RT (k)
1000
100
0.2
0.1
IL,PP = 0
IL,PP = 0.5 IO
IL,PP = 0.75 IO
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VO/VIN)
FIGURE 21. NORMALIZED INPUT-CAPACITOR RMS CURRENT
vs DUTY CYCLE FOR 2-PHASE CONVERTER
0.3
INPUT-CAPACITOR CURRENT (IRMS/IO)
Switching Frequency
INPUT-CAPACITOR CURRENT (IRMS/IO)
0.3
The VCC input of the ISL6564A can be connected either
directly to a +5V supply or through a current limiting resistor
to a +12V supply. An integrated 5.8V shunt regulator
maintains the voltage on the VCC pin when a +12V supply is
used. A 300 resistor is suggested for limiting the current
into the VCC pin to a worst-case maximum of approximately
25mA.
IL,PP = 0
IL,PP = 0.5 IO
IL,PP = 0.25 IO
IL,PP = 0.75 IO
0.2
0.1
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VO/VIN)
FIGURE 22. NORMALIZED INPUT-CAPACITOR RMS CURRENT
vs DUTY CYCLE FOR 3-PHASE CONVERTER
10
10
100
1000
SWITCHING FREQUENCY (kHz)
10000
FIGURE 20. RT vs SWITCHING FREQUENCY
R T = 10
10.886 – 1.0792 log  f S  
(EQ. 30)
Input Capacitor Selection
The input capacitors are responsible for sourcing the ac
component of the input current flowing into the upper
MOSFETs. Their RMS current capacity must be sufficient to
handle the ac component of the current drawn by the upper
MOSFETs which is related to duty cycle and the number of
active phases.
25
For a two phase design, use Figure 21 to determine the
input-capacitor RMS current requirement given the duty
cycle, maximum sustained output current (IO), and the ratio
of the per-phase peak-to-peak inductor current (IL,PP) to IO.
Select a bulk capacitor with a ripple current rating which will
minimize the total number of input capacitors required to
support the RMS current calculated. The voltage rating of
the capacitors should also be at least 1.25 times greater
than the maximum input voltage.
Figures 22 and 23 provide the same input RMS current
information for three and four phase designs respectively.
Use the same approach to selecting the bulk capacitor type
and number as described above.
Low capacitance, high-frequency ceramic capacitors are
needed in addition to the bulk capacitors to suppress leading
and falling edge voltage spikes. The result from the high
current slew rates produced by the upper MOSFETs turn on
FN6285.1
March 20, 2007
ISL6564A
and off. Select low ESL ceramic capacitors and place one as
close as possible to each upper MOSFET drain to minimize
board parasitic impedances and maximize suppression.
INPUT-CAPACITOR CURRENT (IRMS/IO)
0.3
IL,PP = 0
IL,PP = 0.25 IO
IL,PP = 0.5 IO
IL,PP = 0.75 IO
0.1
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VO/VIN)
FIGURE 23. NORMALIZED INPUT-CAPACITOR RMS CURRENT
vs DUTY CYCLE FOR 4-PHASE CONVERTER
MULTIPHASE RMS IMPROVEMENT
Figure 24 is provided as a reference to demonstrate the
dramatic reductions in input-capacitor RMS current upon the
implementation of the multiphase topology. For example,
compare the input RMS current requirements of a two-phase
converter versus that of a single phase. Assume both
converters have a duty cycle of 0.25, maximum sustained
output current of 40A, and a ratio of IL,PP to IO of 0.5. The
single phase converter would require 17.3 Arms current
capacity while the two-phase converter would only require
10.9 Arms. The advantages become even more pronounced
when output current is increased and additional phases are
added to keep the component cost down relative to the
single phase approach.
0.6
INPUT-CAPACITOR CURRENT (IRMS/IO)
The following layout strategies are intended to minimize the
impact of board parasitic impedances on converter
performance and to optimize the heat-dissipating capabilities
of the printed-circuit board. These sections highlight some
important practices which should not be overlooked during the
layout process.
Component Placement
0.2
0
Layout Considerations
Within the allotted implementation area, orient the switching
components first. The switching components are the most
critical because they carry large amounts of energy and tend
to generate high levels of noise. Switching component
placement should take into account power dissipation. Align
the output inductors and MOSFETs such that space between
the components is minimized while creating the PHASE
plane. Place the Intersil MOSFET driver IC as close as
possible to the MOSFETs they control to reduce the parasitic
impedances due to trace length between critical driver input
and output signals. If possible, duplicate the same
placement of these components for each phase.
Next, place the input and output capacitors. Position one
high-frequency ceramic input capacitor next to each upper
MOSFET drain. Place the bulk input capacitors as close to
the upper MOSFET drains as dictated by the component
size and dimensions. Long distances between input
capacitors and MOSFET drains result in too much trace
inductance and a reduction in capacitor performance. Locate
the output capacitors between the inductors and the load,
while keeping them in close proximity to the microprocessor
socket.
The ISL6564A can be placed off to one side or centered
relative to the individual phase switching components.
Routing of sense lines and PWM signals will guide final
placement. Critical small signal components to place close
to the controller include the ISEN resistors, RT resistor,
feedback resistor, and compensation components.
Bypass capacitors for the ISL6564A and ISL66XX driver
bias supplies must be placed next to their respective pins.
Trace parasitic impedances will reduce their effectiveness.
0.4
Plane Allocation and Routing
Dedicate one solid layer, usually a middle layer, for a ground
plane. Make all critical component ground connections with
vias to this plane. Dedicate one additional layer for power
planes; breaking the plane up into smaller islands of
common voltage. Use the remaining layers for signal wiring.
0.2
IL,PP = 0
IL,PP = 0.5 IO
IL,PP = 0.75 IO
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VO/VIN)
FIGURE 24. NORMALIZED INPUT-CAPACITOR RMS
CURRENT vs DUTY CYCLE FOR SINGLE-PHASE
CONVERTER
26
Route phase planes of copper filled polygons on the top and
bottom once the switching component placement is set. Size
the trace width between the driver gate pins and the
MOSFET gates to carry 4A of current. When routing
components in the switching path, use short wide traces to
reduce the associated parasitic impedances.
FN6285.1
March 20, 2007
ISL6564A
Package Outline Drawing
L40.6x6
40 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 3, 10/06
4X 4.5
6.00
36X 0.50
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
40
31
30
1
6.00
4 . 10 ± 0 . 15
21
10
0.15
(4X)
11
20
TOP VIEW
0.10 M C A B
40X 0 . 4 ± 0 . 1
4 0 . 23 +0 . 07 / -0 . 05
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 90 ± 0 . 1
(
C
BASE PLANE
( 5 . 8 TYP )
SEATING PLANE
0.08 C
SIDE VIEW
4 . 10 )
( 36X 0 . 5 )
C
0 . 2 REF
5
( 40X 0 . 23 )
0 . 00 MIN.
0 . 05 MAX.
( 40X 0 . 6 )
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
27
FN6285.1
March 20, 2007
ISL99202
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9001 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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28
FN6285.1
March 20, 2007
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