AN1660

AN1660
A Complete Low-Cost Design and Analysis for Single and Multi-Phase AC
Induction Motors Using an 8-Bit PIC16 Microcontroller
Author:
Justin Bauer
Microchip Technology Inc.
INTRODUCTION
This document provides a detailed analysis of driving a
single and multi-phase AC induction motor (ACIM)
using the PIC16F1509 microcontroller. Laboratory data
is presented alongside the simulated data in an
extensive reasoning of the entire design. The purpose
of this application note is to show how to add speed
control and soft-start to a single and multi-phase AC
induction motor by using a three-phase inverter circuit.
A low-cost solution is delivered, complete with a bill of
materials (BOM), schematic, code and PCB artwork
files. After analyzing this document, the reader can
easily integrate the reference design and associated
materials into a competitive product. The design does
not include power factor correction because of the
low-cost initiative.
The target application of this design is a garage door
opener with a motor size of less than 1 Horsepower
(HP). Motors that are larger than this may require
significant changes to the existing BOM and circuit
layout on the high-voltage side. Hardware
optimizations for different types of motors and input
voltages are detailed.
Here is a short summary of the inverter board benefits
when used to drive a single-phase motor:
• Elimination of relay and run capacitor – a potential
reliability improvement
• Speed/torque control
• No heat sink for up to ½ HP
• Same inverter supports single-phase and
three-phase with one line of code change
• Isolated user interface
• Independent winding drive (load balancing)
The theory on how induction motors work and their
construction will not be covered here. For further
information on this topic, please refer to AN887 in
Reference 2. For a complete listing of Microchip AC
induction motor application notes, please visit
www.microchip.com/motorcontrol/.
The last chapter of this document contains a list of
recommendations that the reader may want to further
investigate.
FIGURE 1:
AC INDUCTION MOTOR
WITH THREE-PHASE
INVERTER BOARD
Laboratory results from running a ¼ HP permanent
split capacitor (PSC) motor directly from the mains at
120V 60 Hz with a run capacitor are directly compared
to the identical motor being driven via a multi-phase
inverter board. A dynamometer and a power analyzer
are used to characterize the reference design. The rest
of this document will analyze these findings along with
other supplementary inverter features such as isolated
I/O, variable frequency (V/f) control, feedback and
voltage regulation. The entire procedure and materials
described in this document will provide the reader with
enough material to design a repeatable experiment.
Three-phase applications can also utilize this board
with minimal software changes; however, this
document does not provide a detailed analysis on the
results. Instructions on the setup and testing procedure
of a three-phase motor are covered in this application
note.
 2014 Microchip Technology Inc.
DS00001660B-page 1
AN1660
TRADITIONAL SINGLE-PHASE AC
INDUCTION MOTOR TOPOLOGY
The PSC single-phase induction motor is often the
simplest and most widely used motor of this type (see
Section “References”). It has low starting torque and
current; however, it incurs inefficiencies from the
expensive, non-polarized run capacitor. This capacitor
commonly fails before the rest of the motor, so there is
a concern for maintenance because of this
dependency.
Topology
A PSC single-phase induction motor is also less
commonly referred to as a two-phase motor since it is
unable to turn without sufficient phase shift between
the two windings. A capacitor can be placed
in-between the input signal and the windings in order to
produce an approximate 90-degree phase shift.
FIGURE 2:
Note:
Losses
A practical capacitor has resistance and radiates heat
as it consumes the RMS AC ripple current within its
equivalent series resistance.
The permanently installed capacitor trades off starting
torque capability at standstill with ripple torque
reduction at running speed. Because of the high VA
rating of the capacitor, capacitors are often selected to
meet the minimum starting performance requirements,
resulting in poor running efficiency.
For motors that do not have identical windings, it is
necessary to feed the two phases with different
voltages. This asymmetry is due to the presence of the
capacitor, which forms a resonant circuit with the
motor’s inductance. Consequently, this raises the
voltage across one of the windings and causes uneven
current flow.
TRADITIONAL
SINGLE-PHASE AC
INDUCTION MOTOR
TOPOLOGY
The capacitor provides the necessary
phase shift to produce a rotating magnetic
field.
The switch is often replaced with a relay that can
control the direction by swapping which phase leads or
lags the other. The value of the capacitor is typically
specified by the motor manufacturer and is sometimes
in the range of 5-50 uF for motors that are less than
1 HP. The capacitor is carefully chosen in an attempt to
correct the power factor for maximum power efficiency.
The voltage rating is typically high at around 220-450V,
depending on the input voltage. The capacitor must not
be polarized, since it is across an alternating voltage. If
this capacitor fails, the motor will cease to turn. The
importance of selecting the correct capacitor is
therefore critical.
DS00001660B-page 2
 2014 Microchip Technology Inc.
AN1660
INVERTER BRIDGE SINGLE-PHASE
AC INDUCTION MOTOR TOPOLOGY
Topology
A three-phase inverter can be used as a substitute to
the permanent capacitor as seen in Figure 3.
FIGURE 3:
SINGLE-PHASE INVERTER WITH THREE HALF-BRIDGES
Note:
Six PWM signals are used to drive the connected squirrel cage PSC motor.
This topology has the benefit of being able to adjust the
speed of the motor and apply the appropriate amount
of voltage on each winding so that the weaker winding
is not overdriven. VDC is created after rectifying the AC
input. See Figure 31 for the entire system overview.
Variable Frequency Drive (VFD) is the control algorithm
used to ramp up and down the motor. For more
information on V/f control, please see Reference 3.
 2014 Microchip Technology Inc.
DS00001660B-page 3
AN1660
EQUATION 3:
Phase Derivation
The motor will still spin without the capacitor if the coils
are driven out of phase from one another. This can be
achieved by creating three phases in software. These
three-phase voltages can then be referenced from one
another to create two resultant waveforms across the
two motor windings. Equation 1 shows how one of the
phases (Vv) is taken as the reference (neutral) to
create two waveforms. The following equations are
explained in more detail in Equations 4 through 6 in
AN967 (see Reference 6), which outline each of the
three phases in the time domain.
EQUATION 1:
MOTOR WINDINGS
EQUATION
V MAIN = V U – V V
V AUX = V W – VV
Note:
INDIVIDUAL PHASE
EXPRESSIONS
V DC
V DC

V U = ------------- SIN  --------- T + ------------ 180 
2
2
V
V
V
V

DC

DC
= ------------- SIN  --------- T – --- + ------------V
 180
2
2
2
W
V DC
V DC

= ------------- SIN  --------- T –  + ------------ 180

2
2
The phase amplitude is equal to the rectified DC
voltage. There is a DC offset of VDC/2 since the sine
wave must swing an equal proportion in both the
positive and negative direction. The PI/2 term indicates
a phase shift of 90 degrees, as seen in Figure 4.
FIGURE 4:
THREE PHASES FROM THE
OUTPUT OF INVERTER
The phasors for each phase are explained
in more detail in Figure 11 in AN967 (see
Reference 6).
VU, VV and VW are the three phases that are created in
software utilizing the PWM technique. Equation 2
shows a general time domain expression of a
sinusoidal wave that will be used to graph and analyze
the voltages on the motor.
EQUATION 2:
A
T
GENERALIZED SINUSOIDAL
EXPRESSION
= A SIN   T    + V
M
Symbol
DC
Description
At
Time-Varying Waveform
Am
Waveform Magnitude
VDC
Direct Current (DC) Offset
Equation 3 shows the individual inverter phases after
applying the general representation of a time varying
sine wave.
DS00001660B-page 4
Note:
VU and VW are each 90 degrees out of
phase from the middle winding, VV.
Phase VV is considered the neutral one. Each winding
has VV as its reference.
 2014 Microchip Technology Inc.
AN1660
Figure 5 shows the two resultant waveforms that are
realized across the two windings of equal magnitude.
FIGURE 5:
FIGURE 6:
ALL-PHASE VOLTAGES
VOLTAGES SEEN ACROSS
THE TWO MOTOR
WINDINGS
Note:
As Figure 5 shows, the peak voltage is much lower
than the original 311V. After substituting values into
Equation 1 from Equation 3 and locating where the
maximum voltage occurs in the auxiliary and main
1  V DC, or
waveforms, the peak voltage evaluates to  -----
2
70% less than that of connecting the motor directly to a
220 VRMS source with capacitor. The two resultant AC
waveforms across the motor’s stator are only utilizing
70% of the available DC supply. Figure 6 shows the
three phases from Equation 3 alongside the two-motor
waveforms from Equation 1.
Please note how the maximum peak
voltage on VMAIN and VSTART never reach
the full potential of the DC bus voltage.
The resultant waveform across the motor does not
utilize 100% of the input DC bus voltage.
Space vector PWM is therefore used to recover this
missing percentage of voltage; however, this strategy
only works for three-phase motors that have each of
their windings 120 degrees out of phase from one
another. Another strategy that can be used is to add the
third harmonic to the sine wave so that close to 100%
of the DC voltage can be utilized. Please see AN955 for
more details on sine PWM generated phases with 120
degree phase shift. The software included with this
inverter board contains a look-up table with the modified values.
Figure 7 shows the current through the windings.
FIGURE 7:
CURRENT THROUGH COILS
USING A CURRENT PROBE
The two current phases in Figure 7 are approximately
90 degrees out of phase from one another.
 2014 Microchip Technology Inc.
DS00001660B-page 5
AN1660
PSC AND INVERTER COMPARISON
FOR SINGLE-PHASE MOTORS
Introduction
This section explains the scientific comparison
between the PSC run method versus the inverter
method with no run capacitor. Table 1 shows a
summary of the tests conducted.
TABLE 1:
SUMMARY OF TESTS
Test
Purpose
Torque
To characterize the effects of Variable Frequency Drive on the motor’s torque.
Acceleration and Speed To identify which method turns the motor shaft the fastest and how quickly the load will
accelerate.
Efficiency
To compare the real component of the output power to the input power and to measure the
power factor and other inefficiencies.
These three tests cover the largest design
considerations in motor control. None of the tests
require a special setup from the other tests. Each
analysis in the three separate tests uses the same
subset of data.
Equipment
Table 2 lists the equipment used during the tests.
TABLE 2:
CURRENT THROUGH COILS USING A CURRENT PROBE
Purpose
Three-Phase Power Analyzer
Model
Yokagawa WT1806
Braking Source (Dynamometer)
Magtrol Hystersis 715-D
Programmable Controller (Dynamometer)
Magtrol DSP6001
PC Control Testing Software (Dynamometer)
Magtrol M-TEST7
1800w AC Power Supply
California Instruments 2001RP
A single motor from WEG was used during the
experiment. Its nameplate readings can be seen in
Table 3.
TABLE 3:
MOTOR NAMEPLATE READINGS
Property
Value
Size
1/4 HP
Type
One-phase ACIM
Poles
4
Frequency
60 Hz
Voltage
220V
Current
1.2A
Run Capacitor
12 uF
Speed
1645 RPM
Main-Winding Resistance
13
Auxiliary-Winding Resistance
13
DS00001660B-page 6
 2014 Microchip Technology Inc.
AN1660
This particular motor has a balanced winding
configuration. Both windings are identical in terms of
resistance and inductance. The results shown in this
document will differ from other motors constructed in
other various ways.
Test Setup
The entire test setup can be seen in Figure 8.
FIGURE 8:
TEST SETUP ON THE DYNAMOMETER
Power
Analyzer
Programmable
Controller
Data Logging
Note:
Power Supply
Motor Board
HD Hysteresis Dynamometer
A Hall effect sensor measures the shaft speed. The programmable controller
applies the test setup from the M-TEST 7 software to the dynamometer and
reads the applied torque. All other readings are performed by the power
analyzer and logged.
The input to the system is single-phase, two-wire
(1P2W) voltage at 220V, 60 Hz. The power analyzer is
also configured for 1P2W on its single element input.
The output from the power analyzer is fed into the two
terminals on the inverter. The inverter has an optional
voltage doubler that is not used during the tests since
the input voltage is greater than 115V. Three wires are
then connected to the motor windings as seen in
Figure 3.
All tests lasted under 60 seconds because of the risk of
damaging the insulation of the motor as a result of large
prolonged currents in the stator.
The PC interface runs the M-TEST software which
configures the programmable controller and hence, the
dynamometer. When a test is running, the
programmable controller adjusts the dynamometer to
apply a specific braking force. Before initiating a test,
the motor is brought up to its maximum free running
speed. When the maximum speed for a specific
frequency is reached, the test begins. The duration of
the test depends on the starting speed of the motor.
TABLE 4:
The M-TEST software runs a dynamic ramp test with
inertia cancellation on the motor. A ramp test
decrements the motor speed in steps of 50 RPM until it
reaches a minimum of 150 RPM. A single locked rotor
reading is extrapolated using 20 data points.
 2014 Microchip Technology Inc.
Note:
Please refer to the motor manufacturer’s
recommendations on the subject of
prolonged stress testing (locked rotor).
The inverter board’s properties for the entire test
duration are listed in Table 4.
INVERTER BOARD
MODULATION PROPERTIES
Property
Value
Switching Frequency
7.82 kHz
Sine Table Look-up Values
64
Dead Band
420 ns max.
When the test is complete, the power analyzer and
motor feedback readings are consolidated into a
spreadsheet and plotted with MATLAB®.
DS00001660B-page 7
AN1660
Results
This section presents the quantitative data taken from
the power analyzer. Table 5 shows a summary of the
tests performed.
TABLE 5:
SUMMARY OF TESTS
Test
Purpose
Torque
To characterize the effects of Variable Frequency Drive on the motor’s torque.
Acceleration and Speed To identify which method turns the motor shaft the fastest and how quickly the load will
accelerate.
Efficiency
To compare the real component of the output power to the input power and to measure the
power factor and other inefficiencies.
These three categories of tests use the same data
points but present different pieces of it.
The main advantage of using an inverter board over the
traditional PSC method is that the user can control the
voltage and drive frequency. Multiple modulation
frequencies are tested on the inverter board to
characterize the effects on the motor, such as
maximum torque, speed and efficiency.
The voltage and frequency can be adjusted with the
inverter board. A 1:1 voltage-to-frequency ratio means
that the voltage scales with the frequency in an exact
linear fashion. The inverter board can control these two
parameters independently of one another to create
unique results. A maximum voltage-to-frequency ratio
implies that the voltage is kept at its maximum whilst
the frequency is adjusted.
Table 6 shows what frequencies are tested and their
maximum torque results when both a linear 1:1
voltage-to-frequency and maximum voltage-tofrequency control algorithms are used for frequencies
lower than 60Hz.
TABLE 6:
INVERTER BOARD TESTS WITH 1:1 AND MAX V/f CURVE
Frequency
(Hz)
Starting Torque
(Nm)
Max. Torque
(Nm)
Max. Speed
(RPM)
30
0.472
0.472
819
30.0
29
1:1 V/f
40
0.664
0.664
1170
29.5
42
1:1 V/f
50
0.765
0.777
1430
34.8
66
1:1 V/f
60
0.742
0.788
1770
37.7
87
Inverter
60
0.830
1.63
1760
47.1
227
PSC method
70
0.670
0.724
2020
39.2
94
—
80
0.53
0.576
2310
37.8
87
—
90
0.375
0.436
2530
33.3
77
—
120
0.195
0.218
3340
28.6
56.6
—
30
2.12
2.49
905
29.6
106
Max. V/f
40
1.61
1.62
1170
35.8
109
Max. V/f
50
1.16
1.16
1490
38.7
105
Max. V/f
In short, the inverter is capable of lifting a larger size
load at start-up; however, its rate of work is much
slower than that of the one-capacitor motor. The rest of
DS00001660B-page 8
Max. Efficiency
(%)
Max. Output
Power (W)
Notes
this section will cover a more detailed analysis of the
data in Table 6.
 2014 Microchip Technology Inc.
AN1660
TORQUE
VARIABLE FREQUENCY DRIVE
The starting torque is one of the most important
characteristics in a motor application. If the motor is
unable to lift the load at 0 RPM (locked rotor), then
none of the other properties matter, since the load will
not move.
The torque developed by the induction motor follows
Equation 4 below.
Figure 9 shows the torque curves plotted against the
speed of the motor’s shaft.
FIGURE 9:
EQUATION 4:
T = K  I
1 M 2
Neglecting the voltage drop caused by the stator
impedance, the magnetizing flux is found to be:
TORQUE CURVES FOR
LINEAR 1:1 V/f
V1
 M = K 2 ------F
1
Where:
Symbol
Description
Units
T
Torque available on the motor
shaft
Nm
Magnetizing Flux (Wb)
Wb
m
I2
Rotor Current (A) ~
proportional to Load
V1
Stator Voltage
V
Constants ~ proportional to
Motor Design
—
K1K2
From the curves, it is apparent that the motor has the
highest starting torque of around 0.75 Nm between
50-60 Hz in line with the motor design. Frequencies
above and below 60 Hz have lower torque profiles. The
V/f slope determines this profile for frequencies below
the rated frequency of 60 Hz. Ideally, a 1:1 ratio
between the voltage and frequency should yield a
constant torque as seen in Figure 10.
FIGURE 10:
V/f RELATIONSHIP TO
TORQUE
A
Given a constant torque load, then, proportionally
varying voltage amplitude and frequency will result in
constant flux and, therefore, constant torque while the
current remains unchanged. For more information,
please refer to the WEG technical guide listed under
Reference 8.
In a real application, the actual voltage/frequency ratio
is usually restricted to a certain range and is not a
perfect 1:1 ratio. One such restriction is the motor
voltage. Overvoltage will damage the insulation of the
stator and cause premature malfunctioning of the
motor.
From the rated frequency upwards, the voltage is kept
constant while the frequency is increased. This causes
the flux to decrease and the motor torque starts to
decrease gradually. For more information please refer
to the WEG technical guide under Reference 8.
However, the low frequency plots in Figure 9 do not
show a constant torque curve. To understand this
phenomenon, the theory on Variable Frequency Drive
must first be discussed.
 2014 Microchip Technology Inc.
Frequencies lower than 60 Hz require a fine tuning of
the voltage-frequency ratio because of losses in the
motor and inaccuracies in the motor drive. The plots in
Figure 9, which do not show a constant torque profile,
are created using a 1:1 ratio. Figure 11 shows the V/f
profile with a slope close to zero by keeping the voltage
at its maximum while adjusting the frequency.
Frequencies below the optimal 60 Hz are kept at
maximum voltage.
DS00001660B-page 9
AN1660
FIGURE 11:
MAXIMUM V/f CONTROL
TORQUE CURVES
If maximum torque is desired, the rate of frequency
increase should be adjusted to jump curves so as to
take advantage of the breakdown torque, as shown in
Figure 13.
FIGURE 13:
It should be apparent from Figure 11 that, when the V/f
slope is lowered from 1:1 to close to 0:1, the torque
rises dramatically as the frequency is decreased. A low
frequency causes a decrease in an inductor’s
impedance. The high voltage applied to this lowered
impedance raises the current in the stator which
produces higher torque.
In order to get a linear torque curve with this specific
motor and inverter board, the V/f profile should be
fine-tuned in code until all frequencies below the
optimal frequency share the same shifted torque
profile. Figure 12 shows a typical plot of torque curves
for a typical NEMA type A motor.
FIGURE 12:
Note:
TYPICAL VARIABLE
FREQUENCY TORQUE
CURVES
CONSTANT TORQUE
PROFILE
The red line shows how the software should control the
frequency and voltage in order to ride along the tops of
each torque curve. This will maximize the delivered
torque at a constant rate.
Figure 14 shows the torque curve for the traditional
PSC method at 60 Hz compared to the 60 Hz inverter
method.
FIGURE 14:
TORQUE CURVES OF THE
CAPACITOR RUN VERSUS
INVERTER RUN
Curves below the optimal 60 Hz have
greater starting torque.
DS00001660B-page 10
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AN1660
The shapes of the curves between the two methods
differ greatly. The capacitor method shows a slightly
larger starting torque and will accelerate faster than the
inverter board at 60 Hz. The inverter board produces a
curve similar to that of a class D design motor, whilst
the PSC run topology produces a similar class A torque
curve. An unequal voltage magnitude caused by the
permanent capacitor creates an unequal magnitude of
magnetizing flux within the stator. The inverter board
attempts to create an equal amount of current in each
winding, since this particular motor has identical
impedance in each. The shape of the torque curves are
not similar because of these discrepancies in the
driving topologies.
Looking at the starting torque when the rotor is locked,
the inverter would be unable to lift the same-sized load
as the PSC method if the inverter were programmed to
simply turn the motor at a 60 Hz modulation frequency.
However, the inverter board can use Variable
Frequency Drive to lift an even larger-sized load as
seen in Figure 11. The designer must also take into
consideration the trade-offs of large starting torque
versus efficiency and speed.
Operating the motor above its rated output torque for a
prolonged period of time will reduce the motor’s
lifespan due to larger-than-rated currents in the stator.
Figure 15 shows a hypothetical motor operation where
the load requires a large starting torque to overcome
stresses such as static friction. This is considered a
hard-start.
FIGURE 15:
HARD-START PROFILE
An important consideration is to determine at what
point to stop incrementing the frequency. If the load
requires a minimum of 0.4 Nm torque to lift, then the
inverter board should be configured to stop its
hard-start at around 80 Hz. As described in Section
“Efficiency”, the full load torque should be matched
with the correct curve, so as to maximize the efficiency
when the motor has been spun up to its running speed.
If the frequency is increased too fast, the motor will stall
and it is unrecoverable until the V/f curve is reset and
adjusted accordingly.
Figure 16 shows a soft-start, which lowers the initial
stress of the motor as it spins up to its running speed.
FIGURE 16:
Note:
SOFT-START PROFILE
Lines in red are a hypothetical course.
Varying the V/f profile should be adjusted according to
each specific application. It may be more practical and
easier to pick one frequency curve and stay on it.
Note:
Lines in red are a hypothetical course.
 2014 Microchip Technology Inc.
DS00001660B-page 11
AN1660
Summary
ACCELERATION AND SPEED
1.
The most obvious benefit of V/f control is that the
designer can control the speed at which the motor shaft
spins. The faster it spins, the sooner the load can be
pushed or pulled to its destination. Getting the gate up
faster than the competition is a critical design win in a
garage door or gating system application.
Starting Torque
The inverter board presents the highest amount of
starting torque, which is arguably the most important
design property in applications where the load is fixed
and requires a large amount of starting torque. The
inverter board must configure the V/f slope accordingly
to achieve these large torque values. A drawback of
this modification is that the stator consumes very large
currents and the maximum speed of the motor will be
slow if the curve is followed to normal run location. The
designer should be aware of the motor specifications
before modifying the applied voltage and frequency.
2.
Nominal Run Speed (60 Hz)
As seen in Figure 14, the PSC method yields greater
starting torque and run-time torque over the motor’s full
range of operation. The PSC method would be a
cheaper and more powerful alternative in terms of
output torque if the inverter board did not take
advantage of manipulating the voltage and/or
frequency and it just ran the motor at its nominal
frequency.
3.
• Angular acceleration  applied torque
• Accelerating torque  available motor torque –
load torque
Large torque will accelerate the motor’s shaft at a faster
pace for a given speed. Figure 17 shows the run
capacitor method at 60 Hz compared to the inverter
board at 80 Hz.
FIGURE 17:
ACCELERATION FROM
TORQUE PROFILES
Torque Range
Settings to accommodate motors that only require low
torque can be easily adjusted with the inverter board.
This typically prolongs the motor’s lifespan, it
conserves energy and displaces heat to its
surroundings, such as a garage. The inverter can
change the frequency and voltage, which is not
possible with the limited on and off control of the PSC.
With the permanent capacitor, the motor must be sized
larger than what is needed and may spend its lifetime
not operating at its fully rated load.
DS00001660B-page 12
The shaded area represents more torque available for
acceleration than the other curve below it. Even though
the 80 Hz curve reaches up to 2300 RPM, the motor will
take longer to reach that speed compared to the
capacitor method. An application where the load must
be moved a large distance would benefit from riding
along the 80 Hz curve since it will be cruising at a
higher speed on average. Short distance applications
may benefit from using the 60 Hz curve with the
capacitor method, since the 80 Hz curve will take a
longer time to ramp up its speed.
 2014 Microchip Technology Inc.
AN1660
Please note that this particular motor has four poles. At
60 Hz, its synchronous speed can be calculated as in
Equation 5.
EQUATION 5:
N
S
120  60
120  F
= ----------------- = -------------------- = 1800 RPM
P
4
Symbol
Description
NS
Synchronous Speed
Applied Power Frequency
Hz
P
Total Number of Poles
—
The synchronous speed of the motor can also be
qualitatively identified by looking at Figure 16 and
locating where the torque curve approaches a value of
‘0’. After synchronous speed is achieved, the
frequency is increased while keeping the voltage
constant. This lowers the available torque on the motor
shaft, but increases the speed at which it turns. To take
advantage of the increase in speed, the designer
should appropriately size the motor and stepping
speed, so as to reach a suitable ending speed. Recall
that if the soft/hard-start increases too rapidly or follows
a curve that cannot continue to lift the load, the motor
will follow the curve back towards (0 RPM) and it will
stall, as shown in Figure 18.
Note:
1.
INCREASING THE
FREQUENCY TOO RAPIDLY
Speed Range
The inverter board allows the motor to out-pace a
replica motor that is driven by the PSC method. The
PSC method can only be driven at one frequency and,
hence, it cannot exceed its synchronous speed.
2.
Units
RPM
ƒ
FIGURE 18:
Summary
Acceleration
The difference between the maximum torque at a
specific speed and what the load actually needs will
determine the rate at which the motor is able to lift the
gate. A large delta will get the gate up faster to a certain
speed.
EFFICIENCY
The system efficiency, as defined in Equation 6, is
analyzed in this section.
EQUATION 6:
PM


 =  ----------------------------------  100
 V RMS  I RMS
Symbol
Description
η
Overall System Efficiency
Units
—
Motor Output
W
Vrms
Input RMS Voltage
V
Irms
Input RMS Current
A
Pm
Note that efficiency is governed by the output power of
the motor shaft divided by the real component input
power to the system. The losses of the motor as well as
the inverter board are lumped together in this model.
Future tests may also want to place a one-phase
three-wire (1P3W) measuring element in-between the
inverter board and the motor to further characterize the
losses. Please see Section “Recommendations” for
more information on other suggestions for future tests.
Lines in red are a hypothetical course.
If the maximum load line is crossed because of the
frequency jumping too soon, the motor will stall by
following its current curve back to 0 RPM (i.e.,
insufficient torque available to drive the load).
Please see Section “Software” to learn more about
the available settings to prevent the motor from stalling.
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Figure 19 shows the efficiency of the PSC and inverter
board at 60 Hz.
FIGURE 19:
EFFICIENCY OF THE PSC
METHOD VERSUS THE
INVERTER
Another important characteristic is the input and output
power of the motor. Figure 21 shows the efficiency
along with the input power.
FIGURE 21:
Note:
The system running at 60 Hz with the PSC method (see
Figure 19) is approximatively 10% more efficient than
with the inverter method.
Figure 20 shows the efficiency and torque plotted on
the same graph.
FIGURE 20:
Note:
EFFICIENCY AND TORQUE
The dotted lines represent torque
curves. Their axis is on the right.
The motor’s peak efficiency is at nearly the same points
in both methods. A typical induction motor will run most
efficiently at a few percentage points from its
synchronous speed. Please note the motor is not
running at its maximum efficiency when the torque is
also at its maximum.
DS00001660B-page 14
EFFICIENCY WITH INPUT
POWER
The dotted lines represent power curves.
Their axis is on the right.
Further analysis shown in Figure 22 shows that much
of the input power is not consumed and it is rather
wasted as reactive power.
FIGURE 22:
POWER FACTOR
More than half of the energy is stored in capacitive and
inductive elements in the inverter board. The PSC
method has a power factor (PF) close to ‘1’ due to the
close matching of the capacitor in relation to the
motor’s inductance. There is also no power stage with
the PSC method to support typical circuitry such as a
microprocessor and sensor inputs. The power factor
comparison is therefore favoring the PSC method due
to its basic, atypical setup.
 2014 Microchip Technology Inc.
AN1660
The poor power factor can be attributed to the
rectification stage of the inverter board’s power supply,
as shown in Figure 23.
FIGURE 23:
MOTOR DRIVEN BY
INVERTER
FULL-WAVE BRIDGE
CURRENT CONSUMPTION
The two large DC capacitors only consume current
when the DC voltage lowers below the peak voltage of
the AC input. Therefore, there are large peaks of
current in short bursts when the incoming voltage
charges the capacitors. The current drawn will have
significant harmonic content due to the presence of the
switching elements.
The motor will also induce reactive power into the input
system which reduces the overall efficiency of the
system. The designer may want to install passive
power factor correction to improve the design.
This causes the voltage and current consumption from
the power utility company to be inefficient. The power
factor correction is not installed because of the low-cost
nature of the inverter. An improved power factor will
greatly increase the real component of the input power.
Infrared imaging was used to compare stator losses in
the motor with the two methods. There is no load
connected to the motor, which is set to run at
60 Hz 220V.
FIGURE 24:
FIGURE 25:
As shown in Figure 24 and Figure 25, under the same
input conditions, the motor driven with the run
capacitor has its stator running warmer compared to
the inverter method. The temperature range on the
bottom indicates an overall increase of the entire
picture. The arrow in the box simply indicates the
warmest part in the box. The designer must take the
current and overall power input/output curves into
consideration before arriving at the conclusion that the
PSC method produces more losses in the stator. An
induction motor fed by PWM voltage presents a lower
efficiency level than when fed by purely sinusoidal
voltage, because of the increased losses caused by
harmonics in the PWM.
Additional analysis into the harmonics produced by the
switching elements on the inverter should be
performed in order to characterize the stator losses.
See Section “Recommendations” for further tests
that can improve the data.
The current in both systems is also analyzed (see
Figure 26 through Figure 28).
FIGURE 26:
TOTAL CURRENT
CONSUMPTION AT 60 HZ
MOTOR WITH PSC
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The amount of current consumed by the overall
system in both methods is similar, but the inverter has
an approximate 500 mA more when the motor starts.
The current in the inverter would theoretically double if
the voltage doubler was used for 115V systems.
FIGURE 29:
OUTPUT POWER AT 60 Hz
For operation under 60 Hz, the current reduces as
dictated by the V/f slope.
FIGURE 27:
TOTAL INVERTER CURRENT
CONSUMPTION AT 1:1 V/f
The inverter has reduced output power capability. The
designer should consider this when the inverter is to be
placed in a direct substitution of a PSC application.
This is primarily caused by the 30% VDC utilization loss,
as shown in Figure 6, as well as by harmonic losses
and a low power factor. The output power does not
increase much at maximum voltage in lower
frequencies.
FIGURE 30:
However, if the voltage is kept at a constant maximum
and the frequency decreases to 30 Hz, the current
increases dramatically, as seen in Figure 28.
FIGURE 28:
OUTPUT POWER OF
INVERTER
TOTAL INVERTER CURRENT
CONSUMPTION AT 0:1 V/f
Note that Figure 29 and Figure 30 use the same
Y axis scale.
Because of the inductive load, a decrease in frequency
lowers its impedance and causes a large increase in
current. The limiting factor in the inverter board is the
ripple current in the DC capacitors. Please see
Table 19 for more information regarding design
limitations.
The mechanical output power of the motor is also
compared to that of the PSC run method, as seen in
Figure 29.
DS00001660B-page 16
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Summary
1.
INVERTER BOARD
Overall Efficiency
The PSC run motor is approximately 10% more
efficient, as seen in Equation 6. Losses in the inverter
run motor can be attributed to the inverter itself and to
the switching elements.
2. Power Factor
The PSC method has an almost unity power factor.
The inverter board has a reduced power factor and,
therefore, it wastes energy in the power grid. Power
factor correction may be added to correct this.
3. Output Power
The PSC run motor has much greater output power at
around 230W. The motor is a 1/4 HP, so the expected
output power is at least 746/4 = 186W. The inverter
board yields slightly under 100W. A drop-in
replacement scenario should note this key difference.
Although the inverter board can lift a heavier gate at
faster speeds, the rate of work is considerably less
when driving a split-phase motor with no run capacitor.
FIGURE 31:
The inverter board is constructed as an
application-specific platform rather than a general
purpose demo board. However, it does provide
numerous inputs and outputs (I/O) for the user to
interact with and modify. The inverter is strictly
engineered to drive a single-phase or three-phase AC
induction motor. Most of the parts in the bill of materials
in Appendix A: “Bill of Materials” have been
optimized to drive up to a ½ HP motor, whilst some
parts, such as the IGBTs, are higher-end and give the
designer more flexibility. After reading this section, the
reader should understand the limitations of this
inverter, as well as its optimizations in software and
hardware.
Overview
Figure 31 shows a top-level overview of the overall
system.
SYSTEM OVERVIEW
The input voltage is first doubled if the jumper is
inserted for 115V systems with a 220V motor.
Note:
Introduction
Please see Table 20 for configuration
settings. Incorrect jumper settings may
cause the board to malfunction.
DC bus voltage. As the motor spins, the microcontroller
monitors the speed and current limit settings, as well as
the feedback current, to ensure overcurrent protection.
It is then rectified to a DC bus voltage. This DC voltage
is then used in a flyback converter to generate two 15V
DC regulated supplies. Each 15V power source is then
down-converted to 5V from a linear regulator. The user
interface uses optocouplers as isolation for safety
issues. The printed circuit board has a clear indicator in
silkscreen that designates what is safe to touch.
PIC16F1509 has six PWM lines driving the
three-phase inverter, which are connected to the motor.
The modulated sine wave is created from the rectified
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Quick Start
The board’s operation should first be validated as seen
in Section “Acceptance Test”. After validation
passes, the user can use the inverter board as
described in Table 7.
TABLE 7:
QUICK-START INPUTS AND OUTPUTS
Element
Purpose
S0
Start/Stop Motor
S1
Start/Stop Motor
POT0
Controls the speed and direction of the motor
POT1
Current Limiter of the motor
W1
Jumper to select between 115V or 230V input. Jumper should be inserted for 220V motors on
115V input (see Table 20)
P3
Input Single-Phase AC Mains Voltage
P4
Motor Output Terminals
P2
Auxiliary Output
P5
Auxiliary Inputs
F1
Motor Fuse
P1
In Circuit Serial Programming™ (ICSP™) Header
The following basic steps show how to get an induction
motor spinning:
1.
2.
3.
4.
5.
6.
7.
8.
9.
10.
11.
12.
13.
14.
Note the size and type of motor to be attached
to the board.
Program either the single-phase hex file or the
three-phase hex file onto the board by using the
header at P1. 5V must be supplied by the
PICkit™ 2 during programming.
Remove the PICkit 2 from the ICSP™ header.
Place a fuse into the fuse holder at F1 that can
pass the rated motor current.
Connect the three motor wires to terminal P4.
Refer to Figure 3 for single-phase motor
connections.
Connect your input voltage of either 115V or
230V to terminal P3. Refer to Table 20 for
jumper settings.
The PWR green LED should now be on.
Place the wiper in POT1 to its halfway point.
Place the wiper in POT0 fully clockwise.
Press either S0 and S1 to start the motor.
The motor should slowly ramp up to its
maximum speed at 120 Hz modulation.
Slowly turn POT0 counterclockwise to slow the
motor to a halt. Turning it fully counterclockwise
will turn the motor in the opposite direction.
Adjusting POT1 will trigger an automatic
shutdown if the trip point threshold is met. The
trip LED will illuminate when this happens.
The user must press S0 or S1 to start up the
motor again after an emergency shutdown
event.
DS00001660B-page 18
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Feature Summary
The inverter board was designed with flexibility in mind;
therefore, some of the features listed in Table 8 can be
omitted to optimize performance with cost.
TABLE 8:
DESIGN SPECIFICATIONS
Design Specifications
Motor Type
Single-phase and three-phase 1/3HP AC induction motors
AC Input Voltage
115 230V, single-phase
Control Method
V/f, with soft-start/stop, current feedback;
Space Vector Modulation (SVM) for three-phase motors
Motor Rated Voltage
120V or 220V
PWM Switching Frequency
3.9 kHz  15 kHz
Motor Frequency Range
15 Hz  120 Hz
Heat Sink
No
Isolation
Yes
Overvoltage Protection
Yes
Undervoltage Protection
Yes
User Configurable Speed
Yes
User Configurable Current Trip Yes
Most of the inputs/ outputs (I/O) are used by the default
code since an optimal microcontroller was selected.
There is still plenty of space for the developer to use
custom modifications. The I2C™ lines are also free to
use for any added slave devices. Table 9 shows how
much code space and how many modules are used to
meet the design specifications.
TABLE 9:
RESOURCES USED
Resource Type
Used
Available to User
2833 (words)
5359 (words – 65.4%)
Data Memory
289
223 (bytes – 43.6%)
CPU Processing (FPWM = 7.8 kHz, FOSC = 16 MHz)
70%
30%
PWM Channels
3
1
CLC Modules
3
1
ADC Channels
2
6
Timers
2
3
Program Memory
Comparators
0
2
I2C™/SPI
0
1
External Interrupts
4
6
I/O Lines
16
1
EEPROM
0
0
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Microcontroller
Figure 32 shows the pin designators and their
purposes, with the exception of programming
functionality.
FIGURE 32:
PIC16(L)F1508/9 PIN
DESIGNATORS
VDD
VSS
2
19
RA0/ ICSPDAT
RA4
3
18
RA1/ICSPCLK
MCLR/VPP/RA3
4
17
RA2
RC5
5
16
RC0
RC4
6
15
RC1
RC3
7
14
RC2
RC6
8
13
RB4
RC7
9
12
RB5
RB7
10
11
RB6
PIC16LF1508/9
20
PIC16F1508/9
1
RA5
Table 10 shows a complete listing of pin functionality.
TABLE 10:
PIN FUNCTIONALITY
Pin
Input/ Output
Digital/ Analog/ Both
RA5
OUT
Digital
RA4
IN
Both
RA3
IN
Digital
Purpose
Notes
PWR LED
Push Buttons
2 x Shared In
Dedicated Input
Dedicated In
RC5
OUT
Digital
Motor PWM
RC4
OUT
Digital
Dedicated Output
RC3
OUT
Digital
Motor
RC6
IN
Analog
Current Sense
RC7
OUT
Digital
Motor LED
RB7
OUT
Digital
Motor
RB6
OUT
Digital
I2C™ Clock
Unused
RB5
IN
Both
Auxiliary Inputs
3 x Shared In
2C™
Data
RB4
Both
Digital
I
RC2
IN
Analog
Motor Speed
RC1
OUT
Digital
Motor PWM
RC0
OUT
Digital
Motor PWM
Unused
RA2
OUT
Digital
Motor PWM
RA1
Unused
Analog
Unused
RA0
IN
Analog
Motor Current Limit
DS00001660B-page 20
Unused
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AN1660
EQUATION 7:
User Interface
The inverter board has numerous inputs and outputs.
Some of these are multiplexed onto a single pin in
order to accommodate a larger number of I/O. All of the
user interface requirements are isolated via two
four-channel optocouplers and a one-channel
optocoupler. Table 11 lists the user interface inputs and
outputs.
TABLE 11:
USER INTERFACE INPUTS AND
OUTPUTS
V
Multiplexed
Purpose
2
Y
Push Buttons
1
N
Speed Control
1
N
Current Limit Control
1
N
Dedicated Digital Output
1
N
Dedicated Digital Input
3
Y
Auxiliary Digital inputs
The board provides two switch buttons and two
potentiometers. There are also headers that provide
connections for external I/O, such as garage door trip
sensors. The two potentiometers have their transistor
in the optocoupler circuitry biased in its amplifying
region. The output is therefore approximately linear,
since the optocoupler LED does not have a linear I-V
curve. Large currents in the 30 mA range are
consumed for each POT.
The digital push buttons and auxiliary inputs are biased
to cause an interrupt-on-change (IOC) when either of
them is used. This alleviates the CPU from constantly
checking the voltage level on the pins. Whenever an
IOC is detected, an ADC reading must be taken in
order to determine which input caused the interrupt.
FIGURE 33:
MULTIPLEXED PUSH
BUTTONS CIRCUIT
10 K
= 5V  ------------------------------ = 4.8V
 10 K + 390 
When switch 1 (SW1) is pressed, the output voltage is
3.3V, as seen in Equation 8.
EQUATION 8:
V
Isolated User Interface
Qty.
OUT
S0 IS ACTIVE
OUT
S1 IS ACTIVE
10 K
= 5V  -------------------------------- = 3.3V
 10 K + 5.1 K 
The large voltage difference between these two
voltages enables the ADC to distinguish between
which switch is active. The electrical specifications for
the PIC16F1509 and all of the enhanced mid-range
devices specify a maximum input low voltage of 0.8V
and a minimum high voltage of 2.0V, as seen in the
design parameter number 41 (D041) under the
Electrical Specifications chapter (see DS40001609).
This means that for IOC to work correctly, an input
should be higher than 2V when active and lower than
1V when inactive.
SOFTWARE
The code has numerous comments surrounding all
functions and important properties to help the reader
easily modify the operation. Some of the code used in
this application is taken from Microchip AN984 (see
Reference 7). The modulation routines in particular are
based around this reference. Parts of the text will be
replicated throughout this chapter for convenience.
Table 12 lists the important developer information of
the software.
TABLE 12:
BUILD INFORMATION
Property
Description
Language
ANSI C89 C
Compiler
XC8 V1.22 PRO
IDE
MPLABX v1.80
Figure 33 shows an example circuit that has two
multiplexed signals (S0 and S1) on a single input pin to
the microcontroller. The 390 and 5.1K resistors form a
voltage divider with the 10K resistor. When switch 0
(SW0) is pressed, it is considered active and the output
voltage reads close to 5V, as seen in Equation 7.
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Summary
Table 13 lists the states in the main state machine
inside of main.c. The main loop consists of a simple
state machine.
TABLE 13:
STATE-MACHINE STATES
State
Description
INIT
The inverter is initialized and the motor is stopped.
IDLE
Motor is off while polling for button presses.
MOTOR_STARTING
Enter here while performing soft-start.
MOTOR_ON
After the motor has been soft-started, the program enters this state. This state will
continuously poll all auxiliary inputs, switches and speed/current potentiometers.
MOTOR_STOPPING
Enter here when program has been instructed to stop the motor. Soft-stop is performed
gradually here until the motor has been stopped completely, or it brakes.
Upon entry, the motor starts in its IDLE state where the
microcontroller initializes pins and stops the motor. If
SW1 is pressed, the motor starts and transitions into its
MOTOR_STARTING state. The motor starts using the
soft-start method where the frequency and voltage are
adjusted in a linear fashion in order to slowly bring the
motor up to operating speed.
The state machine then transitions into the MOTOR_ON
state when soft-start is complete. The motor speed and
current trip points are continuously polled in the main
loop. If an overcurrent scenario is detected, the
MOTOR_STOPPING state is entered and the motor is
stopped and status LEDs blinked in a pattern to
designate that a Fault condition has occurred. If SW1 is
pressed, the motor is stopped by either braking or by
soft-stop. The state machine returns to state IDLE.
DS00001660B-page 22
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A structure diagram of the important files and their
methods and properties are shown in Figure 34.
FIGURE 34:
STRUCTURE DIAGRAM
main.c
modulate.c
It consists of state machine and global ISR flags. The
PWM routine is given priority. No blocking code other
than this routine is within the ISR. All other interrupts
are serviced outside of the interrupt vector to give
precedence to the modulation routines.
It calculates the adjusted sine wave when given a Volt
and phase parameter and it saves all 64 values in a
pre-filled buffer. The soft-start routine pace of
increment is limited by how fast this routine can
complete, since the voltage and phase parameter often
change. When the motor is steadily running, the buffer
will not change often, since the two inputs are typically
constant.
demo_board.c
This file acts as a facade to the main.c file. A facade
is a software design pattern that provides a unified
interface to multiple complex subsystems. In essence,
it alleviates the complex logic from the main.c file. The
majority of the methods called from the main loop to
this file are abstracted away to other files.
pwm.c
This does the actual loading of the PWM registers from
the phase structure. The CLC is configured here to
generate three complementary PWMs.
bsp.h
soft_start_stop.c
The board specific header file contains all hardware
definitions. This is referenced in most other files.
This file contains math-intensive operation methods.
Before doing a soft-start/stop, the speed POT position
is measured. This position divided by the number of
steps for any given sequence, SOFT_START_DURATION, will be added every SOFT_START_DELAY.
Figure 35 shows a flowchart for the while(1) loop in
main.c.
common.h
This file is the only file that should be modified to
fine-tune the general settings, such as switching
frequency, timer intervals, trip points, etc. Any other
modifications to the system may require adjustments
elsewhere. This file is included in every source file.
generio_io.c
This file handles all user interface components, such as
buttons, switches, auxiliary I/O and potentiometers.
The ADC files are also referenced here for multiplexing
purposes.
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FIGURE 35:
MAIN ROUTINE FLOW DIAGRAM
Start
Main Loop
InitiĂlŝze Motor and
board parameters
Stop the Motor
Idle
No
Is SW1 or SW2
pressed?
Yes
Init͘ SoftStart Motor
Calculate
Modulation Buffer
No
Is SoftStart
finished?
Yes
Yes
Measure the speed and
ŽǀĞƌcurrent POTs
Has the set
speed changed?
Yes
Calculate the
Modulation Buffer
No
Is there an
ŽǀĞƌcurrent
condition?
No
Is SWϭ or SW2
pressed?
No
Any auxiliary
interrupts?
No
Yes
Take an ADC reading
on the multiplexed
pin and service it
DS00001660B-page 24
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AN1660
With the exception of the time-keeping TMR1 overflow
and modulation TMR2 overflow, all other ISR flags are
serviced outside of the ISR vector. Figure 36 shows a
flowchart of the ISR.
FIGURE 36:
ISR FLOWCHART
Interrupt Service Routine
ISR entry
Has TMR2 rolled
over?
Yes
Write modulation
values into PWM
registers
Yes
Set the associated
global flag to be
processed later in
main line code
Yes
Increment the
Uime-based
variables
Calculate phase
values from precalculated
modulation buffer
No
Has an IOC
occurred?
No
Has TMR1 rolled
over?
No
Clear all interrupt
flags that have
occured
ISR Exit
The primary focus of the software is driving the
three-phase inverter. The PWM and CLC hardware
modules are utilized to complete this task. The majority
of the software processing power is spent calculating
the next PWM values and polling for user inputs, such
as trip sensors and speed control. Figure 37 shows the
general overview of the driving stage.
FIGURE 37:
SOFTWARE PWM DIAGRAM
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Software Modulation
Generating the Sinusoidal Waveform
This section describes the modulation technique and
its associated routines. These sections are the primary
focus of the code.
The sinusoidal PWM (SPWM) look-up table method is
the easiest way to generate a sine wave. The
calculations are done in an Excel sheet and then
pasted into the static program memory on the
microcontroller. The sine values are read from the table
at predefined intervals to create a specific modulation
frequency.
MODULATION VARIABLES
A global structure, mod, is kept in RAM, which
specifies the pulse width and offset of the PWM
modules. Table 14 lists the three modulation structure
members.
TABLE 14:
Member
phase
MODULATION STRUCTURE
MEMBERS
Size
Purpose
uint16_t Present Phase Offset
deltaPhase uint16_t Phase Increment
(Frequency)
voltage
uint8_t
Pulse Width (Voltage)
Phase
This value represents a full 360 degrees of angle,
where 0x0000 = 0 degrees and 0xFFFF = 359.9
degrees.
deltaPhase
A Sine Pointer variable, deltaPhase, can be adjusted
to scale how fast or slow the modulation moves through
the look-up table. deltaPhase is then added to the
Phase variable. If the variable Phase is kept
increasing/decreasing at a constant rate, then the
resulting sine wave modulation frequency will be kept
constant. This variable can therefore be left to overflow.
Each time a new value is needed from the look-up
table, the upper eight bits of the pointer variable are
used as the pointer index. The lower eight bits of the
pointer variable can be viewed as fractional bits.
To set the modulation frequency with a PWM frequency
of 16 kHz, please see Equation 9.
EQUATION 9:
F
F
PWM
HZ
= --------------- = 0.244 -------RESOLUTION
BIT
16
2
The deltaPhase value is adjusted accordingly to the
voltage parameter. The voltage parameter is derived
directly from the ADC reading of the speed POT. This
variable is added to the frequency variable, phase
value, every TMR2 overflow to get an adjusted
modulation angle value.
Essentially, every increment of deltaPhase will adjust
the modulation frequency by 0.244 Hz. Changing the
PWM frequency will result in a different resolution.
Voltage
EQUATION 10:
The voltage is modified by the ADC reading of POT0.
It has a maximum value of 127 and a minimum value
of 0. A value of 127 indicates that the PWM signal will
have a duty cycle close to 100% when the sine wave
is at its peak value. The voltage sampling rate can be
modified in the SAMPLE_POT_RATE as seen in
Table 14. The 64-byte sine wave table is then
multiplied by this amplitude factor and added to 50%
duty cycle. These scaled values are saved in a
64-byte array in RAM for quick access in the ISR. It
should be noted that the V/f processing is only
executed when the motor voltage is changed such as
in soft-start.
Checks are in place to ensure that the voltage does
not go below the pre-configured minimum value.
DS00001660B-page 26
To find the Table Pointer delta value that will provide a
60 Hz modulation frequency, use the formula in
Equation 10.
DELTA P HASE
F
MOD
60
= -------------- = ------------- = 246 BITS
0.244
0.244
If value 246 is added to delta at each PWM interrupt,
then the resultant modulation will be 60 Hz.
To generate multiple-phase outputs, simply add a
constant offset value to the Phase Pointer. The source
of this method is AN984 listed under Reference 7. For
more information, please refer to pages 5, 6 and 9 of
the above-mentioned application note.
 2014 Microchip Technology Inc.
AN1660
V/f PROFILE
System Configuration Settings
The two subroutines, CalcAmplitude and
CalcPhase, calculate the two parameters that
generate the values for the PWM signals. CalcPhase
takes the scaled POT0 reading as an input and returns
a value between 0 and 512.
The inverter board code contains numerous settings
that are easily changeable by modifying a single
#define. The entire configuration is within a
common.h file. Other #defines, which depend on the
inputs from the common.h file, are located within the
project. These #defines should be modified with
extreme care.
The amplitude calculation checks the minimum and
ideal frequencies to ensure that the voltage is held
within the correct zone (constant voltage past ideal
frequency).
PWM MODULATION
This is called every TMR2 overflow. The angle
parameter has only its upper six bits used as the
pointer index into the pre-calculated and scaled sine
table in RAM. Only the upper six bits are used since the
total size of the table is 64 bytes (26 = 64). The returned
value from the sine table is assigned to the first PWM
pointer variable. The same routine is followed to get the
second and third phase outputs after the angle
parameter is incremented according to the predefined
phase offset. A value of 0x4000 between modulation
outputs will generate a 90-degree phase shift.
LoadModulateBuffer is called to calculate the
scaled PWM values. It takes a single parameter,
volts, with which to multiply the default sine table. It
then adds HALF_DUTY.
DRIVING FREQUENCY
GENERAL
Table 15 lists the settings associated with the general
operation.
TABLE 15:
Name
Description
DEBUG
If this is commented out, it will
disable CLC4 from outputting
PWM4 on Aux4 (P2). This is
used primarily as one of the
tests in the acceptance
document (see Section
“Acceptance Test”).
TMR1_TIMER_SET
TMR1 is used as a generic
timer and multiple items rely
on this as a heartbeat.
SAMPLE_POT_RATE
The frequency at which
Speed and Current Pinch
POTs are sampled as a
function of
TMR1_TIMER_SET (ms)
_XTAL_FREQ
The microcontroller
operational frequency.
Changing this will require vast
changes to the calculations of
the PWM frequency and other
dependences.
An article published by WEG, a major motor
manufacturer (see Reference 7), points out that there
is no simple interrelation between the insulation life and
the switching frequency. General experiences have
shown the following pattern:
• If fS < 5 kHz the probability of insulation failure
occurrence is directly proportional to the switching
frequency.
• If fS > 5 kHz the probability of insulation failure
occurrence is quadratically proportional to the
switching frequency.
GENERAL CONFIGURATION
SETTINGS
High-switching frequencies can also cause bearing
damages. However, a switching frequency increase
results in the motor voltage FFT improvement and,
therefore, it tends to improve the motor thermal
performance besides reducing noise.
 2014 Microchip Technology Inc.
DS00001660B-page 27
AN1660
MODULATION
The modulation routine has a few associated system
settings (see Table 16). The soft-start/stop routine
lengths and delays, as well as the modulation
frequency, can be modified here. The default code
already contains three predefined frequencies of:
1.
2.
3.
3.9 kHz
7.8 kHz
15.6 kHz
Different frequencies can be configured, but changes
to the modulation routines and sine wave values will
need to be made. The frequencies above were
selectively chosen so as to minimize the overhead in
calculations by using powers of 2 (shifts only) for
multiplication and division.
TABLE 16:
MODULATION-RELATED SETTINGS
Name
Description
FREQ_XXXX
Predefined PWM switching frequencies
SOFT_START_DURATION
Amount of times Timer1 rolls over before incrementing soft-start V/f profile
SOFT_START_DELAY
TMR1_TIMER_SET (ms) per each count
SOFT_STOP_DURATION
Amount of times Timer1 rolls over before decrementing soft-stop V/f profile
SOFT_START_DELAY
TMR1_TIMER_SET (ms) per each count
MODULATION_MAX
The maximum modulation frequency
MODULATION_IDEAL
The ideal modulation frequency for the motor
MODULATION_MIN
The minimum modulation frequency
USER INPUT/ OUTPUTS
The ADC is used on multiplexed pins to distinguish
which I/O has been activated. An isolated user input is
labeled as being active when the input on the isolated
side is driven high at typically 5V. Table 17 shows the
relative user configurable #defines.
TABLE 17:
SETTINGS FOR THE USER I/O
Name
Description
BOARD_V
Microcontroller voltage on VDD
S1_V
The expected voltage on pin RA4 when S1 is pressed
S2_V
The expected voltage on pin RA4 when S2 is pressed
AUX2_V
The expected voltage on pin RB5 when Aux2 is active
AUX3_V
The expected voltage on pin RB5 when Aux3 is active
AUX4_V
The expected voltage on pin RB5 when Aux4 is active
ADC_STOP_POS
The upper limit for Idle operation
ADC_STOP_NEG
The lower limit for Idle operation
DS00001660B-page 28
 2014 Microchip Technology Inc.
AN1660
The speed reading returns a signed integer that is then
compared to the ADC_STOP_NEG/POS #defines.
Rotating the POT counter clockwise from the fully
clockwise position will decrease the motor from full
speed to idle and then full speed in reverse, as seen in
Figure 38.
FIGURE 38:
ADC TO MOTOR READINGS
ADC_STOP_NEG
-512
Reverse
Note:
ADC_STOP_POS
Idle
+512
Forward
The ADC value is printed above.
The biasing of S1 and S2, as well as all of the auxiliary
inputs, is designed to allow the greatest voltage
differential between the shared active elements. This is
achieved by carefully-selected resistors in a voltage
divider network.
MOTOR FEEDBACK
The only feedback to the system is the small RC
feedback network that is connected in between the
motor and GND. This network provides a small voltage
reading that is proportional to the current in the motor,
as seen in Table 18.
TABLE 18:
MOTOR FEEDBACK SETTINGS
Name
Description
CURRENT_TRIP_V
In Volts, when generating a Fault condition. Use this setting if the user wishes
to reallocate the current trip POT to some other usage.
CURRENT_TRIP_AVG
Number of ADC measurements to average for the current trip. Make this a
multiple of 2 for code speed.
 2014 Microchip Technology Inc.
DS00001660B-page 29
AN1660
HARDWARE DESIGN
CONSIDERATIONS
This inverter is constructed to run up to a ½ HP motor;
however, components that can exceed those levels
are used. A low-cost solution is one that selects the
minimum amount of parts with as little performance
overhead as possible to run the application. Table 19
highlights some components that limit the flexibility of
the inverter board.
TABLE 19:
HARDWARE COMPONENT LIMITATIONS
Component
Diode Bridge
Limit Description
Maximum Input Current
Approximate Limit Value
10A
Derivation
Data sheet
DC Ripple Capacitors
Maximum Input Current
2.5 Arms through each capacitor Data sheet
Flyback Converter
Input Voltage
250VAC
Data sheet
Flyback Transformer
Secondary Output Current
200 mA
Data sheet
Feedback Resistor
Motor Current
4.5A
 IR  0.22  5W
Various Terminal Connectors Input/Output Voltage on
each terminal
Variable
Data sheet
IGBTs
Fswitch < 25 kHz and
Idrain < 5A without a heat sink
See Section “Motor
Feedback (Current
Sense)”
Switching Frequency and
Motor Current
2
* The derivation source comes from the component’s respective data sheet.
Figure 39 shows a thermal image of the inverter when
running the motor at 220V 60 Hz with no load.
FIGURE 39:
INVERTER BOARD DRIVING
MOTOR WITH NO LOAD
DS00001660B-page 30
Notice how the IGBTs and NTC are the hottest
components on the board. The IGBTs are rated up to
150°C and the NTC up to 200°C. The other parts that
are warm include the linear regulators, flyback
converter and diode bridge. This design ensures that
under full load, none of the components will be
performing out of specification. The designer may wish
to optimize some of the parts for his/her application,
such as the IGBTs and diode bridge. This section will
explore design limitations and optimizations that can be
made.
 2014 Microchip Technology Inc.
AN1660
DC Bus Voltage Capacitors
The default DC bus capacitors have a capacitance of
560 uF and a ripple current rating of 0.235A. Ripple
current is the amount of RMS AC current flowing
through the capacitor’s plates as seen in Figure 40.
FIGURE 40:
The voltage doubler consists of two capacitors that can
optionally be configured by a single jumper (W1) to
double the input voltage, as seen in Figure 41.
FIGURE 41:
VOLTAGE DOUBLER
RIPPLE CURRENT
The ripple current rating is a direct correlation with its
Equivalent Series Resistance (ESR). Current passing
through the capacitor will cause a voltage drop across
its internal resistance which then causes the capacitor
to heat up. Too much heat dissipation will prematurely
damage the capacitor.
The amount of ripple voltage on each of the capacitors
should be carefully monitored when the motor size is
increased and the V/f profile is modified.
Table 20 shows when to connect to the jumper.
TABLE 20:
JUMPER SETTINGS
Input Voltage
(V)
Motor Rated
Voltage (V)
9
8
115
230
115
115
8
230
230
Jumper In (W1)
The ripple current is related to the ripple voltage. Given
the input frequency, capacitor bank size and load
current, Equation 11 outlines the magnitude of ripple
voltage.
WITHOUT VOLTAGE DOUBLER
EQUATION 11:
EQUATION 12:
VOLTAGE RIPPLE ACROSS
CAPACITORS
When the voltage doubler is not engaged, the effective
capacitance of the two large capacitors in series
creates a total capacitance as shown in Equation 12.
DV
V
ICAP = C ------- = C -------- = C  V  F
DT
T
1
1
1
-------------- = ---------- + ---------C TOT
C 12 C 13
Rearranging:
Symbol
Description
Units
V
Ripple Voltage which is equal
to the difference between
VACmax –VACmin
V
Total load current as seen by
the capacitor
A
ƒ
Input frequency as seen by
the capacitor
Hz
C
Total capacitance of the bank
F
A ¼ HP motor that was used throughout the tests will
consume approximately 1.4A at 60 Hz as seen in
Figure 26. The input frequency, or charge/discharge
rate, for the capacitors and total capacitance bank
varies depending on whether the voltage doubler
jumper is inserted.
 2014 Microchip Technology Inc.
C12 and C13 are the two large
capacitors. Please refer to Appendix B:
“Complete Inverter Board Schematic”
for the schematic.
Note:
I LOAD
V = ---------------FC
Iload
TOTAL CAPACITANCE
WITHOUT DOUBLER
Since the capacitors are of the same value, the total
capacitance is (560 uF/2), or 280 uF.
The output frequency from the full-wave rectifier is
double the input frequency, as shown in Figure 23.
Equation 13 shows the voltage across both ends of the
capacitors combined will equal the peak voltage of the
input.
EQUATION 13:
V
DC
= V
VOLTAGE ACROSS BOTH
CAPACITORS IN SERIES
12
+ V13 
= V
IN
 2
DS00001660B-page 31
AN1660
Given the above calculations at 220V input, the ripple
voltage is as shown in Equation 14.
EQUATION 14:
RIPPLE VOLTAGE
1.4A
V = ---------------------------------------  42V  13.5% 
120H Z  280F
WITH VOLTAGE DOUBLER
Driver Stage
The driving stage uses an International Rectifier
three-phase gate driver for high-voltage applications. It
has integrated dead time and drives six external,
N type IGBTs/FETs in bootstrap operation. The inverter
was designed and tested with IGBTs; however, FETs
can easily be placed on the same footprint. The default
IGBTs are rated up to 10A on their drain with a very low
Vce saturation.
If the W1 jumper is inserted, then the voltage across
both ends of the capacitors will be as seen in
Equation 15.
The design consideration between selecting FETs or
IGBTs is usually a function of switching frequency and
current. Table 21 compares a design decision between
placing an FET or an IGBT down.
EQUATION 15:
TABLE 21:
V
DC
= V
12
CAPACITOR VOLTAGE
WITH VOLTAGE DOUBLER
–V
13

=  V IN  2  –  –  V IN  2  
= V
IN
 2  2
The DC voltage is effectively double that of the input
voltage, but at a cost of increased ripple voltage and
current.
Note:
There is no protection for placing input
voltage higher than120 and placing W1 on
the jumper. If this happens, the capacitors
are at risk of irreversible damage since
their voltage rating will be exceeded.
Each capacitor will only see one half of the sine wave
cycle. Therefore, the ripple voltage for the voltage
doubler scenario will have a frequency that reflects the
input frequency on a 1:1 ratio (see Equation 16).
EQUATION 16:
RIPPLE VOLTAGE WITH
VOLTAGE DOUBLER
Type
IGBT AND FET COMPARISON
Part #
Imax
IGBT
IRG4BC20KD-SPbF 16A
FET
IRFIB7N50APBF
RDS(ON)/
VCE(ON)
NA/ 2.27V
6.6A 0.520 Ohm/ NA
IGBTs are inherently more expensive, so the design for
a ½ HP motor should use FETs with a low Rds_on.
From the IR IGBT’s data sheet, Thermal Resistance
(Junction to ambient) is 40°C/W for a TO-263 package,
and TJ is limited to 150°C. The IGBT in this particular
D2Pak can, therefore, dissipate up to 3.15W before it
requires a heat sink for an ambient temperature of
25°C.
Hence, without a heat sink, the IGBT can only pass
1.38A to the emitter through its collector
 3.15W = 2.27  1.38  . The FET can handle 2.46A
2
without needing a heat sink 3.15W = 0.520  2.46  . It is
critical that the designer should note that the data sheet
specification of maximum current does not imply that
the device can handle that amount without a heat sink
or in a warm ambient environment. Figure 42 shows
that the losses for an IGBT scale linearly, whilst the
FET scales at a power of two.
FIGURE 42:
Assuming identical capacitors at 120V input:
IGBT VERSUS FET IN
CURRENT AND POWER
CONSUMPTION
1.4
V = ----------------------------  84V  27% 
60  280F
With the voltage doubler, the ripple voltage is doubled.
Note:
Keep in mind of the typical 20% tolerance
value on the capacitance when doing
calculations.
IGBT
Power (W)
FET
Current (A)
DS00001660B-page 32
 2014 Microchip Technology Inc.
AN1660
Lower current applications will benefit from using FETs,
while higher current applications with larger motors
should use IGBTs.
Figure 43, from an IR application note, AN980 (see
Reference 5), shows that the cost decision point for
choosing IGBTs over FETs coincides with a motor of
1HP.
FIGURE 43:
PER UNIT DOLLARS
VERSUS MOTOR SIZING
Do not replace the fast-acting with general purpose
diodes. The diodes must be fast-acting in order to
charge the bootstrap capacitors efficiently without
losing leakage current from fast-switching cycles.
Inrush Current
An inrush limiter is used to limit the initial current that is
drawn from the AC supply from tripping a household
circuit breaker. This negative temperature coefficient
(NTC) thermistor suppresses the high inrush current
surges that occur when charging the low-impedance
DC smoothing capacitors of C12 and C13. Once the
capacitors are energized, the resistance of the NTC will
decrease rapidly to a very low value.
A few items of data are needed to scale an inrush
current limiter:
1.
2.
3.
Motors that are less than 1 HP in size should use FETs,
according to the International Rectifier application note.
The designer should determine the trade offs of the two
elements.
Essentially, there is a trade off between the cost of the
silicon and the aluminum needed for the heat sink.
Cheaper FETs/IGBTs need more dissipation caused by
higher losses versus expensive parts that need no
external dissipation. As pointed out in the IR application
note, “Often it is more cost-effective to choose silicon
rather than aluminum.”
4.
Load capacitance of device to be protected
Steady-state current (IMAX) and maximum
ambient temperature
Required reduction of inrush current to
determine R25 of NTC inrush current
limiters
Maximum supply voltage
For further details, please see Epcos application note
listed under Reference 9.
The Cantherm MF72 data sheet, Power NTS
Thermistor of the selected NTC, MF72-22D11, already
provides these data points. Important characteristics of
the NTC are placed in Table 22.
SMD thermal dissipation is an alternative route to using
a heat sink, albeit at a cost of heating up other
components on the inverter.
Besides the switching elements, another design issue
is the selection of gate resistors and bootstrap diodes
and capacitors.
The gate resistors were selected at 22 Ohms in order
to lower the dV/dt which reduces the negative voltage
spike caused by fast transitions. The designer should
be careful not to make this transition too slow in
proportion to the switching frequency; if this condition is
not met, the element will never switch full on or off.
Please see Reference 11 from Avago Technologies for
information in selecting the correct gate resistor.
1.0 uf 25V capacitors were chosen as the bootstrap
capacitors. The voltage across the capacitor must not
be lower than Vg_on in between switching times; the
gate will fail to fire if this condition is not respected. The
diodes must be fast-acting in order to preserve the
charge on the capacitors.
 2014 Microchip Technology Inc.
DS00001660B-page 33
AN1660
TABLE 22:
NTC KEY PARAMETERS
Part Number
MF72-22D11
R25
22
Stead State
Current
2A
Approx. R of
Max. Current
0.563
According to the MF72-22D11 data sheet, the
maximum load capacitance at 220V is determined to
be 880 uF, which is higher than the inverter board’s
load capacitors of 560 uF. Its ambient resistance at
room temperature is 2 Ohms, which should limit the
current to a 220V/22 = 10A. The designer should also
check the diode bridge to ensure that the peak current
is within the ratings. The steady-state current must be
taken into consideration when driving larger motors or
when modifying the V/f curve. Recall that large
voltages at low frequencies will cause very large
currents in the stator. If the motor is kept in this mode,
the designer risks damaging the NTC.
The load capacitance is doubled from 280 uF (560/2) to
560 uF and the current is also doubled when using the
voltage doubler at 120V. Please take this into
consideration when choosing an NTC.
The manufacturer’s maximum current derating curve
must be looked at if the inverter board is to be placed
into an application where the ambient temperature is
not 25°Celsius. The resistance of the NTC is inversely
proportional to the rise of the ambient temperature.
The NTC must be given sufficient time to bring itself
back to room temperature after it has been consuming
current, before the inverter board is switched back on.
If the NTC has not properly reset its temperature to a
safe point and the inverter is switched on too soon, the
designer risks irreparable damage of the fuse. Since
large capacitors are being charged in this application,
the large time constant associated with these
capacitors will usually cause the NTC to release
thermal energy faster than the capacitors becoming
fully discharged. The bleed-off resistors of R8 and R13
should be adjusted to tweak the RC time constant.
Lastly, it is not uncommon for the NTC to heat up during
normal operation. According to Table 22, the
temperature for the NTC is as seen in Equation 17.
EQUATION 17:
NTC TEMPERATURE
2
2
P NTC = I R = 2 A  0.563 = 2.252W
T
P
2.252
NTC
= ------------------------------- = -------------  150C
NTC
0.015
C OEF F
NTC
Load Capacitance
Dissipation
Coefficient
15 mW/ °C
@ 120VAC
880 uF
@ 240VAC
220uF
Flyback Converter
The inverter uses a flyback converter to convert the
rectified DC voltage into two separate and isolated
power supplies. The LNK625 provides the regulation
instead of the microcontroller. A lower cost solution
should dedicate a PWM module plus other monitoring
software to control the flyback power supply.
The feedback resistors, R16 and R17, are carefully
chosen to bias the output voltage to around 14-15V. It
is critical that these resistors are 1%. The 15V is used
to supply power to the IGBT/FET driver chip. Its gate
voltage is directly proportional to its supply voltage. The
designer should verify that this gate voltage is sufficient
to drive the switching elements.
The fast-acting diodes used in the input filter and output
rectification should not be exchanged for general
purpose ones. The output rectification capacitors are
chosen to provide ample current to the low voltage
electronics; however, their capacitance value may still
be lowered and hence optimized after careful
considerations of the load. Place these capacitors as
close to the integrated circuits as possible.
The layout of the PCB should closely match that of the
LNK625 data sheet. For more design considerations,
please see the LNK623-626 LinkSwitch©-CV Family
data sheet (Reference 14).
Linear Regulators
The MCP1703A provides the step-down voltage from
16V to 5V on both secondaries of the transformer. The
MCP1703A is operating at its maximum input voltage
of 16V. If the flyback regulation causes voltage spikes
higher than the allowed maximum input voltage,
designers may want to look at the MCP16301 switching
regular for improved efficiency or at the MCP1804 as a
replacement.
Another consideration is the package of the regulators.
The optocouplers consume large currents, higher than
20 mA, while being biased in their active region. The
two potentiometers and the dedicated isolated output
have their associated optocoupler circuits biased this
way. It is important that the designer correctly identifies
the package for heat dissipation. The maximum power
dissipation is shown in Equation 18.
It is evident that the placement of the inrush limiter is
critical; therefore, it should be placed so as not to touch
or heat up any adjacent components.
DS00001660B-page 34
 2014 Microchip Technology Inc.
AN1660
EQUATION 18:
PACKAGE SIZING
FIGURE 44:
T
–T 
J – M AX
A
P MAX = --------------------------------------- JA
MOTOR RC FEEDBACK
FILTER
Where:
Symbol
Description
Units
Pmax
Maximum Power Dissipation
W
Maximum Junction
Temperature
°C
TA
Ambient Temperature
°C
JA
Thermal Resistance from
Junction to Ambient
TJ-max
Symbol
Imotor
Total current through the motor’s
stator.
RFeedback
A power resistor. The value must
be
carefully
calculated
in
proportion to
the expected
locked-rotor current of the motor.
D
A fast-acting diode that is used to
eliminate negative transients.
RFilter and
CFilter
Used to eliminate high-frequency
noise by creating a low-pass filter.
These are placed as close to the
microcontroller pin as possible.
°C/W
Solving for the MCP1703A in SOT-89 package:
 150C – 25C 
P MAX = ---------------------------------------- = 817 M W
153C  W
Therefore, to safely use the MCP1703A in an SOT-89
package, the maximum power must be limited to
817 mW. Power consumption higher than 817 mW
would require reducing the ambient temperature or
adding a heat sink.
The total allotted current for the SOT-89 regulator is as
shown in Equation 19.
EQUATION 19:
I
MAX
POWER CONSUMPTION IN
SOT-89 PACKAGE
P
MAX
= ----------------------------------V – V

IN
OUT
817 M W
= ----------------------------- = 74 M A
 16V – 5V 
If the above calculations were performed for an
MCP1703A in an SOT223 package, the maximum
current drawn from the 5V line can be up to 183 mA
without having to add extra thermal dissipation. The
current design uses the SOT223 package for flexibility.
Motor Feedback (Current Sense)
The current feedback system is a very simple and
low-cost solution. In some motor applications, a
high-side current reading circuit is employed due to its
accuracy and non-invasiveness. The solution provided
in Figure 44 presents a low-side current sense that
consists of only a few discrete components.
 2014 Microchip Technology Inc.
Description
The feedback resistor must be carefully selected. The
voltage drop across the resistor is used to determine if
an overcurrent scenario has been entered. The
resolution of the ADC is 4.88 mV per step (5V / 1023).
A resistor with high-resistance will induce a larger
voltage drop and hence a larger ADC resolution
between the signal and the noise floor. However, the
energy consumed by the resistor will be higher
compared to a resistor of a lower value.
The inverter has a 5W 0.22 Ohm resistor as the
feedback. With an expected maximum current of 4A,
the total voltage drop across it will be 0.88V. The total
power consumed is therefore 3.52W, which is below
5W. The designer has to avoid this voltage drop from
rising high enough so as to cause the IGBTs to switch
on. The minimum IGBT gate-to-emitter (VGS) voltage
for this inverter board is 10V. The IGBT driver supplies
a driving voltage of 15V. If the voltage across the
resistor rises to 5V, the designer risks the inverter
malfunctioning and damaging the driver.
The designer may also want to include a hardware
Fault-protection circuit to handle fast Fault type events.
DS00001660B-page 35
AN1660
Optocouplers
LIMITATIONS
The isolation barrier between the microcontroller and
the outside world is complete with optocouplers. The
optocoupler can be biased in two modes of operation:
Linear mode and Digital Logic mode. These two modes
can easily be modified by the designer by changing
only one resistor (R2), as shown in Figure 45.
This low-cost board has two limitations that the
designer should be aware of:
FIGURE 45:
THE COMMON-COLLECTOR
CIRCUIT
1.
2.
No power factor correction on the input stage
Decreased horsepower output compared to that
of a traditional PSC setup with an identical motor
RECOMMENDATIONS
These tests that were conducted can be further
expanded. The following lists some endeavors that
should enhance this document further:
1.
2.
3.
4.
Note:
When the forward current is ‘0’, the
output is low. When the optocoupler
LED is conducting, the output is high.
The optocouplers are kept within their linear region for
the potentiometer circuits so that every incremental
swipe of the potentiometer can produce an amplified
copied signal on the output. While the output is not a
direct 1:1 relationship, it is a sufficient and
cost-effective method in creating an isolated analog
signal. A downside of this method is that large currents
will be consumed by the optocoupler.
Digital Logic mode is used on the inverter board for all
isolated digital I/O with the exception of the dedicated
output. When the optocoupler LED is active, the output
is high as indicated in Figure 45. Rise and fall times
while operating in Digital Logic mode are much slower
since the load resistor R2 is increased. When R2 is
decreased, the rise (TR) and fall times (TF) decrease
dramatically. This is why the dedicated output and input
circuitries are biased in their active region.
The schematic seen in Appendix B: “Complete
Inverter Board Schematic” shows that the
multiplexed pins appear similar to Figure 45, with the
exception of a voltage divider with a 10K pull-down
resistor. The load resistors are carefully chosen to
create offsets that are equidistant from one another so
that the ADC on the microcontroller can easily
differentiate between signals.
Note:
5.
6.
Characterize more than one split-phase
motor
Test a three-phase motor
Rerun the existing tests at 115V input
with the voltage doubler
Connect another measuring element to
the three output PWM terminals.
Between this measurement and the input
power measurement, the designer can
calculate the efficiency of the inverter and
driving circuitry as separate entities
Power Factor Correction on the input
stage of the inverter
Implement Space Vector PWM (SVPWM)
instead of Sine PWM (SPWM) to
compare total harmonic distortion and
utilization
(please
see
VDC
Reference 15).
ACCEPTANCE TEST
Introduction
This section provides a quantitative test bench and
quick-start operation for the high-voltage and
low-voltage properties of the inverter board, which are
analyzed in this document.
This inverter was originally designed to drive a
single-phase AC induction motor; however, it can be
used to create any arbitrary phase difference between
the three outputs.
The designer must take into consideration
the 20-30 mA current consumption of the
optocoupler LED circuitry and its effect on
the maximum current capability of the
linear regulator (MCP1703A).
DS00001660B-page 36
 2014 Microchip Technology Inc.
AN1660
General Safety Notice
Power Supply
This board consumes high currents and voltages with
no isolation from the supplied input power. It is advised
that extreme caution is taken while performing any
measurements. The following are some general notes:
All measurements below (see Table 23 and Table 24)
are in Volts DC (VDC) unless otherwise specified
• If a scope is used, make sure that it is an isolated
model since the GNDs are floating with respect to
earth ground.
• It is safer to test the inverter using a DC power
supply with current limiting for the initial test.
60VDC can be used to verify the low-voltage
operation.
• The isolated section of the inverter has a separate
return path from the non-isolated one.
• It is recommended to use an isolating transformer
with the appropriate VA rating while testing and
developing.
TABLE 23:
POWER SUPPLY MEASUREMENTS
Description
.
(1)
Abbreviation
TP+
TP-
Min.
Max.
Expected
VIN
VIN(P3)
VIN(P3)
20VAC/
30VDC
230VAC/
325VDC
—
5V Microcontroller
Voltage
PIC_5V
6
PIC_GND
4.52
5.53
5.03
15V Microcontroller
Secondary Voltage
PIC_15V
7
PIC_GND
12.96
15.84
14.4
15V Isolated
Secondary Voltage
ISO_15V
8
ISO_GND
12.24
14.97
13.61
5V Isolated Voltage
ISO_5V
9
ISO_GND
4.51
5.51
5.01
VBUS (representative
of VIN)(3)
VBUS
10
PIC_GND
 VIN 2  – 1.4
 VIN 2  + 1.4
VIN 2
VBOOT
11
PIC_GND
5.38
6.58
5.98
Input Voltage(2)
VBOOTSTRAP LNK
Note 1:
2:
3:
4:
Actual(4)
TP = Test Point
An applied AC voltage that is above 115V must not have the jumper placed in circuit. Doing so will cause
the inverter to malfunction. There is no protection against this user error.
If VIN is an AC signal, VBUS = VIN * 2. If VIN is a DC signal, VBUS = VIN – 1.
To be filled in by the user.
 2014 Microchip Technology Inc.
DS00001660B-page 37
AN1660
Input / Output
TABLE 24:
I/O MEASUREMENTS(1)
Setup
Expected Actual(2) Notes
Abbreviation
TP+
TP-
Min.
Max.
5V connected between C and D
respectively
Aux0
1
PIC_GND
2.98
3.64
3.31
5V connected between E and F
respectively
Aux1
1
PIC_GND
2.25
2.75
2.50
5V connected between G and H
respectively
Aux2
1
PIC_GND
4.32
PIC_5V
4.80
Aux3
2
PIC_GND
4.48
PIC_5V
4.98
Isolated Auxiliary Inputs
Isolated Auxiliary Outputs
5V connected between A and B
Isolated Inverter Inputs (Switches)
Press button S0
SW0
3
PIC_GND
4.29
PIC_5V
4.77
Press button S1
SW1
3
PIC_GND
2.95
3.61
3.28
POT R36 turned fully clockwise
POT0
4
PIC_GND
0.00
0.00
0.00
POT R36 at middle position
POT0
4
PIC_GND
0.99
1.24
1.10
POT R36 turned fully
counter-clockwise
POT0
4
PIC_GND
4.31
PIC_5V
4.79
Isolated Inverter Inputs (POTs)
POT R37 turned fully clockwise
POT1
5
PIC_GND
0.00
0.00
0.00
POT R37 at middle position
POT1
5
PIC_GND
1.07
1.39
1.19
POT R37 turned fully
counter-clockwise
POT1
5
PIC_GND
4.32
PIC_5V
4.80
Note 1:
2:
TP = Test Point
To be filled in by the user.
Figure 46 shows how to test the isolated auxiliary
inputs by applying a +5V DC supply on terminals A and
B, C and D, E and F, G and H. Please connect to the
correct polarities on the terminals.
FIGURE 46:
ISOLATED AUXILIARY
INPUT TEST
The resistors for the multiplexed resources were
chosen to provide the largest voltage differential
between states.
DS00001660B-page 38
 2014 Microchip Technology Inc.
AN1660
Isolated Output
Inverter Board Operation
Figure 4 shows the required setup to test the isolated
digital output (Aux4). Terminal I should be connected to
a +5V power source. Terminal J should be connected
to ground through a 180Ω resistor. Isolated and
non-isolated grounds need to be bridged and
connected with the ground of the probe. Probe 1 should
measure at test point J. Probe 2 should measure at test
point 12. Figure 47 shows the expected output.
After the above measurements are validated, motor
drive operation should be tested.
FIGURE 47:
ISOLATED OUTPUT TEST
The software is capable of producing any output phase
difference. The most common offsets are typically 90 or
120 degrees. The default firmware produces a
90-degree offset for driving single-phase AC induction
motors. A three-phase ACIM may also be used with the
inverter after the configuration file is modified by using
the ONE_TWENTY_DEG
#define inside the
modulate.c file.
Test procedure:
1.
Upon power-up, the default code will present a 7.8 kHz
PWM waveform on the digital output channel as seen
in Figure 48.
FIGURE 48:
EXAMPLE OF PWM
EXPECTED OUTPUT
Connect the motors input wires to the U, V and
W outputs of the inverter.
2. Apply 230V AC power to VIN.
3. The PWR LED should light up, indicating power
to the inverter.
4. Turn POT0 at the midpoint position, facing
down. This is the stop position of the motor.
5. Press SW1 to start the motor.
6. Slowly turn POT0 fully clockwise. The motor
should spin clockwise. The motor light on the
inverter should blink at an increased rate,
proportional to the speed of the motor.
7. Turn POT0 fully counter-clockwise. After the
center-stop position (six o’clock) has been
passed, the motor should start spinning in the
opposite direction. The motor light should again
blink accordingly.
8. Press SW1 again to turn the motor off.
9. POT1 adjusts the current pinch feature of the
inverter. Press SW1 to start the board again.
Lower the current limit and when the limit has
been reached, the motor will shut off.
10. Apply 115V power to VIN with a jumper placed
on W1. Placing a jumper on W1 will double the
voltage going to the inverter.
Note:
The voltage with the jumper W1 in-circuit
must not exceed 115V, as there is no
protection against this user error.
Note 1: Scope 1 is the PWM output on P2.
2: Scope 2 is the PWM output on RC4.
 2014 Microchip Technology Inc.
DS00001660B-page 39
AN1660
APPENDIX A:
TABLE 25:
BILL OF MATERIALS
BILL OF MATERIALS
Quantity
Designator
Description
Part Number
2
C1, C7
CAP CER 470PF 100V 5% NP0 0805
CL21C471JCCNNNC
4
C2, C9, C24, C25
CAP ALUM 10UF 25V 20% RADIAL
UVZ1E100MDD
2
C3, C8
CAP ALUM 47UF 35V 20% RADIAL
ESH476M035AE3AA
4
C4, C5, C10, C18
Cap, Ceramic, 0.1uF, 50V X7R
08055C104MAT2A
10
C6, C14, C15, C16, C17,
C19, C20, C21, C22, C23
CAP CER 1UF 50V 10% X5R 080
UMK212BJ105KG-T
1
C11
CAP CER 820PF 2KV 10% X7R 1210
1210GC821KAT1A
2
C12, C13
CAP ALUM 560UF 250V 20% SNAP
EET-UQ2E561CA
6
D1, D2, D4, D7, D8, D9
DIODE ULTRA FAST 600V 1A SMA
STTH1L06A
1
D3
Full Wave Diode Bridge
GBU10M-BP
1
D5
LED, SMD, YEL, 0805 package
APT2012YC
2
D6, D10
LED, SMD, GRN, 0805 package
APT2012CGCK
1
D11
DIODE_FAST_SOD123
CFRMT107-HF
1
F1
FUSEHOLDER 22.5MM PCB 5X20MM
BK
0031.8201
1
NTC1
Current Limiter Inrush 100R 20%
NTC1MF72-010D11
1
P1
Header, PICkit™ 2, 1X6 0.1sp
TSW-106-07-F-S
1
P2
TERMINAL BLOCK 3.5MM 2POS PCB
ED555/2DS
1
P3
2PHDR-200
OSTTC022162
1
P4
TERMINAL BLOCK 5.08MM 3POS PCB EDZ350/3
1
P5
Term_BLK_8pos 3.5mm
ED555/8DS
2
POT0, POT1
3386P-1-202TLF
3386P-1-202TLF
6
Q1, Q2, Q3, Q4, Q5, Q6
Insulated Gate Bipolar Transistor with
Ultrafast Soft Recovery Diode
RG4BC20KDSTRLP
2
R1
RES 0.0 OHM 1/8W 0805 SMD
5
R2, R3, R45, R47, R48, R49 RES 510 OHM 1/8W 0805 SMD
RRMCF0805FT510R
2
R4, R7
RES 47 OHM 1/8W 0805 SMD
RMCF0805FT47R0
2
R5, R6
RES 820 OHM 1/8W 0805 SMD
RMCF0805FT820R
2
R8, R13
RES 100K OHM 1/4W 5% CARBON
FILM
CF14JT100K
1
R9
RES 270K OHM 1/4W 5% CARBON
FILM
RNMF14FTC270K
2
R10, R11
RES 4.7K OHM 1/8W 1% 0805 SMD
RMCF0805FT4K70
1
R12
ES 330 OHM 1/4W 5% CARBON FILM
CF14JT330R
5
R14,R15, R23, R24, R36
RES 1K OHM 1/8W 1% 0805 SMD
RMCF0805FT1K00
1
R16
ES 30.9K OHM 1/8W 1% 0805 SMD
RMCF0805FT30K9
1
R17
RES 6.04K OHM 1/8W 1% 0805 SMD
RNCP0805FTD6K04
5
R18, R20, R25, R40, R42
RES 10K OHM 1/8W 1% 0805 SMD
RMCF0805FT10K0
1
R19
RES 160K OHM 1/8W 1% 0805 SMD
RMCF0805FT160K
6
R21, R27, R34, R35, R38,
R43
RES 390 OHM 1/8W 0805 SMD
RMCF0805FT390R
2
R22, R44
RES 5.1K OHM 1/8W 1% 0805 SMD
RMCF0805FT5K10
1
R26
RES 0.22 OHM 5W 5% RADIAL
2-1623788-5
6
R28, R29, R30, R31, R32,
R33
RES 20 OHM 1/8W 1% 0805 SMD
RMCF0805FT20R0
DS00001660B-page 40
RMCF0805ZT0R00
 2014 Microchip Technology Inc.
AN1660
TABLE 25:
BILL OF MATERIALS (CONTINUED)
Quantity
Designator
Description
1
R46
2
S0, S1
SWITCH TACTILE SPST-NO 0.05A 24V B3S-1002
1
T1
Four-Winding Transformer (Non-Ideal)
750813144
1
U1
4-Pin Phototransistor Optocoupler
ACPL-217-500E
1
U2
8-Bit Micro
PIC16F1509-E/SO
1
U3
AC/DC Converter
LNK625DG
2
U4, U6
SOLATOR 3KVRMS 4CH TRANS 16SOI ACPL-247-500E
1
U5
IC MOSFET DRIVER
IRS2334SPbF
2
VR1, VR2
POS 5V 250mA, LDO Regulator SOT89
MCP1703T-5002E/MB
1
VR3
Resistor
MOV-07D471K
 2014 Microchip Technology Inc.
RES 6.8K OHM 1/8W 1% 0805 SMD
Part Number
RMCF0805FT6K80
DS00001660B-page 41
COMPLETE INVERTER BOARD SCHEMATIC
Non-Isolated
Isolated
R1
+5V
0R
AN1660
DS00001660B-page 42
APPENDIX B:
P1
AGND
DGND
1
2
3
4
5
6
Vpp/AUX0
PGD/Ramp
PGC
R2
VPP/MCLR
VDD
GND
ICSPDAT
ICSPCLK
NC
R3
U1
P2
Garage Light
510R
Garage Light
1
2
510R
ED1514-ND
AUX4
ACPL-217-500E
DGND
PICKIT2
DGND
VR1
C24
+5V 3
10uF 25V
C6
1.0uF
+15V
2
4
GND
TAB(GND)
10uF 25V
D1
1
VIN
VOUT
C2
MCP1703T-5002E/DB
R5
820R
C1
C3
C4
0.1μF
47uF 35V
750813144
+5V
C5
0.1μF
STTH1L06A R4
47R
470pF
10 T1
VR2
D2
9
1
R6
+5V
DGND
+5V
NTC1
R7
DGND
+Vbus
U2
R15
1K
D5
LED2
F1
MF72-010D11
1
HOT
NEU
C12
560uF 250V
D3
R9
270K
R8
100K
47R
8
C10
0.1μF
47uF 35V
470pF
2
4
VIN
VOUT
GND
TAB(GND)
C9
3 +5V ISO
10uF 25V
MCP1703T-5002E/DB
820R
C11
820pF 2KV
ISOGND
W1
C13
560uF 250V
R13
100K
2
4
R12
330R
AUX1
AUX2
AGND
R36
7
1
C8
2
3
GBU10M-BP
SCL
+5V
DGND
1K
D6
LED2
P3
1
SDA
PIC16F1509-E/SO
R14
2
4.7K
4
R11
4.7K
STTH1L06A
CLC3
R10
PGD/Ramp
PGC
CLC1
CLC2
PWM4
Speed
VR3
Vpp/AUX0
PWM1
Garage Light
PWM2
Isense
MOV
Remote Learn
Open/Close Door
20
19
18
17
16
15
14
13
12
11
VDD
VSS
RA5
RA0/PGD
RA1/PGC
RA4
RA3/MCLR
RA2
RC5
RC0
RC4
RC1
RC2
RC3
RB4
RC6
RC7
RB5
RB7
RB6
MOV-07D471K
1
2
3
4
5
6
7
8
9
10
STTH1L06A C7
AUX3
5
D4
1K
AGND
R16
D10
DGND
U3
EN/UV
BP/M
NC
D
LNK625DG
1
2
3
4
C14
1.0uF
+Vbus
Q3
IRG4BC20KDSTRLP
Q2
IRG4BC20KDSTRLP
Q1
IRG4BC20KDSTRLP
DGND
AGND
+5V ISO
S0
P4
IRG4BC20KDSTRLP
IRG4BC20KDSTRLP
IRG4BC20KDSTRLP
M3
Q6
+5V
R29
R30
R31
20R
20R
20R
R32
R33
20R
20R
C21
1.0uF
DGND
Remote Learn
Isense
D11
AGND
R18
C22
1.0uF
5.1K
10K
R34
390R
DGND
13
4
12
5
6
10
7
9
R38 390R
R20
R23
R27
R35
390R
8
VS3
VB3
IRS2334SPbF
POT1
2K
2K
3386P-1-202TLF
Curent Pinch
C23
1.0uF
C15
1.0uF
+5V
ED1520-ND
P5
284391-8
DGND
14
VS1
HO3
HO2
VS2
VB2
17
VB1
LIN2
LIN3
6
20
5
LO2
HO1
LO3
HIN3
HIN2
LIN1
4
3
2
CCW
ISOGND
C17
1.0uF
16
Vpp/AUX0
R46
6.8K
R44
AUX1
STTH1L06A
5.1K
D8
CLC3
AUX2
CLC1
STTH1L06A
D9
PWM2
U6
15
D7
DGND
PWM4
CCW
Speed
3386P-1-202TLF
+5V
C16
1.0uF
+15V
PWM1
10K
ISOGND
POT0
U5
DGND
CLC2
B3S-1002
R25
390R
ACPL-247-500E
12
13
15
18
8
16
9
19
10
11
VCC
LO1
HIN1
COM
7
10uF 25V
1
 2014 Microchip Technology Inc.
C25
S1
Start/Stop Motor
1K
ISOGND
+15V
C19
1.0uF
10K
ISOGND
1K
DGND
C18
0.1μF
B3S-1002
R24
3
11
PGD/Ramp
+15V
1
2
14
R22
Speed
DGND
U4
15
390R
Isense
R26
0.22R
ISOGND
16
R21
R19
CFRMT107-HF
R28
C20
1.0uF
Open/Close Door
Remote Learn
+5V ISO
ED1644-ND
160K
20R
+5V
1
2
3
M2
Q5
30.9K
R17
6.04K
8
7
6
5
AGND
M1
Q4
S
S
S
S
R42
AUX3
10K
DGND
2
14
3
13
4
12
5
R40
11
6
10K
10
7
R43
9
8
390R
STTH1L06A
1
ACPL-247-500E
1
R45
AUX0
510R
AUX1
510R
4
5
R48
AUX2
510R
6
7
R49
AUX3
510R
2
3
R47
8
AN1660
REFERENCES
1.
2.
3.
4.
5.
6.
7.
8.
9.
10.
11.
12.
13.
14.
15.
Speed Control of 3-Phase Induction Motor
Using PIC18 Microcontrollers, Padmaraja
Yedamale, AN843, Microchip Technology
AC Induction Motor Fundamentals, Rakesh
Parekh, AN887, Microchip Technology
V/f Control of Three-Phase Induction Motors
Using PIC16F7X7 Microcontrollers, Rakesh
Parekh, AN889, Microchip Technology
Determining MOSFET Driver Needs for Motor
Drive Applications, Jamie Dunn, AN898,
Microchip Technology
IGBTs vs HEXFET® Power MOSFETs for
Variable Frequency Motor Drives, Ajit Dubhashi
and Brian Pelly, AN980, International Rectifier
Bidirectional VF Control of Single and
Three-Phase Induction Motors Using the
PIC16F72, Padmaraja Yedamale, AN967,
Microchip Technology
An Introduction to AC Induction Motor Control
Using the dsPIC30F MCU, Steve Bowling,
AN984, Microchip Technology
Induction Motors Fed by PWM Frequency
Inverters, WEG
Application Note for Inrush Current Limiters,
EPCOS
Induction Motor Speed Torque Characteristics,
Yaskawa Electric America, Inc
Gate Drive Optocoupler Basic Design for IGBT/
MOSFET, AN967, Microchip Technology
Bidirectional VF Control of Single and ThreePhase Motors Using the PIC16F72, Padmaraja
Yedamale, AN967, Microchip Technology
Efficiency Improvement of Permanent-Split
Capacitor Motors in HVAC Applications Using a
Two-Phase Asymmetrical Inverter, Anderson, K.
LNK623-626 LinkSwitch-CV Family Data Sheet,
Power Integrations
VF Control of 3-Phase Induction Motor Using
Space Vector Modulation, Rakesh Parekh,
AN955, Microchip Technology
 2014 Microchip Technology Inc.
DS00001660B-page 43
AN1660
NOTES:
DS00001660B-page 44
 2014 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
arising from this information and its use. Use of Microchip
devices in life support and/or safety applications is entirely at
the buyer’s risk, and the buyer agrees to defend, indemnify and
hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights.
Trademarks
The Microchip name and logo, the Microchip logo, dsPIC,
FlashFlex, flexPWR, JukeBlox, KEELOQ, KEELOQ logo, Kleer,
LANCheck, MediaLB, MOST, MOST logo, MPLAB,
OptoLyzer, PIC, PICSTART, PIC32 logo, RightTouch, SpyNIC,
SST, SST Logo, SuperFlash and UNI/O are registered
trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
The Embedded Control Solutions Company and mTouch are
registered trademarks of Microchip Technology Incorporated
in the U.S.A.
Analog-for-the-Digital Age, BodyCom, chipKIT, chipKIT logo,
CodeGuard, dsPICDEM, dsPICDEM.net, ECAN, In-Circuit
Serial Programming, ICSP, Inter-Chip Connectivity, KleerNet,
KleerNet logo, MiWi, MPASM, MPF, MPLAB Certified logo,
MPLIB, MPLINK, MultiTRAK, NetDetach, Omniscient Code
Generation, PICDEM, PICDEM.net, PICkit, PICtail,
RightTouch logo, REAL ICE, SQI, Serial Quad I/O, Total
Endurance, TSHARC, USBCheck, VariSense, ViewSpan,
WiperLock, Wireless DNA, and ZENA are trademarks of
Microchip Technology Incorporated in the U.S.A. and other
countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
Silicon Storage Technology is a registered trademark of
Microchip Technology Inc. in other countries.
GestIC is a registered trademarks of Microchip Technology
Germany II GmbH & Co. KG, a subsidiary of Microchip
Technology Inc., in other countries.
All other trademarks mentioned herein are property of their
respective companies.
© 2014, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
ISBN: 978-1-63276-868-1
QUALITY MANAGEMENT SYSTEM
CERTIFIED BY DNV
== ISO/TS 16949 ==
 2014 Microchip Technology Inc.
Microchip received ISO/TS-16949:2009 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
DS00001660B-page 45
Worldwide Sales and Service
AMERICAS
ASIA/PACIFIC
ASIA/PACIFIC
EUROPE
Corporate Office
2355 West Chandler Blvd.
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Technical Support:
http://www.microchip.com/
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www.microchip.com
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Tel: 905-673-0699
Fax: 905-673-6509
DS00001660B-page 46
China - Chongqing
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Fax: 86-23-8980-9500
China - Hangzhou
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Fax: 86-571-8792-8116
China - Hong Kong SAR
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Fax: 852-2401-3431
China - Nanjing
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China - Qingdao
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China - Shenyang
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Fax: 86-24-2334-2393
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Fax: 86-755-8203-1760
China - Wuhan
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Tel: 82-53-744-4301
Fax: 82-53-744-4302
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Fax: 82-2-558-5932 or
82-2-558-5934
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Tel: 63-2-634-9065
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Fax: 886-3-5770-955
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Fax: 34-91-708-08-91
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Tel: 44-118-921-5800
Fax: 44-118-921-5820
Taiwan - Kaohsiung
Tel: 886-7-213-7830
Taiwan - Taipei
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Fax: 886-2-2508-0102
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Fax: 66-2-694-1350
China - Xiamen
Tel: 86-592-2388138
Fax: 86-592-2388130
China - Zhuhai
Tel: 86-756-3210040
Fax: 86-756-3210049
03/25/14
 2014 Microchip Technology Inc.