AN695 Interfacing Pressure Sensors to Microchip’s Analog Peripherals Author: This application note will concentrate on the signal conditioning path of the piezoresistive sensing element from sensor to microcontroller. It will show how the electrical output of this sensor can be gained, filtered and digitized in order to ready it for the microcontroller’s calibration routines. This theoretical discussion will be followed with a specific pressure sensing design that is specifically designed to measure barometric pressure. Bonnie Baker Microchip Technology Inc. INTRODUCTION Pressure measurement devices can be classified into two groups: those where pressure is the only source of power and those that require electrical excitation. The mechanical style devices that are only excited by pressure, such as bellows, diaphragms, bourdons, tubes or manometers, are usually suitable for purely mechanical systems. With these devices a change in pressure will initiate a mechanical reaction, such as a change in the position of mechanical arm or the level of liquid in a tube. PIEZORESISTIVE PRESSURE SENSORS The piezoresistive is a solid state, monolithic sensor that is fabricated using silicon processing. Piezo means pressure, resistance means opposition to a DC current flow. Since piezoresistive pressure sensors are fabricated on a wafer, 300 to 500 sensors can be produced per wafer. Since these wafers generate a large number of sensors they are available on the market at a reduced cost as compared to mechanical sensors. Electrically excited pressure sensors are most synergistic with the microcontroller environment. These style of sensors can be piezoresistive, Linear Variable Differential Transformers (LVDT), or capacitive sensors. Most typically, the piezoresistive sensor is used when measuring pressure. Voltage or Current Excitation RS1 Voltage or Current Excitation RS1 RS2 RS2 VOUTRS4 RS3 VOUT+ (a.) single element bridge VOUTRS4 RS3 VOUT+ (b.) two element bridge Dielectric Voltage or Current Excitation RS1 Si-P RS2 VOUTRS4 RS3 Contact Contact Silicon Substrate VOUT+ (c.) four element or full bridge Si-N Diaphragm (d.) single side of a sandwiched piezoelectric pressure sensor Figure 1: The resistive wheatstone bridge configuration can have one variable element (a.), two elements that vary with excitation (b.) or four elements (c.). The piezoresistive pressure sensing element is usually a four element bridge and is constructed in silicon (d.). 2000 Microchip Technology Inc. Preliminary DS00695A-page 1 AN695 With this sensor, the resistors are arranged in a full wheatstone bridge configuration, which has improved sensitivity as compared to a single element or two element sensors (see Figure 1.d). When a positive differential pressure is applied to the four element bridge, two of the elements respond by compressing and the other two change to a tension state. When a negative differential pressure is applied to the sensor, the diaphragm is strained in the opposite direction and the resistors that were compressed go into a tension state, while the resistors that were in a tension state change into a compression state. Piezoresistive pressure sensors may or may not have an internal pressure reference. If they do, a pressure reference cavity is generally fabricated by sealing two wafers together. The top side of this fabricated sensor is the resistive material and the bottom is the diaphragm. The high side of the piezoresistive bridges shown in Figure 1 can have a voltage excitation or current excitation applied. Although the magnitude of excitation (whether it is voltage or current) effects the dynamic range of the output of the sensor, the maximum difference between VOUT+ and VOUT- generally ranges from 10s of millivolts to several hundred millivolts. The electronics that follow the sensor are used to change the differential output signal to single ended as well as gain and filter it in preparation for digitization. ELECTRONICS SIGNAL PATH There are several ways of capturing the small differential output signal of the sensor and transforming it into a usable digital code. One approach that can be taken is shown in the block diagram in Figure 2.a. With this approach, the small differential output of the bridge is gained and converted from differential to single ended with an instrumentation amplifier (IA). The signal may or may not travel through a multiplexer. The signal then passes through a low pass filter. The low pass filter eliminates out-of-band noise and unwanted frequencies in the system before the A/D conversion is performed. This is followed by a stand-alone A/D converter which transforms the analog signal into a usable digital code. The microcontroller takes the converter code, further calibrates and translates if need be for display purposes. In this signal path only one analog filter is required and it is positioned at the output of the multiplexer. The second signal path shown in Figure 2.b also has an instrumentation amplifier (IA) in the signal path. Following the instrumentation amplifier stage the signal is filtered in the analog domain and then digitized with an on-chip microcontroller’s A/D converter. When this type of signal path is used, every signal going into the multiplexer will require its own analog filter. Additionally, the accuracy and speed of the converter in the microcontroller is less than a stand-alone A/D converter. This may or may not be an issue in a particular application. IA (a.) MUX FILTER SAR A/D MicroController REF (b.) IA FILTER MUX A/D µC VREF (c.) MUX Low Pass Filter COMP µC VREF Figure 2: Three block diagrams for the piezoresistive pressure sensor signal conditioning path are shown in this Figure. The top two block diagrams, a. and b., are discussed in detail in this application note. The bottom block diagram (c.) is discussed in detail in AN717 (Microchip Technology Inc.). DS00695A-page 2 Preliminary 2000 Microchip Technology Inc. AN695 INSTRUMENTATION AMPLIFIER OPTIONS AND DESIGN ational amplifiers, thereby significantly reducing source impedance mismatch problems at DC. The transfer function of this circuit is equal to: With this application, the two low voltage signals from the bridge need to be subtracted in order to produce a single ended output signal. The results of this subtraction also need to be gained so that it matches the input range of the A/D converter. The implementation of the subtraction and gain functions are done so that the sensor signal is not contaminated with additional errors. The instrumentation amplifier circuits shown in Figure 3 and Figure 4 achieve all of these goals. Both of these configurations take two opposing input signals, subtract them and apply gain. The subtraction process inherently rejects common-mode voltages. Combined with these functions the signal is level shifted, making it synergistic with the signal supply environment. R4 1 R2 R3 V OUT = ( V IN+ – V IN- ) ------- 1 + --- ------- + ------- R3 2 R1 R4 R2 + R3 + -------------------- + V CM RG 4 R3 R 2 R - ------- – ------- + V REF -----R 3 R4 R 1 It should be noted from this transfer function that the input signals are gained along with the common-mode voltage of the two signals. The common-mode voltage can be rejected when R1 = R4 and R2 = R3. Given this change the transfer function becomes: R 1 2R1 VOUT = ( V IN+ – V IN- ) 1 + ------- + ---------- + VREF R2 RG The Two Op Amp Instrumentation Amplifier The common-mode rejection error that is caused by resistor mismatch is equal to: A solution to the circuit problem discussed above is shown in Figure 3. The circuit in Figure 3 uses two operational amplifiers and five resistors to solve this gain and subtraction problem. R2 1 + ------R1 CMR = 100* ---------------------------------------------------------- ( % of mismatch error ) Dual amplifiers are usually used in this discrete design because of their good matching of bandwidth and over temperature performance. This instrumentation amplifier design uses the high impedance inputs of the oper- RG R2 R4 VDD - VREF R1 R3 A1 MCP602 VIN- A2 + MCP602 + VIN+ R 1 2R 1 4 + 60kΩ VOUT = 1 + ------- + ---------- ( V IN+ – V IN- ) + VREF = ------------------------ ( VIN+ – VIN- ) + VREF RG R 2 RG Where R1 = 30kΩ and R2 = 10kΩ Figure 3: The two op amp instrumentation amplifier takes the difference of two input signals, gains that difference, while rejecting any voltage that is common to both of the input signals. 2000 Microchip Technology Inc. Preliminary DS00695A-page 3 AN695 The ac common mode rejection for this configuration is poor. This is due to the fact that the common mode signal at VIN- is inverted once with A1 and then it travels through A2 causing a second propagation delay. The common mode signal at VIN+ only travels through one operational amplifier (A2). Additionally, the two operational amplifiers have different closed loop gains, and consequently different closed loop bandwidths. The second factor that limits the common-mode input range of this circuit comes from the input swing restrictions of the amplifiers themselves. If this circuit is in a single supply environment, it will typically require a reference that is centered at the common-mode voltage of the input signals. In Figure 4, VREF serves that function. This voltage can be implemented discretely with a precision reference chip as shown in Figure 4.a or with two equal resistors in series between the power supply as shown in Figure 4.b. In terms of common-mode input range, there are two factors that limit the range of this instrumentation amplifier. The first factor involves the operation of A1 as it responds to the VIN- and VIN+ input signals and the voltage reference, VREF. The signal at the non-inverting input to A1 and A2 gained by the output of A1 by: V OUT – A1 = V IN- Another added benefit to matching R2/R1 = R3/R4 is that the gain of the circuit can be changed with one resistor, RG. This instrumentation amplifier circuit has high impedance inputs and programmable gain capability. The features that could be improved in this circuit solution is to have the common-mode rejection independent of gain and better over frequency. These performance characteristics can only be obtained by an instrumentation amplifier configuration that has three operational amplifiers. ( R G R 2 + R 1 R 2 + R 1 R G ) --------------------------------------------------------------- ( R 1 R2 ) R2 R2 – V IN+ -------- – V REF ------- RG R1 R1 = R1A || R1B VDD VDD VREF Precision Voltage Reference R1A R2 R1B RG R1 VDD - VREF OR VIN- R2 MCP602 - + MCP602 + VOUT VIN+ (a) (b) Figure 4: The reference voltage for a two op amp instrumentation amplifier in a single supply environment can be implemented with a stand-alone voltage reference (a) or a resistor divider across a voltage reference or the supply voltage (b). DS00695A-page 4 Preliminary 2000 Microchip Technology Inc. AN695 all gains as long as the signals stay within A1 and A2 input and output head room limitations. If the common mode errors of the input amplifiers track they will be cancelled by the output stage. The Three Op Amp Instrumentation Amplifier An example of a more versatile instrumentation amplifier configuration is shown in Figure 5. With this circuit configuration, two of the three amplifiers (A1 and A2) gain the two input signals. The third amplifier, A3, is used to subtract the two gained input signals, thereby providing a single ended output. The transfer function of this circuit is equal to: If the assumption that R1/R2 equals R3/R4 is not correct, there could be a noticeable common mode voltage error. The calculated common-mode rejection (CMR) error that is attributed to resistor mismatches in this circuit is equal to: 2R F2 R1 + R2 V OUT = VIN+ 1 + -------------- R 4 ---------------------------------- – R G ( R 3 + R 4 )R 1 R2 1 + ------R1 CMR = 100* ---------------------------------------------------------- ( % of mismatch error ) R1 + R2 2R F1 R 2 V IN- 1 + -------------- ------- + V REF R 3 ---------------------------------- ( R 3 + R 4 )R 1 RG R1 for R 1 = R 3 and R 2 = R 4 If RF2 = RF1, R1 = R3, and R2 = R4, this formula can be simplified to equal: 2R F VOUT = ( V IN+ – V IN- ) 1 + ----------- + V REF RG An example of the impact of this error is demonstrated with a 12-bit, 5V system, where the gain of the circuit is 100V/V, the common-mode voltage ranges 0 to 5V and the matching error can be as large as ±1%. Using the formula above, the contributed error of this type of common-mode excursion is equal to 1mV. This voltage is slightly less than 1LSB. Quad amplifiers are typically used in the three op-amp instrumentation amplifier discrete designs because of the matching qualities of amplifiers with the same silicon. In contrast to the two op-amp instrumentation amplifier, the input signal paths (at VIN+ and VIN-) are completely balanced. This is achieved by sending VIN+ and VIN- signals through the same number of amplifiers to the output and using a common gain resistor, RG. In a single supply environment, the voltage reference should be equal to the center of the input signals. This voltage is represented in the circuit in Figure 5 as VREF. The purpose and effects of this reference voltage is to simply shift the output signal into the linear region of the amplifier. Since this input stage is balanced, common mode currents will not flow through RG. The common-mode rejection of this circuit is primarily dependent on the resistor matching around A3. When R1 = R2 = R3 = R4, common mode signals will be gained by a factor of one regardless of gain of the front end of the circuit. Consequently, large common mode signals can be handled at VIN- + A1 MCP604 R2 RF1 R1 A3 RG MCP604 RF2 R3 A2 VOUT R4 MCP604 VIN+ + + VREF Where RF1 = RF2 and R1 = R2 = R3 = R4 2R F VOUT = 1 + ----------- ( V IN+ – V IN- ) + VREF RG Figure 5: This is a three op amp implementation of an instrumentation amplifier. 2000 Microchip Technology Inc. Preliminary DS00695A-page 5 AN695 R2 From the output of A1 Precision Voltage Reference VDD From the output of A2 R1 - A3 R1 MCP604 + VOUT VDD VSHIFT OR VSHIFT R2A R2B R2 = R2A || R2B (a) (b) Figure 6: The reference voltage in Figure 5 can be implemented by using a precision reference circuit (a.) or a resistive voltage divider circuit (b.). The VREF circuit function can be implemented with a precision voltage reference or with the resistive network shown in Figure 6. ANALOG FILTERING A big topic for debate in digital design circles is whether or not an analog filter is needed and more importantly, can a digital filter replace the analog filter. A common assumption with designers that are trying to tackle analog challenges of this type is that they claim that they are only measuring DC so they don’t have to worry about filtering. Unfortunately, the noise generators in the electronics and the environment do not have the “intelligence” to accommodate the designer’s desires. Consequently, if a filter is not included, the circuit will be surprisingly noisier than anticipated. Once it is accepted that a filter is required, the next debate that ensues is whether the filtering strategy should be analog, digital or both. These signals are usually unintentional, but almost always destructive if not controlled. On the down side, analog filters can add to the noise floor particularly if a noisy amplifier is used with a large gain. Where analog filters earn their worth by rejecting noise in the out of band region, digital filters can be utilized to reduce the in-band noise floor. This is implemented with oversampling algorithms. These types of filters are much easier to change on the fly because it is a matter of programming instead of a matter of changing resistors and capacitors as it is with analog filters. With all of these benefits there is a price to pay in terms of response time. Digital filters must collect a certain amount of conversion data before calculations can be performed. The digital filter algorithms tend to slow down the response time as well as delay the output. If real time responses are not critical, the digital filter disadvantages are not detrimental to the operation of circuit. A common assumption that is made by programmers is that they can eliminate all ills with digital filtering. To some extent this is true, however, it is at a high price of time and memory and truthfully, it may not be possible to succeed. Analog filtering removes a considerable amount of headaches for the programmer from the start. Analog filters have their place in circuit designs as do digital filters. For instance, analog filters will eliminate aliasing errors that will occur through the A/D conversion process if they are allowed to go through. Once these errors are allowed in the conversion it is impossible to discriminate good signal from aliased signal in the digital domain. The analog filter also removes large signal noise that is generated by spikes or spurs in the signal. DS00695A-page 6 Preliminary 2000 Microchip Technology Inc. AN695 C1 R2 VIN R1 C2 R3 + MCP 606 - VOUT R4 Figure 7: By using FilterLabTM software, this 2nd order low pass filter that has a non-inverting gain in the pass band can be configured as a Butterworth, Bessel, or Chebyshev filter. As discussed previously, the hardware implementation of a low pass filter at its most fundamental level requires a capacitor and resistor for each pole. Active filters, which have one amplifier for every two poles, have the added benefit of preventing conflicting impedances and degrading the signal path. Close inspection of this filter shows that the circuit can be configured in a gain of +1V/V by shorting R4 and opening R3. In this configuration it is likely that the input of the amplifier will be exercised across a full rail-to-rail input range. The second order Multiple Feedback circuit implementation of a low pass filter uses an amplifier, three resistors and two capacitors, as is shown in Figure 8. The DC gain of this filter is negative and easily adjusted with the ratio of R3 and R1. When used in a single supply environment, this circuit usually needs a voltage reference on the non-inverting input of the amplifier. The 2nd order lower pass filter shown in Figure 7 is one of a class of circuits that were described in 1995 by R.P Sallen and E.L. Key. With this filter the DC gain is positive. In a single supply environment this eases the implementation considerably, because a mid-supply reference is not required. This circuit not only filters high frequencies, but it can be used to gain the incoming signal. This is the filter circuit that will be used in a barometric pressure application. An adjustable voltage reference will be Included in this filter design. R3 VIN R1 R2 C2 C1 MCP 606 + VOUT VREF Figure 8: By using FilterLabTM software, this 2nd order low pass filter that has an inverting gain in the pass band can be configured as a Butterworth, Bessel, or Chebyshev filter. 2000 Microchip Technology Inc. Preliminary DS00695A-page 7 AN695 BAROMETRIC PRESSURE SENSING Parameter (w/ 5V excitation) Specification The considerations for the design of a barometric sensing system encompasses altitude and resolution. The expected altitude that our sensor will be placed in is approximately from sea level to 20,000 ft. The nominal pressure at sea level is 14.7 psi and the nominal pressure at 20,000 ft. is 6.75 psi. The difference in pressure between these two altitudes is 7.95 psi. With this range, the appropriate pressure sensor should be an absolute version that is referenced to an on-chip vacuum and have a range up to 15psi. Since the change in pressure for major weather changes is approximately 0.18 psi, a resolution of 0.015 psi is over ten times more accurate than the measured value. The circuit that will be used for this design discussion is shown in Figure 9. Operating Pressure Range 0 to 15 psi Sensitivity 6.0mV/psi (typ) Full-Scale Span 90mV (typ) Zero Pressure Offset ± 300µV Temperature Effect on Span (0 to 70°C) ± 0.5% FSO Temperature Effect on Offset (0 to 70°C) ± 500µV Table 1: The specifications of the SCX015 from SenSym indicates that this is a good pressure sensor that can be used to measure barometric pressure. The specifications of the SCX015 from SenSym indicates that this is a good pressure sensor that can be used to measure barometric pressure. The critical pressure sensor specifications for this application include the operating pressure range, sensitivity, room temperature (25°C) span and offset errors as well as over temperature (see Table 1). Although, the range of this sensor extends from 0 psi to 15 psi, this application will not be using that lower range. The minimum differential output voltage from the sensor will be 40.5mV (6.75psi or 20,000 ft.) and the maximum sensor voltage will be 88.2mV (14.7psi or sea level). The voltage at the output of the sensor is gained before it is digitized using an instrumentation amplifier. Note that temperature issues are beyond the scope of this application note. Detailed information about temperature sensing circuits can be found in Microchip’s AN679, AN684, AN685, and AN687. Instrumentation Amplifier RG R2 2nd Order Butterworth Low Pass Filter 1/2 R1 R2 R2 R4 R2 MCP602 + VDD = 5V R1 VDD A1 R1 1/2 MCP602 + A2 R3 C1 R5 C2 SCX 015 1 MCP 606 + 2 A3 3 8 SCLK MCP3201 DOUT CS A4 4 R1 = 30kΩ R2 = 10kΩ RG = 1.15 kΩ R3 = 95.7 kΩ R4 = 172 kΩ R5 = 304 kΩ C1 = 0.22 µF C2 = 0.22 µF A1 = A2 = A3 = Single Supply, CMOS op amp A4 = 12-bit, A/D SAR Converter 35.7kΩ SCK SI CS PIC16C6xx R1 A6 A5 10kΩ 68.1kΩ Level Shift Voltage and Offset Adjust A4 and A6 can be replaced with a PICmicro that has an on-chip A/D converter A5 = 10kΩ Digital Potentiometer Figure 9: The voltage at the output of the SCX015 pressure sensor is gained by the instrumentation amplifier (A1 and A2) then filtered, gained and level shifted (A4) with a 2nd order low pass filter (A3) and digitized with a 12-bit A/D converter (A5). DS00695A-page 8 Preliminary 2000 Microchip Technology Inc. AN695 Instrumentation Amplifier Design This sensor requires voltage excitation. In order to determine the required gain of the circuit in Figure 9 the relationship between the maximum sensor output and allowable instrumentation amplifier output is used in the calculation. As stated previously, the maximum differential output of the sensor is 88.2mV. The allowable output range of the instrumentation amplifier is equal to VDD - 100mV. In a five volt system where VDD = 5V, the amplifier output maximum is equal to 4.9V. The minimum output of the sensor is 40.5mV. Since this is a positive voltage and the instrumentation amplifier is in a single supply environment, this minimum sensor output voltage will not drive the output of the instrumentation amplifier below ground. Consequently, the reference voltage called out in Figure 3 and Figure 4 is made to be equal to ground. Gain is calculated by dividing the maximum output voltage with the maximum input voltage. Using this calculation, the appropriate gain for our system is 55.6V/V. By using the gain formula in Figure 5: 2R 1 R G = ---------------------------------------R 1 Gain – 1 – ----- R 2 The gain and offset adjust features of this filter are also used in this segment of the application circuit. Given that the output from the instrumentation amplifier is 2.3V to 4.9V, the peak-to-peak voltage of this signal is 2.6V. A gain of 1.8V/V will produce an output swing of approximately 4.7V peak-to-peak. The adjustable offset voltage of this circuit which is gained by 2.8V/V will be configured to insure that the signal will fall at the output of the amplifier between the supplies. This adjustment circuit can also be used to remove system offset errors that originate in the sensor or instrumentation amplifier. The filter circuit in Figure 9 can be designed with the FilterLabTM software from Microchip Technology. The two capacitors are adjusted using the FilterLab program to be equal to 0.22uF. This adjustment is made in order to keep the capacitor packages small enough so surface mount capacitors can be used. If R 1 = 30kΩ and R 2 = 10kΩ, 60kΩ R G = -------------------------( 55.6 – 4 ) = 1.15kΩ (closest 1% value) With this gain, the maximum output of the IA will be 88.2mV*55.6V/V or 4.9V and the minimum output will be 40.5mV*55.6V/V or 2.3V. Since the gain of this instrumentation amplifier stage is relatively large, it is desirable to use an amplifier that has a low offset voltage. The MCP607, dual CMOS amplifier has a guaranteed input offset voltage of 250µV (max). This amplifier’s low quiescent current of 25µA (max) make this device attractive for battery powered applications. The offset adjust of the filter circuit is implemented with a 10kΩ digital potentiometer in series with a 68.1kΩ and 35.7kΩ resistors. The range of the offset adjust portion of this circuit at the wiper of the digital potentiometer is from 3.0V to 3.4V. This offset circuitry is gained by the filter/amplifier circuit so that the nominal value of the offset circuitry in combination with the sensor signal is equal to: VOUT-FILTER = -1.8V/V (Nominal Input Signal) + 2.8V/V (Nominal Reference voltage) VOUT-FILTER = -1.8V/V (3.6V) + 2.8V/V (3.2V) VOUT-FILTER = 2.48V A key amplifier specification for this filtering circuit is input voltage noise. The MCP601, single CMOS amplifier has a typical noise density of 29 nV/√Hz @ 1kHz. A/D Converter Design Filter Design Now that the signal from the pressure sensor has been properly differentiated and gained, noise is removed to making the results from the 12-bit A/D conversion repeatable and reliable. Remember that the output of the instrumentation amplifier circuit does not swing a full 0V to 5V. Consequently, the filter stage will also be used to implement a second gain cell as well as offset adjust. The stop frequency of this filter is 60Hz. This will removes any mains frequencies that may be aliased back into the signal path during conversion. This being the case, the cut-off frequency is selected to be 10Hz. 2000 Microchip Technology Inc. Any cut-off frequency lower than 10Hz, requires capacitors that are too large, making the board implementation awkward. The total attenuation between 10Hz and 60Hz is approximately -30dB. In other words, a 60Hz signal that is part of the output signal of the instrumentation amplifier is attenuated by 0.031 times. Keeping in mind that the instrumentation amplifier has already rejected a major portion of any 60Hz common-mode signal, this level of attenuation is enough to remove any remaining 60Hz noise that exists in the signal path. The final design step for this analog signal path is to insert the analog-to digital converter. The converter quantizes a continuous analog signal into discrete buckets. The appropriate converter can be selected once it is determined how many bits the application requires. The range of the analog signal has been closely matched to the input range of a zero to 5V in A/D converter. The barometric pressure range is 14.7 psi to 6.75 psi. The expected increase from good weather to a strong storm system would be approximately 0.18 psi. Given Preliminary DS00695A-page 9 AN695 this, the equipment should resolve to at least 0.015psi. This is easily achieved with a 10-bit converter. If resolution to 0.002 is needed a 12-bit a/d converter would be more suitable. Microchip has a large variety of analog to digital converters that can be used for this application. If the stand-alone solution is appealing, the MPC320X family of 12-bit and the MCP300X 10-bit family of converters are available. Generally speaking, stand-alone A/D converters have better accuracy than those compared to on-board converters. They also have features such a pseudo differential inputs and faster conversion speeds. The pseudo differential capability of these devices allow for configurations that reject small common mode signals. Additionally, the single channel devices can be used in simultaneous sampling applications, such as motor control. The application circuits using the singe converter also require fewer analog filters because the multiplexer is typically placed before the anti-aliasing filter. If an on-board a/d converter fits the application better, the PICmicro line has a large array of converters combined with other peripherals on a variety of micros that can be used. The integrated solution offers a degree of flexibility that the stand-alone solution does not have. This flexibility comes in the form of operational flexibility where the device’s voltage reference and sampling speed can be reconfigured on the fly. The I/O configuration is also very flexible allowing for easy implementation of the board layout. The stand-alone and integrated A/D converters from Microchip are both suitable for the pressure sensor circuit that is shown in Figure 9. CONCLUSION The design challenge that has been tackled in this application note is gaining, filtering, and digitizing the small differential signal of a pressure sensor bridge. In order to achieve this goal, we used a two-op amp instrumentation amplifier which gained the differential signal from the pressure sensor and converted it to a signal ended output. After this gain stage, a 2nd order, Butterworth, anti-aliasing filter was used to reduce noise so that the A/D converter could achieve a full 10bit accuracy. The suggested A/D converter strategy could be on board or off board and the trade-offs were presented. Digital filtering was not needed in this application. REFERENCES Tandeske, Duane, Pressure Sensors, Marcel Dekker, Inc., 1991 “Anti-Aliasing Analog Filters for Data Acquisition Systems”, Baker, Bonnie C., AN699, Microchip Technology Inc. “Making a Delta-Sigma converter with a Microcontroller”, Baker, Peter, Darmawaskita, Butler, AN700, Microchip Technology, Inc. “Using Operational Amplifiers for Analog Gain in Embedded System Design”, Baker, Bonnie C., AN682, Microchip Technology, Inc. “Building a 10-bit Bridge Sensing Circuit using the PIC16C6XX and MCP601 Operational Amplifier”, Baker, Bonnie C., AN717, Microchip Technology, Inc. “Precision Temperature Sensing with RTD Circuits”, Baker, Bonnie C., AN687, Microchip Technology, Inc. “Temperature Sensing Technologies”, Baker, Bonnie C., AN679, Microchip Technology, Inc. “Thermistors in Single Supply Temperature Sensing Circuits”, Baker, Bonnie C., AN685, Microchip Technology, Inc. “Single Supply Temperature Sensing with Thermocouples”, Baker, Bonnie C., AN684, Microchip Technology, Inc. DS00695A-page 10 Preliminary 2000 Microchip Technology Inc. AN695 NOTES: 2000 Microchip Technology Inc. Preliminary DS00695A-page 11 WORLDWIDE SALES AND SERVICE AMERICAS AMERICAS (continued) Corporate Office Toronto Singapore Microchip Technology Inc. 2355 West Chandler Blvd. Chandler, AZ 85224-6199 Tel: 480-786-7200 Fax: 480-786-7277 Technical Support: 480-786-7627 Web Address: http://www.microchip.com Microchip Technology Inc. 5925 Airport Road, Suite 200 Mississauga, Ontario L4V 1W1, Canada Tel: 905-405-6279 Fax: 905-405-6253 Microchip Technology Singapore Pte Ltd. 200 Middle Road #07-02 Prime Centre Singapore, 188980 Tel: 65-334-8870 Fax: 65-334-8850 Atlanta Microchip Technology, Beijing Unit 915, 6 Chaoyangmen Bei Dajie Dong Erhuan Road, Dongcheng District New China Hong Kong Manhattan Building Beijing, 100027, P.R.C. Tel: 86-10-85282100 Fax: 86-10-85282104 Microchip Technology Inc. 500 Sugar Mill Road, Suite 200B Atlanta, GA 30350 Tel: 770-640-0034 Fax: 770-640-0307 Boston ASIA/PACIFIC China - Beijing ASIA/PACIFIC (continued) Taiwan Microchip Technology Taiwan 10F-1C 207 Tung Hua North Road Taipei, Taiwan Tel: 886-2-2717-7175 Fax: 886-2-2545-0139 EUROPE China - Shanghai Denmark Microchip Technology Unit B701, Far East International Plaza, No. 317, Xianxia Road Shanghai, 200051, P.R.C. Tel: 86-21-6275-5700 Fax: 86-21-6275-5060 Microchip Technology Denmark ApS Regus Business Centre Lautrup hoj 1-3 Ballerup DK-2750 Denmark Tel: 45 4420 9895 Fax: 45 4420 9910 Hong Kong France Microchip Asia Pacific Unit 2101, Tower 2 Metroplaza 223 Hing Fong Road Kwai Fong, N.T., Hong Kong Tel: 852-2-401-1200 Fax: 852-2-401-3431 Arizona Microchip Technology SARL Parc d’Activite du Moulin de Massy 43 Rue du Saule Trapu Batiment A - ler Etage 91300 Massy, France Tel: 33-1-69-53-63-20 Fax: 33-1-69-30-90-79 India Germany Microchip Technology Inc. Two Prestige Place, Suite 150 Miamisburg, OH 45342 Tel: 937-291-1654 Fax: 937-291-9175 Microchip Technology Inc. India Liaison Office No. 6, Legacy, Convent Road Bangalore, 560 025, India Tel: 91-80-229-0061 Fax: 91-80-229-0062 Arizona Microchip Technology GmbH Gustav-Heinemann-Ring 125 D-81739 München, Germany Tel: 49-89-627-144 0 Fax: 49-89-627-144-44 Detroit Japan Microchip Technology Inc. Tri-Atria Office Building 32255 Northwestern Highway, Suite 190 Farmington Hills, MI 48334 Tel: 248-538-2250 Fax: 248-538-2260 Microchip Technology Intl. Inc. Benex S-1 6F 3-18-20, Shinyokohama Kohoku-Ku, Yokohama-shi Kanagawa, 222-0033, Japan Tel: 81-45-471- 6166 Fax: 81-45-471-6122 Arizona Microchip Technology SRL Centro Direzionale Colleoni Palazzo Taurus 1 V. Le Colleoni 1 20041 Agrate Brianza Milan, Italy Tel: 39-039-65791-1 Fax: 39-039-6899883 Microchip Technology Inc. 5 Mount Royal Avenue Marlborough, MA 01752 Tel: 508-480-9990 Fax: 508-480-8575 Chicago Microchip Technology Inc. 333 Pierce Road, Suite 180 Itasca, IL 60143 Tel: 630-285-0071 Fax: 630-285-0075 Dallas Microchip Technology Inc. 4570 Westgrove Drive, Suite 160 Addison, TX 75248 Tel: 972-818-7423 Fax: 972-818-2924 Dayton Los Angeles Microchip Technology Inc. 18201 Von Karman, Suite 1090 Irvine, CA 92612 Tel: 949-263-1888 Fax: 949-263-1338 New York Microchip Technology Inc. 150 Motor Parkway, Suite 202 Hauppauge, NY 11788 Tel: 631-273-5305 Fax: 631-273-5335 Korea Microchip Technology Korea 168-1, Youngbo Bldg. 3 Floor Samsung-Dong, Kangnam-Ku Seoul, Korea Tel: 82-2-554-7200 Fax: 82-2-558-5934 San Jose Microchip Technology Inc. 2107 North First Street, Suite 590 San Jose, CA 95131 Tel: 408-436-7950 Fax: 408-436-7955 All rights reserved. © 2000 Microchip Technology Incorporated. Printed in the USA. 4/00 Italy United Kingdom Arizona Microchip Technology Ltd. 505 Eskdale Road Winnersh Triangle Wokingham Berkshire, England RG41 5TU Tel: 44 118 921 5858 Fax: 44-118 921-5835 03/23/00 Microchip received QS-9000 quality system certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona in July 1999. The Company’s quality system processes and procedures are QS-9000 compliant for its PICmicro® 8-bit MCUs, KEELOQ® code hopping devices, Serial EEPROMs and microperipheral products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001 certified. Printed on recycled paper. Information contained in this publication regarding device applications and the like is intended through suggestion only and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. No representation or warranty is given and no liability is assumed by Microchip Technology Incorporated with respect to the accuracy or use of such information, or infringement of patents or other intellectual property rights arising from such use or otherwise. Use of Microchip’s products as critical components in life support systems is not authorized except with express written approval by Microchip. No licenses are conveyed, implicitly or otherwise, except as maybe explicitly expressed herein, under any intellectual property rights. The Microchip logo and name are registered trademarks of Microchip Technology Inc. in the U.S.A. and other countries. All rights reserved. All other trademarks mentioned herein are the property of their respective companies. DS00695A-page 12 2000 Microchip Technology Inc.