AN695

AN695
Interfacing Pressure Sensors to Microchip’s Analog
Peripherals
Author:
This application note will concentrate on the signal conditioning path of the piezoresistive sensing element
from sensor to microcontroller. It will show how the
electrical output of this sensor can be gained, filtered
and digitized in order to ready it for the microcontroller’s
calibration routines. This theoretical discussion will be
followed with a specific pressure sensing design that is
specifically designed to measure barometric pressure.
Bonnie Baker
Microchip Technology Inc.
INTRODUCTION
Pressure measurement devices can be classified into
two groups: those where pressure is the only source of
power and those that require electrical excitation. The
mechanical style devices that are only excited by pressure, such as bellows, diaphragms, bourdons, tubes or
manometers, are usually suitable for purely mechanical systems. With these devices a change in pressure
will initiate a mechanical reaction, such as a change in
the position of mechanical arm or the level of liquid in a
tube.
PIEZORESISTIVE PRESSURE
SENSORS
The piezoresistive is a solid state, monolithic sensor
that is fabricated using silicon processing. Piezo means
pressure, resistance means opposition to a DC current
flow. Since piezoresistive pressure sensors are fabricated on a wafer, 300 to 500 sensors can be produced
per wafer. Since these wafers generate a large number
of sensors they are available on the market at a
reduced cost as compared to mechanical sensors.
Electrically excited pressure sensors are most synergistic with the microcontroller environment. These style
of sensors can be piezoresistive, Linear Variable Differential Transformers (LVDT), or capacitive sensors.
Most typically, the piezoresistive sensor is used when
measuring pressure.
Voltage or
Current
Excitation
RS1
Voltage or
Current
Excitation
RS1
RS2
RS2
VOUTRS4
RS3
VOUT+
(a.) single element bridge
VOUTRS4
RS3
VOUT+
(b.) two element bridge
Dielectric
Voltage or
Current
Excitation
RS1
Si-P
RS2
VOUTRS4
RS3
Contact
Contact
Silicon
Substrate
VOUT+
(c.) four element or full bridge
Si-N
Diaphragm
(d.) single side of a sandwiched
piezoelectric pressure sensor
Figure 1: The resistive wheatstone bridge configuration can have one variable element (a.), two elements that vary
with excitation (b.) or four elements (c.). The piezoresistive pressure sensing element is usually a four element bridge
and is constructed in silicon (d.).
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Preliminary
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AN695
With this sensor, the resistors are arranged in a full
wheatstone bridge configuration, which has improved
sensitivity as compared to a single element or two element sensors (see Figure 1.d). When a positive differential pressure is applied to the four element bridge,
two of the elements respond by compressing and the
other two change to a tension state. When a negative
differential pressure is applied to the sensor, the diaphragm is strained in the opposite direction and the
resistors that were compressed go into a tension state,
while the resistors that were in a tension state change
into a compression state. Piezoresistive pressure sensors may or may not have an internal pressure reference. If they do, a pressure reference cavity is
generally fabricated by sealing two wafers together.
The top side of this fabricated sensor is the resistive
material and the bottom is the diaphragm.
The high side of the piezoresistive bridges shown in
Figure 1 can have a voltage excitation or current excitation applied. Although the magnitude of excitation
(whether it is voltage or current) effects the dynamic
range of the output of the sensor, the maximum difference between VOUT+ and VOUT- generally ranges from
10s of millivolts to several hundred millivolts. The electronics that follow the sensor are used to change the
differential output signal to single ended as well as gain
and filter it in preparation for digitization.
ELECTRONICS SIGNAL PATH
There are several ways of capturing the small differential output signal of the sensor and transforming it into
a usable digital code. One approach that can be taken
is shown in the block diagram in Figure 2.a. With this
approach, the small differential output of the bridge is
gained and converted from differential to single ended
with an instrumentation amplifier (IA). The signal may
or may not travel through a multiplexer. The signal then
passes through a low pass filter. The low pass filter
eliminates out-of-band noise and unwanted frequencies in the system before the A/D conversion is performed. This is followed by a stand-alone A/D converter
which transforms the analog signal into a usable digital
code. The microcontroller takes the converter code,
further calibrates and translates if need be for display
purposes. In this signal path only one analog filter is
required and it is positioned at the output of the multiplexer.
The second signal path shown in Figure 2.b also has
an instrumentation amplifier (IA) in the signal path. Following the instrumentation amplifier stage the signal is
filtered in the analog domain and then digitized with an
on-chip microcontroller’s A/D converter. When this type
of signal path is used, every signal going into the multiplexer will require its own analog filter. Additionally,
the accuracy and speed of the converter in the microcontroller is less than a stand-alone A/D converter. This
may or may not be an issue in a particular application.
IA
(a.)
MUX
FILTER
SAR
A/D
MicroController
REF
(b.)
IA
FILTER
MUX
A/D
µC
VREF
(c.)
MUX
Low
Pass
Filter
COMP
µC
VREF
Figure 2: Three block diagrams for the piezoresistive pressure sensor signal conditioning path are shown in this
Figure. The top two block diagrams, a. and b., are discussed in detail in this application note. The bottom block
diagram (c.) is discussed in detail in AN717 (Microchip Technology Inc.).
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Preliminary
 2000 Microchip Technology Inc.
AN695
INSTRUMENTATION AMPLIFIER
OPTIONS AND DESIGN
ational amplifiers, thereby significantly reducing source
impedance mismatch problems at DC. The transfer
function of this circuit is equal to:
With this application, the two low voltage signals from
the bridge need to be subtracted in order to produce a
single ended output signal. The results of this subtraction also need to be gained so that it matches the input
range of the A/D converter. The implementation of the
subtraction and gain functions are done so that the
sensor signal is not contaminated with additional
errors. The instrumentation amplifier circuits shown in
Figure 3 and Figure 4 achieve all of these goals. Both
of these configurations take two opposing input signals,
subtract them and apply gain. The subtraction process
inherently rejects common-mode voltages. Combined
with these functions the signal is level shifted, making
it synergistic with the signal supply environment.
R4
1 R2 R3
V OUT = ( V IN+ – V IN- ) -------  1 + ---  ------- + ------- 
R3 
2  R1 R4 
R2 + R3
+  --------------------   + V CM
RG
4  R3 R 2  
R
- ------- – ------- + V REF
 -----R 3  R4 R 1  
It should be noted from this transfer function that the
input signals are gained along with the common-mode
voltage of the two signals. The common-mode voltage
can be rejected when R1 = R4 and R2 = R3. Given this
change the transfer function becomes:
R 1 2R1
VOUT = ( V IN+ – V IN- )  1 + ------- + ----------  + VREF
R2 RG
The Two Op Amp Instrumentation Amplifier
The common-mode rejection error that is caused by
resistor mismatch is equal to:
A solution to the circuit problem discussed above is
shown in Figure 3. The circuit in Figure 3 uses two
operational amplifiers and five resistors to solve this
gain and subtraction problem.
R2


1 + ------R1


CMR = 100*  ---------------------------------------------------------- 
(
%
of
mismatch
error
)




Dual amplifiers are usually used in this discrete design
because of their good matching of bandwidth and over
temperature performance. This instrumentation amplifier design uses the high impedance inputs of the oper-
RG
R2
R4
VDD
-
VREF
R1
R3
A1
MCP602
VIN-
A2
+
MCP602
+
VIN+
R 1 2R 1
4 + 60kΩ
VOUT =  1 + ------- + ----------  ( V IN+ – V IN- ) + VREF =  ------------------------  ( VIN+ – VIN- ) + VREF

 RG 
R 2 RG 
Where R1 = 30kΩ and R2 = 10kΩ
Figure 3: The two op amp instrumentation amplifier takes the difference of two input signals, gains that difference,
while rejecting any voltage that is common to both of the input signals.
 2000 Microchip Technology Inc.
Preliminary
DS00695A-page 3
AN695
The ac common mode rejection for this configuration is
poor. This is due to the fact that the common mode signal at VIN- is inverted once with A1 and then it travels
through A2 causing a second propagation delay. The
common mode signal at VIN+ only travels through one
operational amplifier (A2). Additionally, the two operational amplifiers have different closed loop gains, and
consequently different closed loop bandwidths.
The second factor that limits the common-mode input
range of this circuit comes from the input swing restrictions of the amplifiers themselves.
If this circuit is in a single supply environment, it will typically require a reference that is centered at the common-mode voltage of the input signals. In Figure 4,
VREF serves that function. This voltage can be implemented discretely with a precision reference chip as
shown in Figure 4.a or with two equal resistors in series
between the power supply as shown in Figure 4.b.
In terms of common-mode input range, there are two
factors that limit the range of this instrumentation amplifier. The first factor involves the operation of A1 as it
responds to the VIN- and VIN+ input signals and the voltage reference, VREF. The signal at the non-inverting
input to A1 and A2 gained by the output of A1 by:
V OUT – A1 = V IN-
Another added benefit to matching R2/R1 = R3/R4 is
that the gain of the circuit can be changed with one
resistor, RG.
This instrumentation amplifier circuit has high impedance inputs and programmable gain capability. The
features that could be improved in this circuit solution is
to have the common-mode rejection independent of
gain and better over frequency. These performance
characteristics can only be obtained by an instrumentation amplifier configuration that has three operational
amplifiers.
( R G R 2 + R 1 R 2 + R 1 R G )
 ---------------------------------------------------------------

( R 1 R2 )
R2
R2
– V IN+  --------  – V REF  ------- 
 RG 
 R1 
R1 = R1A || R1B
VDD
VDD
VREF
Precision Voltage
Reference
R1A
R2
R1B
RG
R1
VDD
-
VREF
OR
VIN-
R2
MCP602
-
+
MCP602
+
VOUT
VIN+
(a)
(b)
Figure 4: The reference voltage for a two op amp instrumentation amplifier in a single supply environment can be
implemented with a stand-alone voltage reference (a) or a resistor divider across a voltage reference or the supply
voltage (b).
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Preliminary
 2000 Microchip Technology Inc.
AN695
all gains as long as the signals stay within A1 and A2
input and output head room limitations. If the common
mode errors of the input amplifiers track they will be
cancelled by the output stage.
The Three Op Amp Instrumentation Amplifier
An example of a more versatile instrumentation amplifier configuration is shown in Figure 5.
With this circuit configuration, two of the three amplifiers (A1 and A2) gain the two input signals. The third
amplifier, A3, is used to subtract the two gained input
signals, thereby providing a single ended output. The
transfer function of this circuit is equal to:
If the assumption that R1/R2 equals R3/R4 is not correct, there could be a noticeable common mode voltage error. The calculated common-mode rejection
(CMR) error that is attributed to resistor mismatches in
this circuit is equal to:
2R F2
R1 + R2
V OUT = VIN+  1 + --------------  R 4  ----------------------------------  –

R G   ( R 3 + R 4 )R 1 
R2


1 + ------R1


CMR = 100*  ---------------------------------------------------------- 
(
%
of
mismatch
error
)




R1 + R2
2R F1 R 2
V IN-  1 + --------------   -------  + V REF R 3  ---------------------------------- 
 ( R 3 + R 4 )R 1 

RG   R1 
for R 1 = R 3 and R 2 = R 4
If RF2 = RF1, R1 = R3, and R2 = R4, this formula can be
simplified to equal:
2R F
VOUT = ( V IN+ – V IN- )  1 + -----------  + V REF

RG 
An example of the impact of this error is demonstrated
with a 12-bit, 5V system, where the gain of the circuit is
100V/V, the common-mode voltage ranges 0 to 5V and
the matching error can be as large as ±1%. Using the
formula above, the contributed error of this type of common-mode excursion is equal to 1mV. This voltage is
slightly less than 1LSB.
Quad amplifiers are typically used in the three op-amp
instrumentation amplifier discrete designs because of
the matching qualities of amplifiers with the same silicon. In contrast to the two op-amp instrumentation
amplifier, the input signal paths (at VIN+ and VIN-) are
completely balanced. This is achieved by sending VIN+
and VIN- signals through the same number of amplifiers
to the output and using a common gain resistor, RG.
In a single supply environment, the voltage reference
should be equal to the center of the input signals. This
voltage is represented in the circuit in Figure 5 as VREF.
The purpose and effects of this reference voltage is to
simply shift the output signal into the linear region of the
amplifier.
Since this input stage is balanced, common mode currents will not flow through RG. The common-mode
rejection of this circuit is primarily dependent on the
resistor matching around A3. When R1 = R2 = R3 = R4,
common mode signals will be gained by a factor of one
regardless of gain of the front end of the circuit. Consequently, large common mode signals can be handled at
VIN-
+
A1
MCP604
R2
RF1
R1
A3
RG
MCP604
RF2
R3
A2
VOUT
R4
MCP604
VIN+
+
+
VREF
Where RF1 = RF2 and R1 = R2 = R3 = R4
2R F
VOUT =  1 + ----------- ( V IN+ – V IN- ) + VREF
RG
Figure 5:
This is a three op amp implementation of an instrumentation amplifier.
 2000 Microchip Technology Inc.
Preliminary
DS00695A-page 5
AN695
R2
From the
output of
A1
Precision Voltage
Reference
VDD
From the
output of
A2
R1
-
A3
R1
MCP604
+
VOUT
VDD
VSHIFT
OR
VSHIFT
R2A
R2B
R2 = R2A || R2B
(a)
(b)
Figure 6: The reference voltage in Figure 5 can be implemented by using a precision reference circuit (a.) or a
resistive voltage divider circuit (b.).
The VREF circuit function can be implemented with a
precision voltage reference or with the resistive network shown in Figure 6.
ANALOG FILTERING
A big topic for debate in digital design circles is whether
or not an analog filter is needed and more importantly,
can a digital filter replace the analog filter.
A common assumption with designers that are trying to
tackle analog challenges of this type is that they claim
that they are only measuring DC so they don’t have to
worry about filtering. Unfortunately, the noise generators in the electronics and the environment do not have
the “intelligence” to accommodate the designer’s
desires. Consequently, if a filter is not included, the circuit will be surprisingly noisier than anticipated.
Once it is accepted that a filter is required, the next
debate that ensues is whether the filtering strategy
should be analog, digital or both.
These signals are usually unintentional, but almost
always destructive if not controlled. On the down side,
analog filters can add to the noise floor particularly if a
noisy amplifier is used with a large gain.
Where analog filters earn their worth by rejecting noise
in the out of band region, digital filters can be utilized to
reduce the in-band noise floor. This is implemented
with oversampling algorithms. These types of filters are
much easier to change on the fly because it is a matter
of programming instead of a matter of changing resistors and capacitors as it is with analog filters. With all of
these benefits there is a price to pay in terms of
response time. Digital filters must collect a certain
amount of conversion data before calculations can be
performed. The digital filter algorithms tend to slow
down the response time as well as delay the output. If
real time responses are not critical, the digital filter disadvantages are not detrimental to the operation of circuit.
A common assumption that is made by programmers is
that they can eliminate all ills with digital filtering. To
some extent this is true, however, it is at a high price of
time and memory and truthfully, it may not be possible
to succeed.
Analog filtering removes a considerable amount of
headaches for the programmer from the start. Analog
filters have their place in circuit designs as do digital filters. For instance, analog filters will eliminate aliasing
errors that will occur through the A/D conversion process if they are allowed to go through. Once these
errors are allowed in the conversion it is impossible to
discriminate good signal from aliased signal in the digital domain. The analog filter also removes large signal
noise that is generated by spikes or spurs in the signal.
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Preliminary
 2000 Microchip Technology Inc.
AN695
C1
R2
VIN
R1
C2
R3
+
MCP
606
-
VOUT
R4
Figure 7: By using FilterLabTM software, this 2nd order low pass filter that has a non-inverting gain in the pass band
can be configured as a Butterworth, Bessel, or Chebyshev filter.
As discussed previously, the hardware implementation
of a low pass filter at its most fundamental level
requires a capacitor and resistor for each pole. Active
filters, which have one amplifier for every two poles,
have the added benefit of preventing conflicting impedances and degrading the signal path.
Close inspection of this filter shows that the circuit can
be configured in a gain of +1V/V by shorting R4 and
opening R3. In this configuration it is likely that the input
of the amplifier will be exercised across a full rail-to-rail
input range.
The second order Multiple Feedback circuit implementation of a low pass filter uses an amplifier, three resistors and two capacitors, as is shown in Figure 8. The
DC gain of this filter is negative and easily adjusted with
the ratio of R3 and R1. When used in a single supply
environment, this circuit usually needs a voltage reference on the non-inverting input of the amplifier.
The 2nd order lower pass filter shown in Figure 7 is one
of a class of circuits that were described in 1995 by R.P
Sallen and E.L. Key. With this filter the DC gain is positive. In a single supply environment this eases the
implementation considerably, because a mid-supply
reference is not required. This circuit not only filters
high frequencies, but it can be used to gain the incoming signal.
This is the filter circuit that will be used in a barometric
pressure application. An adjustable voltage reference
will be Included in this filter design.
R3
VIN
R1
R2
C2
C1
MCP
606
+
VOUT
VREF
Figure 8: By using FilterLabTM software, this 2nd order low pass filter that has an inverting gain in the
pass band can be configured as a Butterworth, Bessel, or Chebyshev filter.
 2000 Microchip Technology Inc.
Preliminary
DS00695A-page 7
AN695
BAROMETRIC PRESSURE SENSING
Parameter (w/ 5V excitation)
Specification
The considerations for the design of a barometric sensing system encompasses altitude and resolution. The
expected altitude that our sensor will be placed in is
approximately from sea level to 20,000 ft. The nominal
pressure at sea level is 14.7 psi and the nominal pressure at 20,000 ft. is 6.75 psi. The difference in pressure
between these two altitudes is 7.95 psi. With this range,
the appropriate pressure sensor should be an absolute
version that is referenced to an on-chip vacuum and
have a range up to 15psi. Since the change in pressure
for major weather changes is approximately 0.18 psi, a
resolution of 0.015 psi is over ten times more accurate
than the measured value. The circuit that will be used
for this design discussion is shown in Figure 9.
Operating Pressure Range
0 to 15 psi
Sensitivity
6.0mV/psi (typ)
Full-Scale Span
90mV (typ)
Zero Pressure Offset
± 300µV
Temperature Effect on Span
(0 to 70°C)
± 0.5% FSO
Temperature Effect on Offset
(0 to 70°C)
± 500µV
Table 1: The specifications of the SCX015 from
SenSym indicates that this is a good pressure
sensor that can be used to measure barometric
pressure.
The specifications of the SCX015 from SenSym indicates that this is a good pressure sensor that can be
used to measure barometric pressure.
The critical pressure sensor specifications for this
application include the operating pressure range, sensitivity, room temperature (25°C) span and offset errors
as well as over temperature (see Table 1). Although,
the range of this sensor extends from 0 psi to 15 psi,
this application will not be using that lower range. The
minimum differential output voltage from the sensor will
be 40.5mV (6.75psi or 20,000 ft.) and the maximum
sensor voltage will be 88.2mV (14.7psi or sea level).
The voltage at the output of the sensor is gained before
it is digitized using an instrumentation amplifier.
Note that temperature issues are beyond the scope of
this application note. Detailed information about temperature sensing circuits can be found in Microchip’s
AN679, AN684, AN685, and AN687.
Instrumentation Amplifier
RG
R2
2nd Order Butterworth
Low Pass Filter
1/2
R1
R2
R2
R4
R2
MCP602
+
VDD = 5V
R1
VDD
A1
R1
1/2
MCP602
+
A2
R3
C1
R5
C2
SCX
015
1
MCP
606
+
2
A3
3
8
SCLK
MCP3201
DOUT
CS
A4
4
R1 = 30kΩ
R2 = 10kΩ
RG = 1.15 kΩ
R3 = 95.7 kΩ
R4 = 172 kΩ
R5 = 304 kΩ
C1 = 0.22 µF
C2 = 0.22 µF
A1 = A2 = A3 = Single Supply, CMOS op amp
A4 = 12-bit, A/D SAR Converter
35.7kΩ
SCK
SI
CS
PIC16C6xx
R1
A6
A5
10kΩ
68.1kΩ
Level Shift Voltage
and Offset Adjust
A4 and A6 can be
replaced with a
PICmicro that has
an on-chip A/D
converter
A5 = 10kΩ Digital Potentiometer
Figure 9: The voltage at the output of the SCX015 pressure sensor is gained by the instrumentation amplifier (A1
and A2) then filtered, gained and level shifted (A4) with a 2nd order low pass filter (A3) and digitized with a 12-bit A/D
converter (A5).
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Preliminary
 2000 Microchip Technology Inc.
AN695
Instrumentation Amplifier Design
This sensor requires voltage excitation. In order to
determine the required gain of the circuit in Figure 9 the
relationship between the maximum sensor output and
allowable instrumentation amplifier output is used in
the calculation. As stated previously, the maximum differential output of the sensor is 88.2mV. The allowable
output range of the instrumentation amplifier is equal to
VDD - 100mV. In a five volt system where VDD = 5V, the
amplifier output maximum is equal to 4.9V. The minimum output of the sensor is 40.5mV. Since this is a
positive voltage and the instrumentation amplifier is in
a single supply environment, this minimum sensor output voltage will not drive the output of the instrumentation amplifier below ground. Consequently, the
reference voltage called out in Figure 3 and Figure 4 is
made to be equal to ground.
Gain is calculated by dividing the maximum output voltage with the maximum input voltage. Using this calculation, the appropriate gain for our system is 55.6V/V.
By using the gain formula in Figure 5:
2R 1
R G = ---------------------------------------R 1
 Gain – 1 – -----
R 2
The gain and offset adjust features of this filter are also
used in this segment of the application circuit. Given
that the output from the instrumentation amplifier is
2.3V to 4.9V, the peak-to-peak voltage of this signal is
2.6V. A gain of 1.8V/V will produce an output swing of
approximately 4.7V peak-to-peak. The adjustable offset voltage of this circuit which is gained by 2.8V/V will
be configured to insure that the signal will fall at the output of the amplifier between the supplies. This adjustment circuit can also be used to remove system offset
errors that originate in the sensor or instrumentation
amplifier.
The filter circuit in Figure 9 can be designed with the
FilterLabTM software from Microchip Technology. The
two capacitors are adjusted using the FilterLab program to be equal to 0.22uF. This adjustment is made in
order to keep the capacitor packages small enough so
surface mount capacitors can be used.
If R 1 = 30kΩ and R 2 = 10kΩ,
60kΩ
R G = -------------------------( 55.6 – 4 )
= 1.15kΩ (closest 1% value)
With this gain, the maximum output of the IA will be
88.2mV*55.6V/V or 4.9V and the minimum output will
be 40.5mV*55.6V/V or 2.3V.
Since the gain of this instrumentation amplifier stage is
relatively large, it is desirable to use an amplifier that
has a low offset voltage. The MCP607, dual CMOS
amplifier has a guaranteed input offset voltage of
250µV (max). This amplifier’s low quiescent current of
25µA (max) make this device attractive for battery powered applications.
The offset adjust of the filter circuit is implemented with
a 10kΩ digital potentiometer in series with a 68.1kΩ
and 35.7kΩ resistors. The range of the offset adjust
portion of this circuit at the wiper of the digital potentiometer is from 3.0V to 3.4V. This offset circuitry is
gained by the filter/amplifier circuit so that the nominal
value of the offset circuitry in combination with the sensor signal is equal to:
VOUT-FILTER = -1.8V/V (Nominal Input Signal)
+ 2.8V/V (Nominal Reference voltage)
VOUT-FILTER = -1.8V/V (3.6V) + 2.8V/V (3.2V)
VOUT-FILTER = 2.48V
A key amplifier specification for this filtering circuit is
input voltage noise. The MCP601, single CMOS amplifier has a typical noise density of 29 nV/√Hz @ 1kHz.
A/D Converter Design
Filter Design
Now that the signal from the pressure sensor has been
properly differentiated and gained, noise is removed to
making the results from the 12-bit A/D conversion
repeatable and reliable. Remember that the output of
the instrumentation amplifier circuit does not swing a
full 0V to 5V. Consequently, the filter stage will also be
used to implement a second gain cell as well as offset
adjust.
The stop frequency of this filter is 60Hz. This will
removes any mains frequencies that may be aliased
back into the signal path during conversion. This being
the case, the cut-off frequency is selected to be 10Hz.
 2000 Microchip Technology Inc.
Any cut-off frequency lower than 10Hz, requires capacitors that are too large, making the board implementation awkward. The total attenuation between 10Hz and
60Hz is approximately -30dB. In other words, a 60Hz
signal that is part of the output signal of the instrumentation amplifier is attenuated by 0.031 times. Keeping in
mind that the instrumentation amplifier has already
rejected a major portion of any 60Hz common-mode
signal, this level of attenuation is enough to remove any
remaining 60Hz noise that exists in the signal path.
The final design step for this analog signal path is to
insert the analog-to digital converter. The converter
quantizes a continuous analog signal into discrete
buckets. The appropriate converter can be selected
once it is determined how many bits the application
requires.
The range of the analog signal has been closely
matched to the input range of a zero to 5V in A/D converter.
The barometric pressure range is 14.7 psi to 6.75 psi.
The expected increase from good weather to a strong
storm system would be approximately 0.18 psi. Given
Preliminary
DS00695A-page 9
AN695
this, the equipment should resolve to at least 0.015psi.
This is easily achieved with a 10-bit converter. If resolution to 0.002 is needed a 12-bit a/d converter would
be more suitable.
Microchip has a large variety of analog to digital converters that can be used for this application. If the
stand-alone solution is appealing, the MPC320X family
of 12-bit and the MCP300X 10-bit family of converters
are available.
Generally speaking, stand-alone A/D converters have
better accuracy than those compared to on-board converters. They also have features such a pseudo differential inputs and faster conversion speeds. The pseudo
differential capability of these devices allow for configurations that reject small common mode signals. Additionally, the single channel devices can be used in
simultaneous sampling applications, such as motor
control. The application circuits using the singe converter also require fewer analog filters because the
multiplexer is typically placed before the anti-aliasing
filter.
If an on-board a/d converter fits the application better,
the PICmicro line has a large array of converters combined with other peripherals on a variety of micros that
can be used.
The integrated solution offers a degree of flexibility that
the stand-alone solution does not have. This flexibility
comes in the form of operational flexibility where the
device’s voltage reference and sampling speed can be
reconfigured on the fly. The I/O configuration is also
very flexible allowing for easy implementation of the
board layout.
The stand-alone and integrated A/D converters from
Microchip are both suitable for the pressure sensor circuit that is shown in Figure 9.
CONCLUSION
The design challenge that has been tackled in this
application note is gaining, filtering, and digitizing the
small differential signal of a pressure sensor bridge. In
order to achieve this goal, we used a two-op amp
instrumentation amplifier which gained the differential
signal from the pressure sensor and converted it to a
signal ended output. After this gain stage, a 2nd order,
Butterworth, anti-aliasing filter was used to reduce
noise so that the A/D converter could achieve a full 10bit accuracy.
The suggested A/D converter strategy could be on
board or off board and the trade-offs were presented.
Digital filtering was not needed in this application.
REFERENCES
Tandeske, Duane, Pressure Sensors, Marcel Dekker,
Inc., 1991
“Anti-Aliasing Analog Filters for Data Acquisition Systems”, Baker, Bonnie C., AN699, Microchip Technology
Inc.
“Making a Delta-Sigma converter with a Microcontroller”, Baker, Peter, Darmawaskita, Butler, AN700, Microchip Technology, Inc.
“Using Operational Amplifiers for Analog Gain in
Embedded System Design”, Baker, Bonnie C., AN682,
Microchip Technology, Inc.
“Building a 10-bit Bridge Sensing Circuit using the
PIC16C6XX and MCP601 Operational Amplifier”,
Baker, Bonnie C., AN717, Microchip Technology, Inc.
“Precision Temperature Sensing with RTD Circuits”,
Baker, Bonnie C., AN687, Microchip Technology, Inc.
“Temperature Sensing Technologies”, Baker, Bonnie
C., AN679, Microchip Technology, Inc.
“Thermistors in Single Supply Temperature Sensing
Circuits”, Baker, Bonnie C., AN685, Microchip Technology, Inc.
“Single Supply Temperature Sensing with Thermocouples”, Baker, Bonnie C., AN684, Microchip Technology,
Inc.
DS00695A-page 10
Preliminary
 2000 Microchip Technology Inc.
AN695
NOTES:
 2000 Microchip Technology Inc.
Preliminary
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