DN18 - A Battery Powered Lap Top Computer Power Supply

A Battery Powered Lap Top Computer Power Supply
Design Note 18
Brian Huffman
Most battery powered lap top computers require regulated multiple output potentials. Problems associated
with such a supply include magnetic and snubber design,
loop compensation, short circuit protection, size and
efficiency. Typical output power requirements include
5V @ 1A for memory and logic circuitry and ±12V
@ 300mA to drive the analog components. Primary
power may be either a 6V or 12V battery. The circuit
in Figure 1 meets all these requirements. The LT®1071
simplifies the power supply design by integrating most
of the switching regulator building blocks. Also, the
off-the-shelf transformer eliminates all the headaches
associated with the magnetic design.
The circuit is a basic flyback regulator. The transformer
transfers the energy from the 12V input to the 5V and
±12V outputs. Figure 2 shows the voltage (trace A) and
the current (trace B) waveforms at the VSW pin. The
VSW output is a collector of a common emitter NPN,
so current flows through it when it is low. The circuit’s
40kHz repetition rate is set by the LT1071’s internal
oscillator. During the VSW (trace A) “on” time, the input
voltage is applied across the primary winding. Notice
MUR120
12VIN
+
L1
4
22μF
•
2k
2W
0.2μF
+
7
LT1086-12
12V
+
470μF
10μF
8
2
9
3
10
that the current in the primary (trace C) rises slowly
as the magnetic field builds up. The magnetic field in
the core induces a voltage on the secondary windings.
This voltage is proportional to the input voltage times
the turns ratio. However, no power is transferred to
the outputs because the catch diodes are all reversed
biased. The energy is stored in the magnetic field. The
amount of energy stored in the magnetic field is a function of the current level, how long the current flows, the
primary inductance and the core material. When the
switch is turned “off” energy is no longer transferred
to the core, causing the magnetic field to collapse. The
voltage on the transformer windings is proportional to
time-rate-of-change of the magnetic field. Hence, the
collapsing magnetic field causes the voltages on the
windings to change. Now the catch diodes are forward
biased and the energy is transferred to the outputs.
Trace D is the voltage seen on the 5V secondary and
trace E is the current flowing through it. The energy
transfer is controlled by the LT1071’s internal error
amplifier, which acts to force the feedback (FB) pin to
a 1.24V reference. The error amplifiers high impedance
output (VC pin) uses an RC damper for stable loop
compensation. If a 6V input is desired, use just one
primary winding and an LT1070.
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respective owners.
MUR120
LT1086-12
•
MUR120
+
+
470μF
10μF
•
1
VIN
11
–12V
VSW
MUR360
LT1071
•
5
2700μF
FB
GND
VC
+
VOUT
5V
1A
3.01k*
6
A = 50V/DIV
LT1070 SWITCH VOLTAGE
B = 2A/DIV
LT1070 SWITCH CURRENT
C = 2A/DIV
PRIMARY CURRENT
D = 10V/DIV
5V SECONDARY VOLTAGE
E = 4A/DIV
5V SECONDARY CURRENT
F = 50V/DIV
LT1070 SWITCH VOLTAGE
G = 2A/DIV
1k
SNUBBER DIODE CURRENT
+
DN018 F02
1μF
L1 = PULSE ENGINEERING, INC. #PE-65108
PULSE ENGINEERING, INC.
P.O. BOX 12235
SAN DIEGO, CA 92112
PHONE: 619-268-2400
* = 1% FILM RESISTOR
Figure 1. Multi-Output Flyback Converter
11/88/18_conv
1.00k*
A, B, C, D, E HORIZ = 5μs/DIV
F, G HORIZ = 1μs/DIV
DN018 F01
Use LT1171 and LT3080 for Higher Efficiency
Figure 2. Waveforms for Continuous Mode Operation
OPTIONAL FOR LOWEST
IQ CURRENT
OFF
MUR120
L1
•
ON
7
12V
LT1086-12
+
+
470μF
10μF
8
9
MUR120
12VIN
LT1086-12
+
4
22μF
•
2k
2W
0.2μF
10
+
+
470μF
10μF
11
2
MUR360
3
•
MUR120
+
5
C1
2700μF
R3
1M
R1
1M*
R7
10k
–12V
VOUT
5V
1A
•
VIN
1
VSW
6
10pF
C3
0.005
LT1071
12VIN
VC
+
C2
47μF
74C04
(5/6)
74C04
(1/6)
A1
1/2 LT1017
R6
200Ω
+
GND
NC
–
FB
1N4148
1N4148
L1 = PULSE ENGINEERING, INC. #PE-65108
* = 1% FILM RESISTOR
R5
180k
R2
453k*
12VIN
LT1004
1.2
DN018 F03
Figure 3. Multi-Output, Transformer Coupled Low Quiescent Current Converter
This is not an ideal transformer so not all the energy
is coupled into the secondary. The energy left in the
primary winding causes the overvoltage spike seen on
the VSW pin (trace F). This phenomenon is modeled
by a leakage inductance term placed in series with
the primary winding. When the switch is turned “off”
current continues to flow in the inductor, causing the
snubber diode to conduct (trace G). The snubber network clamps the voltage spike, preventing excessive
voltage at the LT1071’s VSW pin. When the snubber
diode current reaches zero, the VSW pin voltage settles
to a potential related to the turns ratio, output voltage
and input voltage.
Post regulators are needed to the unregulated outputs if
the cross regulation error is too great. Such error can be
as much as 20% depending upon output loading conditions. Note that the floating secondaries allow a –12V
output to be obtained with a positive voltage regulator.
The isolation allows the input of the regulator to float
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above ground. The LT1086 positive voltage regulators
maintain both positive and negative outputs with 1%.
If battery capacity is limited by size or weight this
circuits 9mA quiescent current may be too high. Figure 3’s modification offers output current in the ampere
range with only microamps of quiescent drain. Further
information about this circuit can be found in the LTC
Application Note AN29 “Some Thoughts on DC-DC
Converters, “ page 8.
By using standard magnetics and a simplified switching regulator the design time needed to implement this
power supply is greatly reduced. Although these circuits
demonstrated in flyback topology, the LT1070/LT1071/
LT1072 can easily handle other configurations including
buck, boost, forward and inverting. Examples are given
in the LTC Application Notes; AN19 “LT1070 Design
Manual, “ AN25 “Switching Regulators for Poets,” and
AN29 “Some Thoughts on DC-DC Converters.”
For applications help,
call (408) 432-1900
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