cd00190357

AN2738
Application note
L6390 half-bridge gate driver
Introduction
The L6390 is a versatile high voltage gate driver IC which is particularly suited for field
oriented control (FOC) motor driving applications. It simplifies the design of control systems
for a wide range of motor applications such as home appliances, industrial drives, DC
motors and fans.
Designed using BCD off-line technology, this device is capable of operating with voltage rails
up to 600 V. The gate driver provides all the functions and current capability necessary for
high side and low side power MOSFET and IGBT driving.
The L639x series are high voltage half-bridge gate drivers. These devices can be used in all
applications where high voltage shifted control is necessary. The devices have a driver
current capability best suited for home appliance motor driving ratings, and they are also
equipped with patented internal circuitry which replaces the external bootstrap diode. This
feature is achieved by means of a high voltage DMOS synchronously driven with the low
side gate driver.
The L6390 is a half-bridge driver with several functions such as externally adjustable deadtime, interlocking, smart shutdown (patented), fault comparator and a dedicated high
performance op-amp for advanced current sensing. The outputs can be driven by two
dedicated logic signals or, alternatively, only one logic signal by connecting the two inputs
together. The device is available in the DIP16 or SO16 packages.
Figure 1.
L6390 application block diagram
VCC VCC
BOOTSTRAP DRIVER
4
16
FLOATING STRUCTURE
BOOT
+
from LVG
UV
DETECTION
UV
DETECTION
FROM CONTROLLER
HIN
H.V.
3
S
LEVEL
SHIFTER
SHOOT
THROUGH
PREVENTION
LIN
GND
14
OUT
TO LOAD
VCC
SD/OD
HVG
1
VBIAS
FROM/TO
CONTROLLER
15
R
LOGIC
5V
FROM CONTROLLER
2
8
Cboot
HVG
DRIVER
LVG
11
SD
LATCH
SMART
SD
LVG
DRIVER
5V
COMPARATOR
10
+
CP+
+
VBIAS
VREF
DT
5
DEAD
VCC
TIME
OPOUT
OPAMP
7
+
9
6
TO ADC
August 2009
OP+
OP-
-
Doc ID 14589 Rev 2
1/54
www.st.com
Contents
AN2738
Contents
1
Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2
Logic inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
3
UVLO function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
4
Dead time and interlocking function management . . . . . . . . . . . . . . . . 8
5
Smart shutdown function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
6
L6390 op-amp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
7
Bootstrap driver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
7.1
8
CBOOT selection and charging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Application example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
8.1
VCC supply pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
8.2
BOOT (floating) supply pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
8.3
Logic input pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.4
Shutdown pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.5
Dead time pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.6
Op-amp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.7
Comparator input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.8
Sense resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.9
Gate driver outputs: gate lines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.10
Gate driving: principle of working with inductive load . . . . . . . . . . . . . . . . 25
9
Induced turn-on phenomenon . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
10
How to increase the gate driver output current capability . . . . . . . . . 37
11
The below-ground voltage on the OUT pin . . . . . . . . . . . . . . . . . . . . . . 39
2/54
11.1
The below-ground voltage phenomenon . . . . . . . . . . . . . . . . . . . . . . . . . 39
11.2
How to reduce the below ground spike voltage . . . . . . . . . . . . . . . . . . . . 40
Doc ID 14589 Rev 2
AN2738
Contents
11.3
11.4
Issues related to the below-ground voltage phenomenon . . . . . . . . . . . . 43
11.3.1
VBOOT voltage safe operating condition . . . . . . . . . . . . . . . . . . . . . . . . 43
11.3.2
Bootstrap capacitor over-charging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Functionality of L6390 outputs in below-ground condition . . . . . . . . . . . . 46
11.4.1
Steady state (DC) conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
11.4.2
Transient conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
11.4.3
Below-ground voltage spikes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
12
Layout suggestions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
13
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
Doc ID 14589 Rev 2
3/54
List of figures
AN2738
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Figure 33.
Figure 34.
Figure 35.
Figure 36.
Figure 37.
Figure 38.
on OUT pin
Figure 39.
Figure 40.
Figure 41.
Figure 42.
Figure 43.
Figure 44.
Figure 45.
Figure 46.
4/54
L6390 application block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Input configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
L6390 gate driver outputs in UVLO condition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Timing waveforms. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Typical dead time vs. DT resistor value . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Smart shutdown timing waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Smart shutdown equivalent circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Protection scheme . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Disable time vs. SD capacitance (typical values) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
3-phase system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
General advanced current sense scheme . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Advanced current sensing waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
L6390 op-amp, application example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Typical L6390 ideal output . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Detail on single PWM cycle in advanced current sensing for FOC systems . . . . . . . . . . . 17
Bootstrap driver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
External charge pump. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
3-phase drive- typical scheme . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Typical application schematic of a 3-phase FOC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Ex. of an application circuit for one of the three half-bridges of a 3-phase power stage . . 23
Layout suggestion for the gate driving circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Gate driver output: equivalent circuit for turn-on. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Gate driver output: equivalent circuit for turn-off. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Hard switching . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Soft switching . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Turn-on hard switching details with induction load: gate charge and plateau phase . . . . . 29
Total equivalent circuit for the turn-on . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Turn-off hard switching details with induction load: gate charge and plateau phase . . . . . 31
Total equivalent circuit for the turn-off . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Power dissipation during switching (approximation) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
RGATE dimensioning criteria . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Induced turn-on phenomenon - circuital description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Block diagram of output current capability enhancement using external current buffers . . 37
Example of a gate driving circuit with current buffers for current capability increasing. . . . 38
below-ground voltages in L6390 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
Transient peak forward voltage vs. dIF/dt of STTH1L06 diode . . . . . . . . . . . . . . . . . . . . . . 41
Use of OUT resistor to limit the below ground voltage spike on OUT pin. . . . . . . . . . . . . . 41
Use of combination of OUT resistor and OUT diode to limit the below ground voltage spike
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
Bootstrap over-charging due to below-ground voltage on OUT pin . . . . . . . . . . . . . . . . . . 44
Different bootstrap network characteristics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
L6390 safe operating range when the OUT pin is below ground voltage (in steady state). 46
Driver functionality in below-ground voltage condition on OUT pin . . . . . . . . . . . . . . . . . . 46
OUT below-ground voltage in transient conditions: limited boot over-charging . . . . . . . . . 48
Example of below-ground voltage spike . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
Layout suggestion for a 3-phase power system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
Layout example from the STEVAL-IHM021V1 3-phase board . . . . . . . . . . . . . . . . . . . . . . 52
Doc ID 14589 Rev 2
AN2738
1
Pin description
Pin description
Table 1.
Pin description
Pin n°
Pin name
Type
1
LIN
I
2
SD/OD(1)
I/O
3
HIN
I
High side driver logic input (active high)
4
VCC
P
Lower section supply voltage
5
DT
I
Dead time setting
6
OP-
I
Op-amp inverting input
7
OPOUT
O
Op-amp output
8
GND
P
Ground
9
OP+
I
Op-amp non inverting input
10
CP+
I
Comparator input
O
Low side driver output
11
LVG
(1)
12,13
NC
14
OUT
Function
Low side driver logic input (active low)
Shutdown logic input (active low)/open drain
(comparator output)
Not connected
P
High side (floating) common voltage
15
(1)
HVG
O
High side driver output
16
BOOT
P
Floating section (bootstrap) supply voltage
1. The circuit provides less than 1 V on the LVG and HVG pins (@ Isink = 10 mA), with VCC > 3 V. This allows
the omission of the “bleeder” resistor connected between the gate and the source of the external MOSFET
normally used to hold the pin low. The gate driver ensures low impedance in SD conditions also. See
Section 4.
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5/54
Logic inputs
2
AN2738
Logic inputs
The L6390 has two logic inputs, HIN and LIN, to separately control the high side and low
side outputs, HVG and LVG. HIN is in phase with HVG, while LIN is out of phase with LVG.
The signal inversion on the low side input allows control of the half-bridge output with only
one control signal (see Figure 2).
Figure 2.
Input configuration
HIN
HVG
LIN
VPULSE_LS
DOUBLE INPUT
CONFIGURATION
HIN
LIN
DRIVER
VPULSE_HS
LVG
HVG
LVG
HVG
LIN
SINGLE INPUT
CONFIGURATION
HIN
DRIVER
VPULSE
HIN
LIN
HVG
LVG
LVG
Note that by connecting the two logic input signals together, the resulting dead time is
defined by the resistor connected between pin 5 and ground. The dead time can be set to a
wide range of values from hundreds of nanoseconds to a few microseconds (see Figure 5 or
the L6390 datasheet). All the logic inputs are provided with hysteresis (~1 V) for low noise
sensitivity and are TTL//CMOS 3.3 V compatible. Thanks to this low voltage interface logic
compatibility, the L6390 can be used with any kind of high performance controller, such as
microcontrollers, DSPs or FPGAs.
As shown in the block diagram in Figure 1, the logic inputs have internal pull-down (or pullup) resistors. The purpose of these resistors is to set a proper logic level in case, for
example, there is an interruption in the logic lines or the controller outputs are in tri-state
conditions. If logic inputs are left floating, the gate driver outputs LVG and HVG are set to
low level. The internal resistors are:
●
HIN logic input: 85 kΩ (typ.) pull-down
●
LIN logic input: 720 kΩ (typ.) pull-up connected to an internal 5 V regulator through a
diode
●
SD logic input: 375 kΩ (typ.) pull-down
If the logic inputs are connected together as in the single input configuration (Figure 2) and
they are left floating, the internal pull-down and pull-up resistors form a resistive divider
providing a voltage value (about 460 mV) which keeps the HVG off and LVG on, thus turning
on the low side power switch.
6/54
Doc ID 14589 Rev 2
AN2738
3
UVLO function
UVLO function
The L6390 supply voltage VCC is continuously monitored by an under-voltage lockout
(UVLO) circuitry which turns off the IC outputs when the supply voltage goes below the
VCC_thOFF threshold (see L6390 datasheet for values) and turns on the device when the
supply voltage goes above the VCC_thON voltage. A hysteresis of about 1.5 V is provided for
noise rejection purpose. The high voltage floating supply VBOOT is provided with a similar
under-voltage lockout circuitry also. When the L6390 is in UVLO condition, both gate driver
outputs are set to low level, setting the half-bridge power stage output to high impedance.
Figure 3 below shows the I-V characteristics of the output buffers at different VCC values.
Figure 3.
L6390 gate driver outputs in UVLO condition
ILVG/HVG
(mA)
VCC = 15V
VCC = 9V
VCC = 7V
100
VCC = 6V
90
80
VCC = 5V
70
60
VCC = 4V
50
40
30
VCC = 3V
20
10
VCC = 2V
VCC = 0V
0
1
VCC = 1V
2
3
4
5
VLVG/HVG
(V)
NOTE: when VCC < VLVG/HVG
the body diode of the p-channel (source)
turns ON and clamps the VLVG/HVG voltage
TEST CIRCUIT
GATE DRIVER
OUTPUT BUFFER
(LS/HS)
VCC/BOOT
p-channel
(source)
OFF
LVG/HVG
ON
ILVG/HVG
A
+
VCC
+
VLVG/HVG
n-channel
(sink)
GND/OUT
Doc ID 14589 Rev 2
7/54
Dead time and interlocking function management
4
AN2738
Dead time and interlocking function management
In order to avoid any possible cross-conduction between the power MOSFETs/IGBTs of the
half-bridge, the L6390 provides both the dead time and the interlocking functions. The
interlocking function is a logic operation which sets both the outputs to low level when the
inputs are simultaneously active (HIN to high level and LIN to low level). The dead time
function is a safety time introduced by the device between the falling edge transition of one
driver output and the rising edge of the other output. If the rising edge set externally by the
user occurs before the end of this dead time, it is ignored and results delayed until the end of
the dead time. The dead time can be adjusted externally through the value of the DT resistor
connected between pin 5 and pin 8 (see Figure 5). A 100 nF ceramic capacitor in parallel
with this resistor is recommended for noise immunity. In Figure 4 the details of dead time
and interlocking function management are described.
Table 2.
L6390 truth table
Inputs
Outputs
SD
LIN
HIN
LVG
HVG
L
X(1)
X(1)
L
L
H
H
L
L
L
H
L
H
L
L
H
L
L
H
L
H
H
H
L
H
1. Don’t care
8/54
Doc ID 14589 Rev 2
AN2738
Dead time and interlocking function management
Figure 4.
Timing waveforms
INTE
RLO
CKIN
G
HIN
INTE
CONTROL SIGNAL EDGES
OVERLAPPED:
INTERLOCKING + DEAD TIME
RLO
CKIN
G
LIN
LVG
DT
DT
HVG
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
LIN
CONTROL SIGNALS EDGES
SYNCHRONOUS (*):
DEAD TIME
HIN
LVG
DT
DT
HVG
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
LIN
CONTROL SIGNALS EDGES
NOT OVERLAPPED,
BUT INSIDE THE DEAD TIME:
DEAD TIME
HIN
LVG
DT
DT
HVG
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
LIN
CONTROL SIGNALS EDGES
NOT OVERLAPPED,
OUTSIDE THE DEAD TIME:
DIRECT DRIVING
HIN
LVG
DT
DT
HVG
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
gate driver outputs OFF
(HALF-BRIDGE TRI-STATE)
(*) HIN and LIN can be connected togheter and driven by just one control signal
Figure 5.
Typical dead time vs. DT resistor value
3.5
3
2.5
DT (us)
2
1.5
1
0.5
0
0
50
100
150
200
250
300
Rdt (kOhm)
Doc ID 14589 Rev 2
9/54
Smart shutdown function
5
AN2738
Smart shutdown function
The L6390 integrates a comparator for fault sensing purposes. The comparator has an
internal reference voltage Vref on its inverting input (see L6390 datasheet), while the noninverting input is available on pin 10. The comparator input can be connected to an external
shunt resistor in order to implement a simple over-current detection function. The output
signal of the comparator is fed to an integrated MOSFET with the open drain available on
pin 2, shared with the SD input.
When the comparator triggers, the device is set in shutdown state and both its outputs are
set to low level leading the half-bridge in tri-state.
Figure 6.
Smart shutdown timing waveforms
comp
Vref
CP+
PROTECTION
HIN/LIN
HVG/LVG
SD/OD
upper
threshold
lower
threshold
1
2
open drain gate
(internal)
real disable time
Fast shut down:
the driver outputs are set in SD state
immediately after the comparator
triggering even if the SD signal
has not yet reach
the lower input threshold
TIME CONSTANTS
1
= (RON_OD // RSD) CSD
2
= RSD CSD
SHUT DOWN CIRCUIT
VBIAS
RSD
FROM/TO
CONTROLLER
SD/OD
CSD
10/54
Doc ID 14589 Rev 2
RON_OD
SMART
SD
LOGIC
AN2738
Smart shutdown function
In common over-current protection architectures the comparator output is usually connected
to the SD input and an RC network is connected to this SD/OD line in order to provide a
mono-stable circuit, which implements a protection time that follows the fault condition.
Unlike common fault detection systems, the L6390 smart shutdown architecture allows to
immediately turn-off the output gate driver in case of fault, by minimizing the propagation
delay between the fault detection event and the actual outputs switch-off. In fact the time
delay between the fault and the outputs turn-off is no more dependent on the RC value of
the external network connected to the pin. In the smart shutdown circuitry, the fault signal
has a preferential path which directly switches off the outputs after the comparator
triggering. At the same time the internal logic turns on the open drain output and holds it on
until the SD voltage goes below the SD logic input lower threshold. The smart SD system
provides the possibility to increase the time constant of the external RC network (that is the
disable time after the fault event) without increasing the delay time of the protection. Any
external signal provided to the SD pin is not latched and can be used as control signal in
order to perform, for instance, PWM chopping through this pin. In fact when a PWM signal is
applied to the SD input and the logic inputs of the gate driver are stable, the outputs switch
from the low level to the state defined by the logic inputs and vice-versa.
A block diagram of the smart shutdown architecture is depicted in Figure 7.
Figure 7.
Smart shutdown equivalent circuitry
LIN
LVG
HIN
HVG
CP+
FSD
VBIAS
Vref
SD
Q
Q
S
FF
R
SET dominant FF
In normal operation the outputs follow the commands received from respective input signals.
When a fault detection event occurs the fault signal (FSD) is set to high by the fault detection
circuit output (LVG, HVG) and the FF receives a SET input signal. Consequently the FF
outputs set output signals to low level and, at the same time, turn-on the open drain
MOSFET which works as active pull-down for the SD signal. Note that the gate driver
outputs stay at low level until the SD pin has experienced both a falling edge and a rising
edge, although the fault signal could be returned to low level immediately after the fault
sensing. In fact even if the FF is reset by the falling edge of the SD input, the SD signal also
works as enable for the outputs, thanks to the two AND ports. Moreover once the internal
open-drain transistor has been activated, due to the latch, it cannot be turned-off until the
SD pin voltage reaches the low logic level. Note that, since the FF is SET dominant,
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Smart shutdown function
AN2738
oscillations of the SD pin are avoided if the fault signal remains steady at high level. The
block diagram of a power system using the gate driver with the smart shutdown architecture
is shown in Figure 8. An RC network is used to implement the disable time after a fault
detection event.
Figure 8.
Protection scheme
bias
In Figure 9 the typical duration of the disable time vs. the SD capacitance with different Rsd
values and Vbias values is shown.
Figure 9.
Disable time vs. SD capacitance (typical values)
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12/54
Doc ID 14589 Rev 2
AN2738
6
L6390 op-amp
L6390 op-amp
The L6390 integrates an operational amplifier optimized for field oriented control (FOC)
applications. In a typical FOC application a tri-phase power bridge is used and the currents
in the three half-bridges are sensed using a shunt resistor on the source of each low side
power switch. The analog current information is transformed in a discontinuous sense
voltage signal, available only when the current is flowing in the low side path and having the
same frequency of the PWM signal driving the bridge. The sense voltage is a bipolar analog
signal, which sign depends on the direction of the current (see Figure 10):
Figure 10. 3-phase system
SENSING:
3-PHASE DRIVER
Discontinuos Voltage at fPWM frequency
SINUSOIDAL VECTOR CONTROL
Vs
Vs
V
Iload x Rs
Vs
POWER
STAGE
t
3-PHASE
MOTOR
IPHASE
-Iload x Rs
PWM frequency in a typical home appliance motor driver application is in the range of 5-20
kHz. The sense voltage signals must be provided to an A/D converter in order to perform the
matrix calculation related to a certain control technique. Those sense signals are usually
shifted and amplified by dedicated op-amps in order to exploit the full range of the A/D
converter. The typical scheme shown in Figure 11 is used.
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L6390 op-amp
AN2738
Figure 11. General advanced current sense scheme
H.V.
ROUT resistor is usually required
in order to make the OPAMP stable
when the COUT capacitance increases
TO LOAD
VREF
R3
OPAMP
OP+ +
OPOUT
OP- -
R5
R4
Vsense
ROUT
to ADC
COUT
Rs
R2
Half-Bridge
Current Sensing
Capacitor required
by the ADC
for sampling purpose
C2
R1
Voltage Gain
and
Filtering
Voltage Shifting
of the Vsense
Principle waveforms corresponding to the above scheme are provided in Figure 12:
Figure 12. Advanced current sensing waveforms
V
0.6 V
H.V.
0V
t
TO LOAD
V
3.3 V
VREF
R3
OPAMP
R5
OP+
+
OPRs
OPOUT
to ADC
ROUT
-
R4
COUT
R2
C2
R1
0V
V
V
t
3.3 V
Iload x Rs
FILTERED SIGNAL
0V
t
-Iload x Rs
0V
t
ADCs used in vector control applications have a typical FSR of about 3.3 V. The sense
signals have to be shifted and centered on FSR/2 voltage (about 1.65 V) and amplified with
a gain which provides the matching between the maximum value of the sensed signal and
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AN2738
L6390 op-amp
the FSR of the ADC. In Figure 13 an application circuit of the op-amp with typical passive
components values is shown.
Figure 13. L6390 op-amp, application example
Following Figure 14 shows the output voltage waveform of the L6390 op-amp considering
the voltage level shifting and the gain amplification provided by the amplifier configuration of
the op-amp, neglecting the low-pass filtering action of the feedback network. In particular
the waveform is zoomed on the maximum amplitude of PWM sinusoidal voltage signal
which represents the extreme condition for the external slew rate of the op-amp. The
required external slew rate increases proportionally to the voltage excursion and reverse
proportionally to the ON time of PWM output.
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L6390 op-amp
AN2738
Figure 14. Typical L6390 ideal output
Typically the maximum voltage variation on op-amp output is FSR/2 (about 1.65 V) and the
ON time usually does not show its minimum value where the PWM voltage amplitude is
higher. In fact the duty cycle of each half-bridge of the 3-phase power stage is proportional
to the average voltage applied to each 3-phase terminal which normally is not in phase with
the output current of the same bridge, due to the load angle and the BEMF of the running
motor. The ADC front-end usually samples the op-amp output voltage in the middle of the
ON time, so the output must approach its final value as much as possible within one half of
the ON-time. Considering a minimum ON time of about 6 µs, the op-amp output settling time
must be lower than 3 µs. If the maximum output voltage variation is 1.5 V and the maximum
time needed to reach the desired value is 3 µs, the minimum SR requirement for proper
operation would be 1.5 V/ 3 µs = 0.5 V/µs (see Figure 15).
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AN2738
L6390 op-amp
Figure 15. Detail on single PWM cycle in advanced current sensing for FOC systems
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Bootstrap driver
7
AN2738
Bootstrap driver
A bootstrap circuitry is needed to supply the high voltage section. This function is normally
accomplished by a high voltage fast recovery diode (Figure 16 a). In the L6390 a patented
integrated structure replaces the external diode. It is realized with a high voltage DMOS
driven synchronously with the low side driver (LVG), with a diode in series, as shown in
Figure 16 b. An internal charge pump (Figure 16.b) provides the DMOS driving voltage.
7.1
CBOOT selection and charging
To choose the proper CBOOT value the external MOSFET can be seen as an equivalent
capacitor. This capacitor CEXT is related to the MOSFET total gate charge:
Equation 1
Q GATE
C EXT = ----------------V GATE
The ratio between the capacitors CEXT and CBOOT is proportional to the cyclical voltage
loss. It has to be:
Equation 2
C BOOT » C EXT
For example: if QGATE is 30 nC and VGATE is 10 V, CEXT is 3 nF. With CBOOT = 100 nF the
drop would be 300 mV.
If HVG has to be supplied for a long time, the CBOOT selection has to take into account also
the leakage and quiescent losses.
For example: since the HVG steady state consumption is lower than 200 μA, if HVG TON is 5
ms then CBOOT has to supply 1 µC to CEXT. This charge on a 1 µF capacitor means a
voltage drop of 1 V.
The internal bootstrap driver gives a great advantage: the external fast recovery high
voltage diode can be avoided (it usually has great leakage current). This structure works if
VOUT is close to GND (or lower) and in the meanwhile the LVG is on. The charging time
(Tcharge) of CBOOT is the time in which both conditions are fulfilled and it has to be long
enough to charge the capacitor.
The bootstrap driver introduces a voltage drop due to the DMOS RDS(on). At low frequency
of operation this drop can be neglected, but if the frequency is increased the drop must be
taken into account.
The following equation is useful to compute the drop on the bootstrap DMOS:
Equation 3
Q GATE
V DROP = I CHARGE ⋅ R DS ( on ) → V DROP = ------------------------- ⋅ R DS ( on )
T CHARGE
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AN2738
Bootstrap driver
where QGATE is the gate charge of the external power MOSFET, RDS(on) is the on resistance
of the bootstrap DMOS and TCHARGE is the charging time of the bootstrap capacitor.
For example: given the typical value of 120 Ω for RDS(on) and using a power MOS with a total
gate charge of 30 nC, the drop on the bootstrap DMOS is about 1 V if the TCHARGE is 5 µs.
In fact:
Equation 4
30nc
V DROP = ------------- ⋅ 120Ω ≈ 0.7V
5μs
VDROP has to be considered when the voltage drop on CBOOT is calculated: if this drop is too
high, or the circuit topology doesn't allow a sufficient charging time, an external diode can be
used.
Working at very low frequencies the high side driver on-time can be very long. So CBOOT
voltage can drop because of HVG steady state consumption. To avoid extremely large
capacitor (> 1-2 µF) an external charge pump can be added (see Figure 17 as example). It
is mandatory for the diodes to have a low parasitic capacitance, because C1 and C2 should
be greater than diodes capacitance. The oscillator has to work in order to balance the high
voltage side consumption, and the minimum frequency is fixed by C1 and C2 values (with
C1 = C2 = 33 pF then f > 250 - 300 kHz).
Figure 16. Bootstrap driver
DBOOT
BOOT
VCC
BOOT
VCC
H.V.
HVG
H.V.
HVG
CBOOT
CBOOT
OUT
OUT
TO LOAD
TO LOAD
LVG
LVG
a
b
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Bootstrap driver
AN2738
Figure 17. External charge pump
HV
VBOOT
Cboot
200nF
HVG
L639x
OUT
LOAD
330pF
LVG
GND
C2
33pF
C1
33pF
VCC
HCF4069UB
Cx
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AN2738
Application example
A typical scheme for a 3-phase inverter application is shown in Figure 18. In order to drive a
3-phase load as for example a BLDC or Induction AC motor, three L6390 ICs can be used.
Figure 18. 3-phase drive- typical scheme
L6390
CONTROLLER
L6390
HV BUS
L6390
8
Application example
ADC
3-phase
MOTOR
Each L6390 IC drives one half-bridge of the 3-phase power stage. The three gate drivers
are driven by the controller and the sensing signals coming from the power stage are
managed directly by the L6390 analog blocks (comparators and op-amps) which provide the
proper feedback signals to the ADC and the system controller. Each half-bridge comprises
two power switches such as IGBTs (or MOSFETs), one high side and one low side. When
the high side switch is ON, it brings the output voltage of the half-bridge to the HV bus
voltage, which can be a high DC voltage power supply with large power availability, while the
low side switch shorts the output to power ground voltage when it is ON. Thanks to the
internal floating structure for the high side switch driving, n-type IGBTs (or N-channel
MOSFETs) can be used. The gate drivers work exactly as digital/analog-power interface
between the controller and the power stage. A shunt resistor can be placed between each
low side switch and the power ground, to sense the current on each low side. The
information contained in each current sense signal can be conditioned by the L6390 op-amp
as described in Section 6.
In Figure 19 a more detailed schematic of a 3-phase power stage topology is shown. Note
that usually is recommended (not mandatory) to have some gate resistances between each
power switch gate and the correspondent gate driver output, in order to limit the current
during the gate charge. Then some filtering and/or level shifting RC networks should be
added between the shunt resistors and the correspondent comparator and op-amp input
pins of each gate driver. A unique RC network can be used for the SD pin even if it is
strongly recommended to split the capacitor in three equal components in parallel in order to
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Application example
AN2738
place each one very close to the SD input pin of each IC. This capacitance and, in general,
each capacitance indicated in the schematic must be placed as close as possible to the IC
in order to guarantee a good noise filtering action.
Figure 19. Typical application schematic of a 3-phase FOC
H.V.
+
VCC
VBOOT
VCC
+
-
GND
L6390
HVG
HIN
OUT
LIN
LVG
SD/OD
CP+
DT
OP+
U1
OPOUT
OP-
H.V.
+
VCC
IN1
-
GND
IN2
CONTROLLER
VBOOT
VCC
+
HVG
HIN
IN3
OUT
LIN
LVG
3.3V
SD
+
L6390
SD/OD
CP+
3-phase
MOTOR
DT
OP+
A1
ADC
U2
OPOUT
OP-
A2
A3
H.V.
+
VCC
-
VBOOT
VCC
+
GND
L6390
HVG
HIN
OUT
LIN
LVG
SD/OD
CP+
DT
OP+
OPOUT
U3
OP-
In the example of Figure 19 the logic inputs HIN and LIN are connected together, using just
one single signal to drive each L6390.
In Figure 20 a special focus on a single gate driver circuit of the 3-phase L6390 scheme is
provided.
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AN2738
8.1
Application example
VCC supply pin
Regarding the VCC pin, a local filtering of the supply voltage very close to the L6390 IC is
recommended. Generally the suggestion is to use two capacitors, one electrolytic with
greater value (CVCC1 = 10 µF, for example) which has a great energy capability but also a
not negligible ESR (so is quite slow in providing the current) and a second smaller ceramic
capacitor (CVCC2 = 100 nF) which has a lower ESR value but a lower energy capability. The
first capacitor works mainly as bulk energy storage while the second one is able to supply
the dynamic current spikes required by the commutations of the device, so is better for high
frequency decoupling of the IC supply voltage. On the other hand, the selection of the
proper value for the CBOOT capacitor is already described in Section 7.
Figure 20. Ex. of an application circuit for one of the three half-bridges of a 3-phase
power stage
In the example a VCC supply voltage of 15 V is used. This voltage is approximately the
same voltage provided to the gate of the power switches.
8.2
BOOT (floating) supply pin
The bootstrap capacitor to supply the high-side floating section of the gate driver must be
placed between the BOOT pin 16 and the OUT pin 14. For a deep description of the proper
dimensioning of the bootstrap capacitor please refer to Section 7 on page 18. The capacitor
must be placed as close as possible to the related IC pins. The bootstrap diode required for
the charge of the bootstrap capacitor is integrated inside the L6390 device.
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Application example
8.3
AN2738
Logic input pins
The logic input pins can be connected directly to the controller with the suggestions
provided in Section 2 on page 6. If the application environment is very noisy and the logic
input voltage is low (e.g. 3.3 V), it can be useful to place some small RC network (not
showed in Figure 20) in series with the logic input lines, in order to avoid false input
triggering due external noise.
8.4
Shutdown pin
Dimensioning of the SD network (RSD and CSD) is provided in Section 5.
8.5
Dead time pin
The resistance value on the DT pin must be selected following the indications in Section 4
and in Figure 5. It is recommended to connect between DT and GND pins a capacitor with a
value of at least 100 nF, as close as possible to the IC and with short PCB tracks.
8.6
Op-amp
As explained in the previous paragraphs, the L6390 op-amp is completely uncommitted so a
large amount of amplifier configurations can be implemented by the application designer. An
example of amplifying network is provided in Section 6. Typically low tolerance resistors are
used, if high accuracy is required in the conditioning of analog signals.
8.7
Comparator input
No particular external circuits are required apart some noise filtering network useful to avoid
false triggering of the comparator due to voltage spikes on sense resistor. In the example of
Figure 20 a common RC network is implemented (time constant ~1 µs).
8.8
Sense resistor
The value of the sense resistor must be chosen considering mainly the current rating of the
application and the amplitude of the sense voltage desired. The power rating of the sense
resistor must be large enough to withstand the maximum current of the application. More
resistors in parallel are often used in order to withstand higher power requirements and to
obtain low resistance values with low parasitic inductance, which must be as low as possible
in order to reduce the dynamic below-ground voltage on the OUT pin of the gate driver.
8.9
Gate driver outputs: gate lines
The gates of the power switches and the gate driver outputs can be connected directly, but
usually some gate resistors are placed in series on the gate lines in order to limit the gate
current during commutations. The final target is to control the dVOUT/dt of each half-bridge
output and then reducing the EMI. A more detailed explanation of the mechanisms behind
the dVOUT/dt control through the gate resistors is provided in the next sub-paragraph.
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AN2738
Application example
Consider that all the following considerations should be considered as approximated
analyses of the gate charge phenomenon, so for a proper sizing of the gate resistor is
always strongly recommended to evaluate the resulting power bridge transitions in bench
analyses.
As shown in Figure 20, the gate line is split into separated paths, one for the turn-on (with, in
the example, a gate resistor of 33 Ω) and the other one for the turn-off, using a small signal
diode as path selector the equivalent turn OFF resistance is in a first approximation the
parallel of the turn OFF and the turn ON resistances (neglecting the diode drop). In the
example the turn-off resistance is set to 0 Ω to provide the lowest resistance path for the
turn-off of the IGBT. In fact, as explained in the two following paragraphs, low impedance on
the gate driver turn-off helps in reducing the induced turn-on phenomena.
Regarding the layout of such gate lines, it is always strongly recommended to place the
power IGBT (or MOSFETs) very close to the gate driver. It is important to reduce as much
as possible the lengths of such line paths as well the areas included in the gate circuits,
because these can act as weak antennas and could catch noise from the surrounding
environment. The larger these areas, the higher the gain of such undesired antenna circuits.
Figure 21. Layout suggestion for the gate driving circuits
MINIMIZE THIS LENGHT
HVG
MINIMIZE THIS AREA
OUT
MINIMIZE THIS LENGHT
LVG
MINIMIZE THIS AREA
GND
8.10
Gate driving: principle of working with inductive load
In this paragraph a deeper description of the power IGBT (or MOSFET) gate driving, in case
of inductive load, will be provided. The following explanations and calculations are just
intended to provide a general understanding of physical principles lying behind the
phenomenon of the dVOUT/dt control through the gate current limitation performed by the
gate resistors. The target is to help the application designer which uses L6390 ICs to be
aware of the various contributes that different parameters of the gate driver system have on
the power transitions and also to distinguish the main effects on which he should focus on.
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Application example
AN2738
Calculations and formulae should be considered as qualitative indications and not for an
accurate quantitative use since, as explained in the above paragraph, they are the result of a
first approximation study. The experimental verification of the design choices is always
recommended.
The description distinguishes and explores the two main actors of the gate driver system: by
one side the inner structure of the gate driver output buffers is described and on the other
side the switching mechanism of a generic IGBT (or MOSFET) is analyzed in detail. In
Figure 22 a simplified view of the L6390 gate driver output buffers is provided. Each one can
be considered as a CMOS push-pull stage where a P-channel MOSFET works as source
driver while an N-channel MOSFET works as sink driver. The structure is similar both for the
low side and for the high side and the behavior can be considered the same. In fact the high
side driver can be thought as a floating buffer having as supply the VBOOT voltage and as
reference the OUT pin. The CBOOT capacitor represents the floating supply voltage source
of the high side driver. During the charge of the power switch gate, each source/sink
MOSFET can be considered (in first approximation, for simplicity) as if it would be in the
ohmic region, so it can be represented as an equivalent resistor with a value equal to its
RDS(on). Thanks to this approximation it is possible to use a simplified equivalent circuit for
the turn-on and the turn-off commutation (see Figure 22 and 23). Regarding the turn-on, this
gate charge circuit has two resistors in series (RGATE_ON and RDS(on)_SOURCE) and a supply
voltage which is VCC for the low side and VBOOT-OUT for the high side. Regarding the
turn-off the equivalent circuit is composed just by two resistors (RGATE_OFF and
RDS(on)_SINK) connected to the source of the power switch.
Gate driver output buffers: equivalent circuit
Figure 22. Gate driver output: equivalent circuit for turn-on
HVG or LVG
OUTPUT DRIVER
BOOT or VCC
OUT or HVbus
RDSon_source
RGATE
Vge
RDSon_source
RGATE
Vge
+
HVG/LVG
(Vboot-Vout) or VCC
OUT or GND
Figure 23. Gate driver output: equivalent circuit for turn-off
HVG or LVG
OUTPUT DRIVER
BOOT or VCC
OUT or HVbus
Vge
HVG or LVG
RDSon_sink
RGATE
OUT or GND
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Doc ID 14589 Rev 2
RDSon_sink
RGATE
Vge
AN2738
Application example
This first simplification is related to the inner structure of the gate driver circuitry. Let now
focus for a while on the side of the power bridge and its commutations. In a common halfbridge composed by a high side and a low side switch the transitions of the power IGBTs (or
MOSFETs) are not all similar, but they can be distinguished in two main types: soft switching
and hard switching (see Figure 24 and 25). The main element playing a key role in the
dynamic of transitions is the direction of the current flowing in the power IGBT (or MOSFET)
under evaluation which depends on the direction of the load current respect to the halfbridge. Mainly if the current has the same direction of the power switch the commutation is
hard. If it is in opposite direction the commutation is soft.
Figure 24. Hard switching
CURRENT GOING IN THE BRIDGE
+
LOW SIDE SWITCHING
CURRENT GOING OUT FROM THE BRIDGE
+
HIGH SIDE SWITCHING
OFF
OFF
OFF
ON
ON
OFF
ON
OFF
Iload
ON
Iload
OFF
Figure 25. Soft switching
CURRENT GOING OUT FROM THE BRIDGE
+
LOW SIDE SWITCHING
OFF
OFF
ON
Iload
ON
CURRENT GOING IN THE BRIDGE
+
HIGH SIDE SWITCHING
OFF
ON
ON
OFF
OFF
Iload
OFF
when the power switches ON
the diode is already turned ON
As shown in Figure 24 and 25, there is hard switching when the high side is commutating
and the load current is exiting the bridge or when the low side is commutating and the load
Doc ID 14589 Rev 2
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Application example
AN2738
current is entering the bridge. This type of transition is called “hard” because the power
switch turns on when the related VCE (VDS) is at maximum; VCE (VDS) goes back to
maximum during turn-off (VCE is close to zero while the switch is steadily ON). The transition
is completely managed by the switch that, during the commutation, dissipates some energy.
On the other hand, if the switch turns on when its VCE is already closed to zero or VCE
remains close to zero after turn-off, the transition is soft and, during the commutation, the
switch dissipates almost no energy. This last condition happens when the free-wheeling
diode of the switch is bringing the current because the diode is in the same direction of the
load current. As a general rule, consider that the dynamic of the transition (dVOUT/dt, VCE
rise/fall time etc.) is always managed and controlled by the power switch in hard switching,
while its companion switch is necessarily in soft switching because it is in the opposite
direction of the load current. Considering this, in order to understand the power IGBT
(MOSFET) transition, the hard switching transition will be further investigated. Note that all
of the descriptions that follow consider an inductive load connected to the half-bridge stage
output.
In Figure 26, the dynamics of the hard switching for the turn-on transition are described. The
first graph is related to the gate charge curves of the power IGBTs (or MOSFETs). This is a
typical graph which is also available in the datasheets for the IGBT or MOSFET and
describes the dependence of the amount of charge required by the IGBT gate on the
voltage drop between the gate and the emitter (or source). In the second graph the collector
current (IC) and VCE voltage versus time are shown. The third graph shows the working
point of the IGBT traced on the various Ic vs. VCE characteristic curves for different VGE
voltages. For a single turn-on transition, four main phases related to the power IGBT
(MOSFET) commutation can be distinguished. On the left side of Figure 26, the IGBT
conditions for each phase are described. During phase T1 the gate begins to be charged,
but the power IGBT still is not conducting because VGE is below the threshold of the IGBT. In
this state, the IGBT current is zero and its VCE is at maximum, while VGE is gradually
increasing. In phase T2 the VGE voltage goes above to the IGBT threshold and the IGBT
starts to bring part of the load current. VCE remains fixed at maximum level because the
free-wheeling diode of the companion IGBT is still bringing the rest of the load current and is
still conducting, thereby clamping the VCE voltage. Both in T1 and T2 phases, the gate
current contributes to charge the equivalent VGE parasitic capacitance of the IGBT. In the Ic
vs. VCE characteristic plot the T2 phase is the vertical section of the working curve trace,
because the IGBT is moving on its own characteristic with a constant VCE and an increasing
current, while its VGE is increasing also. When the amount of current flowing in the IGBT is
equal to the load current, the diode turns off and the VCE voltage starts to decrease because
it is no longer clamped by the free-wheeling diode of the other IGBT. The working point on Ic
vs. VCE curve reaches the load value and starts to move horizontally on the Ic constant
curve in the direction of decreasing VCE voltages. This is the T3 phase, usually called the
“plateau” phase. This name derives from the fact that the gate charge curve is horizontal for
the entire T3 phase, until VCE reaches the VCE_sat value corresponding to the load current.
Note that the VGE voltage is constant, although the gate current flowing in the IGBT gate is
not zero. The reason for this is that the whole gate current is used to charge the CGC (Miller)
parasitic capacitance which then experiences a dV/dt on its terminals because the VCE
voltage is decreasing after the diode turn-off. As a first approximation, if CGC was constant
(currently it is not), the dVOUT/dt could be calculated as follows:
Equation 5
I SOURCE
OUT⎞
⎛ dV
----------------= ---------------------⎝ dt ⎠ FALL
C GC
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Doc ID 14589 Rev 2
AN2738
Application example
The formula above shows how the source current coming from the gate driver can directly
control the dVOUT/dt of the power half-bridge. This calculation is just a first approximation,
since typically the CGC Miller capacitance value is not constant but depends on the VCE
voltage, CGC is not linear. The main result of the abrupt variation of the CGC is that the
actual slope of the OUT transition is composed of two different dVOUT/dt slopes, one faster
and the other slower compared to the value obtained by the above formula (see Figure 26).
In this document only the first approximation approach will be used.
Figure 26. Turn-on hard switching details with induction load: gate charge and
plateau phase
GATE CHARGE CURVES
T1
Vge
OFF
Qg
diode ON
Vge_max
Vge
Qge
Qgc
Isource
plateau
Cge
Vce
=
HVbus
Ic = 0
Vgs_p
Ton
Vth
T2
OFF
t, Q
diode ON
Vce, Ic
Vge
HVbus
Isource
Ic
APPROX.
Vce
=
HVbus
Cge
Iload
REAL
T3
OFF
T2
Cgc
T3
Ton
Cgc = Isource
diode OFF
t
T1
dVce
dt
T4
Isource
Tfall
Vge = Vge_p
Cge
Ic vs Vce CHARACTERISTICS
Ic
Vge = Vge_max
Vce
Ic
=
Iload
PLATEAU
PHASE
Tfall
T4
Iload
T4
Vge = Vge_p
T3
OFF
diode OFF
plateau
T2
Vge
Vge
Isource
Cge
Vce_sat
HVbus
Ic
=
Iload
Vce = Vce_sat
Vce
WORKING
CURVE
Due to CGC non-linearity, the IGBT (or MOSFET) datasheet also reports the equivalent total
amount of charge required during the different gate charge phases: the total gate charge Qg
required for turning on the IGBT (or MOSFET) completely, the QGE required for increasing
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Application example
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the VGE up to the plateau voltage and the QGC required during the whole plateau phase.
This last amount of charge is also called the plateau charge. The time required to provide
the complete plateau charge is the Tfall time, and this is the same time necessary for
completing the VCE transition. For Ton time is intended the time delay between the
beginning of the gate charge and the full conduction of the IGBT (or MOSFET), when the
power switch current equals the full load current. So Tfall can be calculated as follows:
Equation 6
Q GC
T fall = ---------------------I SOURCE
It is now possible to merge the equivalent gate driver output circuit reported in figures 24
and 25 and the considerations regarding the different gate charge phases in the total
equivalent circuit reported in Figure 27.
Figure 27. Total equivalent circuit for the turn-on
Ton
RDSon_source
BOOT/VCC
+
Tfall
RGATE
Isource
RDSon_source
Vge
BOOT/VCC
+
RGATE
Isource
+
Vge_p
Cge =
Qge
Vge_p
= Ciss_min
From the above circuit the value of the transition times could be calculated (the following
formulae are related to the low side transition, but the same are suitable also for the high
side by exchanging “VCC” with “VBOOT-VOUT”):
Equation 7
VCC
T ON = ( R DS ( on ) + R GATE ) ⋅ C ISSmin ⋅ In ⎛⎝ ----------------------------------⎞⎠
VCC – V GEP
Equation 8
( R DS ( on ) + R GATE )
Q GC
T FALL = ---------------------- = Q GC ⋅ -------------------------------------------------VCC – V GEP
I SOURCE
CISS_min is used because when the VCE (or VDS) of the power IGBT (or MOSFET) is
maximum (equal to HVbus), CISS (which depends on the VCE) shows its minimum value.
During the turn-off of the power IGBT (or MOSFET), the behavior is the same as the turnon, but in reversed time order. In Figure 28, the gate charge characteristics and VCE vs. IC
curves are provided. The time required to sink the complete plateau charge is the Trise time,
while for Toff time is intended the time delay between the beginning of the gate charge and
the increase of Vce (or Vds) voltage.
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Doc ID 14589 Rev 2
AN2738
Application example
Figure 28. Turn-off hard switching details with induction load: gate charge and
plateau phase
T1
GATE CHARGE CURVES
Vge
OFF
diode OFF
Qg
Vge_max
Vge
Qge
Qgc
Isink
Cge
plateau
Vgs_p
Ic
=
Iload
Vce = Vce_sat
Toff
Vth
T2
dVce
dt
OFF
t, Q
Vce, Ic
Cgc = Isink
diode OFF
Cgc
HVbus
Isink
APPROX.
Vge = Vge_p
Final tail
only for
IGBTs
Iload
REAL
Cge
Vce
Ic
=
Iload
PLATEAU
PHASE
Trise
T3
OFF
diode ON
t
Vge
T1
T2
Toff
T3
T4
Isink
Trise
Iload
Vce
=
HVbus
Cge
Ic vs Vce CHARACTERISTICS
Ic
Ic
Vge = Vge_max
T1
Vge = Vge_p
T2
T4
OFF
plateau
diode ON
T3
Vge
Vge
Isink
Cge
HVbus
Vce_sat
Vce
Ic = 0
Vce
=
HVbus
WORKING
CURVE
The rising dVOUT/dt and Trise can be calculated as follows:
Equation 9
I SINK
OUT⎞
⎛ dV
----------------= ------------⎝ dt ⎠ RISE
C GC
Equation 10
Q GC
T RISE = ------------I SINK
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Application example
AN2738
As discussed above, the equivalent gate driver output circuit can be represented as in
Figure 29:
Figure 29. Total equivalent circuit for the turn-off
Trise
RDSon_sink
Toff
RGATE
Isink
RDSon_sink
+
RGATE
Isink
Vge
Vge_p
Cge = Ciss_max
Given the equivalent circuits in Figure 29 and the approximations of the power IGBT
(MOSFET) switching behavior, the timing equations are as follows:
Equation 11
V GE⊥MAX
T OFF = ( R DS ( on ) + R GATE ) ⋅ C ISSmax ⋅ In ⋅ ⎛⎝ -------------------------⎞⎠
V GEP
Equation 12
( R DS ( on )SINK + R GATE )
Q GC
T RISE = ------------= Q GC ⋅ --------------------------------------------------------------V GEP
I SINK
As above, CISS_max is used because when the VCE (or VDS) of the power IGBT (or
MOSFET) is minimum (equal to VCE_sat) the CISS (which depends on the VCE) shows its
maximum value. Even if they are just approximations, the Tfall and Trise are very important
values to be estimated because they provide an indication of the power dissipation during
the switching of the power IGBT (or MOSFET), which can be approximated as indicated in
Figure 30.
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AN2738
Application example
Figure 30. Power dissipation during switching (approximation)
turn ON
turn OFF
Vce, Ic
Vce, Ic
HVbus
HVbus
approx.
approx.
Iload
Iload
real
real
t
t
T1
T2
T3
Ton
T4
T1
T2
Toff
Trise
T3
T4
T3
T4
Tfall
HVbus
Pdiss_sw
Iload
Pdiss_sw
HVbus
Iload
conservative
approximation
conservative
approximation
real
real
t
t
T1
T2
T3
Ton
T4
T1
T2
Toff
Trise
Tfall
T2 is typically very short
compared to T3
T3 is typically very short
compared to T2
Since the T2 time for the turn-on and T3 time for the turn-OFF are much shorter compared
to Tfall and Trise, respectively, the approximated instantaneous amount of energy dissipated
during the commutation can be calculated as the triangular area having as its base the Tfall,
or Trise time and as its height the product of the maximum VCE voltage (equal to HVbus) and
the load current value:
Equation 13
( HV bus ⋅ I LOAD ) ⋅ ( T fall + T rise )
E DISS – SW ≅ --------------------------------------------------------------------------------2
Equation 14
( HV bus ⋅ I LOAD ) ⋅ ( T fall + T rise ) ⋅ f SW
P DISS – SW ≅ ----------------------------------------------------------------------------------------------2
The term fSW represents the switching frequency of the power IGBT (or MOSFET).
Based the results obtained by these approximate calculations, it should be clear that the
main consideration when dimensioning the gate resistor values is the trade-off between
electromagnetic emission and the power dissipated during power IGBT (or MOSFET)
switching, as shown in Figure 31.
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Application example
AN2738
Figure 31. RGATE dimensioning criteria
Pdiss_sw
Electromagnetic
emissions
Maximum allowed
EMI
Rgate dimensioning trade off
Rgate
RGATE influences the power dissipation on one side and the EMI effects on the other.
Therefore, the best trade-off must be determined during the application design-in phase.
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9
Induced turn-on phenomenon
Induced turn-on phenomenon
One possible phenomenon to be analyzed is the induced turn-on that could occur on a
turned-off power IGBT (or MOSFET) when its companion on the same half-bridge is
switching on. This phenomenon is due to the current injected on the gate by the Miller
parasitic capacitance (CGC) which causes an undesired voltage increase on the gate of the
power IGBT (or MOSFET) which should be kept well turned off. Induced gate voltage
depends on the absolute value of the parasitic capacitance CGC, its relative ratio with CGE,
the value of the dVOUT/dt of the half-bridge and the value of the equivalent (turn-off)
resistance between the emitter (or the source) and the gate. Figure 32 illustrates this
phenomenon.
Figure 32. Induced turn-on phenomenon - circuital description
Note:
Typically the induced turn-on phenomenon does not cause a complete turn-on of the power
MOSFET, so it is rare to have destructive cross-conduction on the half-bridge during
commutations. Nevertheless, weak conduction of the opened power switch could increase
the power dissipation of the power stage, increasing the overall temperature of the power
MOSFETs and reducing the efficiency. This is why this phenomenon deserves particular
attention also in terms of thermal performance of the power application.
Some strategies to reduce this phenomenon are:
a)
Reducing resistance path between gate and emitter (source). Reducing as
much as possible the gate resistance in which the injected current flows, the
voltage drop on the gate becomes lower. The drawback is a possible increase of
the dVOUT/dt during turn-off of the power IGBT (or MOSFET) which commutates in
hard switching, and then an increase of electrical noise and EMI issues (as
explained in the previous paragraph, the power IGBT or MOSFET in hardswitching is always the one which has the channel in the same direction as the
current). On the other hand, usually the dVOUT/dt during turn-off is lower than
during turn-on, because the plateau voltage is closer to the source voltage than
the supply voltage of the driver. This results in a lower gate current for the
discharge of the gate charge.
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Induced turn-on phenomenon
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AN2738
b)
Reducing the maximum dVOUT/dt. This reduction can be achieved by increasing
the gate resistor value which limits the gate current for the turn-on of the power
MOSFET. The drawback is a consequent increase of the switching time which
means dissipating more power during commutations.
c)
Using a MOSFET with lower CGD/CGS (or CGC/CGE for IGBTs) ratio. This
solution is not always the simplest, but it could be the best strategy where the
induced turn-on phenomenon is dominant.
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AN2738
10
How to increase the gate driver output current capability
How to increase the gate driver output current
capability
In some cases, certain applications may require more gate driver output current capability.
This requirement could be found in systems having a power rating higher than about 1 kW.
In those applications, in fact, typically the power MOSFET/IGBTs have a large gate charge
which contributes to a slowing down of the power switch transition (the dVOUT/dt), increasing
the power dissipation during each commutation. Note that also in applications with power
rating higher than 1 kW, the output current capability of the L6390 could be enough if the
power dissipation for commutation is acceptable (the power dissipation due to RDS(on) or
VCESAT is not dependent on the gate current). But if the limitation of this power contribution
is a constraint, the dVOUT/dt of each transition must be increased by enhancing the current
capability of the gate driver.
A simple way to increase the current capability of the gate driver outputs is to insert, in
series with the two gate lines of one half-bridge gate driver IC, two external current buffers
(Figure 33).
Figure 33. Block diagram of output current capability enhancement using external
current buffers
CURRENT
BUFFERS
H.V.
BOOT
HVG
GATE
DRIVER
OUT
TO LOAD
VCC
LVG
GND
Typically the current buffers are implemented using bipolar NPN-PNP push-pull noninverting (emitter follower configuration) structures. In Figure 34, a typical circuit using a
gate driver with bipolar push-pull current buffers is shown.
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How to increase the gate driver output current capability
AN2738
Figure 34. Example of a gate driving circuit with current buffers for current
capability increasing
Very close to push-pull stage
H.V.
BOOT
GATE
DRIVER
100 nF
25V
HVG
+
15V
-
VCC
+
10 uF
25V
100 nF
25V
STS01DTP06
100 nF
25V
+
100 uF
400V
0-10 Ω
0.25W
0-10 Ω
0.25W
GND
OUT
VCC
TO LOAD
STS01DTP06
100 nF
25V
LVG
0-10 Ω
0.25W
0-10 Ω
0.25W
Very close to push-pull stage
In the example above, the STMicroelectronics devices STS01DTP06 are dual NPN-PNP
complementary bipolar transistors rated for 1 A of current and available in the small SO-8
package. The two push-pull structures are placed on each gate driving path and must be
provided with a decoupling capacitor very close to device pins (see Figure 34).
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11
The below-ground voltage on the OUT pin
The below-ground voltage on the OUT pin
A typical phenomenon found in a great number of power applications is the below-ground
voltage experienced by the OUT pin, which sometimes can be very high. Contrary to
common opinion, the real problem of the below-ground voltage is not in the maximum
absolute value of the OUT pin, but rather the voltage on BOOT pin and the over-charging of
the bootstrap capacitance. In the paragraph that follows the root causes of the below-ground
voltages will be explained in detail, and the description of the real issues which could result
from the phenomenon will be provided.
11.1
The below-ground voltage phenomenon
In power applications using half-bridge topologies and typically driving Ioads with a
significant inductive component, the output of the power half-bridge experiences,
systematically, a below-ground voltage transition, which can be distinct in a dynamic
contribution consisting of a greater undershoot spike and in a static contribution, which is a
below-ground static voltage with lower absolute value (Figure 35b). This phenomenon
occurs when the bridge carries out a so-called hard switching transition towards a low
voltage level and the load current is outgoing (from the bridge to the load). When the high
side switch turns off, the output current tends to remain quite constant due to the inductive
component of the load, and then has to flow through the low side freewheeling diode, which
turns on going from a high voltage reverse condition to a forward condition. It is evident that
until the output bridge voltage has reached the “zero” value, the diode is turned off, so the
output transition is dominated by the high side turn-off commutation. After the output voltage
reaches the zero-voltage level, the diode can turn on and it begins to bring the entire load
current in a very brief time, so the high dIF/dt causes the well-known forward peak voltage,
which is the main contribution to the undershoot spikes. Other contributions to dynamic
below-ground voltage are the spikes due to the high dI/dt experienced by the parasitic
inductances in series with the free-wheeling diode located along the turn-off current path of
the half-bridge (Figure 35).
Figure 35. below-ground voltages in L6390
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The below-ground voltage on the OUT pin
AN2738
The overall below-ground voltage on the OUT pin can be calculated as follows:
Equation 15
V OUTmin⊥static = – ( R SENSE ⋅ I LOAD + V F )
Equation 16
dIF
VBGV _ spike = VFPK + LPARASITIC ⋅
+ VOUT min_ static
144244
3
dt
1444424444
3 static
contribution
dynamic
contribution
where:
11.2
●
VFPK is the free-wheeling diode transient peak forward voltage, which depends mainly
on the device technology and on the dIF/dt of the current in the diode. Typical values
could be from a few volts to more than 10 V. In Figure 36, VFP vs. dIF/dt of the
STTH1L06 diode is shown.
●
the dIF/dt is the current slope in the low side IGBT/MOSFET and could have a value
from a few tens to several hundreds of A/µs. The value depends mainly on the power
switch characteristics and in part on the driving current.
●
LPARASITIC represents the sum of all parasitic inductances on the current path and
mainly depends on the PCB layout. Generally, during the design of the power
application it is important to pay attention to the layout of the power bridges in order to
limit this parameter. Typical values of a good layout are in the order of some tens of nH.
Note that it is useful also to use RSENSE resistors with low parasitic inductances for the
same reason.
●
RSENSE * ILOAD product is the value of the VSENSE voltage and typically is less than 1 V,
also for thermal dissipation issues on the same resistor.
●
VF is the forward voltage of the free-wheeling diode and it is usually less than 2 V.
How to reduce the below ground spike voltage
In order to reduce the below ground spike, the following action should be taken:
●
Reduce the parasitic inductances (see Figure 35).
●
Reduce the dIF/dt by slowing down the turn-off of the high side IGBT/MOSFET.
In most of application the two previous strategies result to be enough to reduce properly the
below ground spike voltage, increasing the robustness of power system and then the margin
for safe operation of the application. On the other hand in some cases, where the below
ground spike voltage is significantly higher, the above suggestions may be not sufficient to
limit that value and then it could be useful to add some external components to improve
further the noise robustness of the power stage section:
●
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Add a small resistor (2÷10 Ω typically) in series to the OUT line (see Figure 37). The
series resistor has the positive effect of limiting the spike voltage on the BOOT pin,
thanks to the filtering effect of such resistor coupled with the bootstrap capacitor. Note
that, in order to obtain an effective filtering effect, the OUT resistor must be placed
between the minus terminal of bootstrap capacitor and the output of the power stage,
as indicated in Figure 37.
Doc ID 14589 Rev 2
AN2738
The below-ground voltage on the OUT pin
Figure 36. Transient peak forward voltage vs. dIF/dt of STTH1L06 diode
Figure 37. Use of OUT resistor to limit the below ground voltage spike on OUT pin
V
BOOT
OUT
HB_OUT
0
VBOOT without Rout
VCC
DBOOT
RBOOT
t
BOOT
H.V.
CBOOT
RGH_ON
HS
HVG
DH
RGH_OFF
HB_OUT
OUT
ROUT
RGL_ON
LS
LVG
DL
RGL_OFF
Note that this resistor is in series with both the turn ON and the turn OFF path, so it must be
considered in the sizing of overall turn-ON and turn-OFF resistance.
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The below-ground voltage on the OUT pin
AN2738
In case neither the OUT resistor would not be enough to limit the below ground voltage, the
following final resolving action could be taken:
●
Add a high voltage fast diode (e.g. STTH1L06) between the GND pin and the OUT pin
(very close to the device pins) in order to clamp directly on the gate driver OUT pin the
below ground voltage spike. Note that, in any case, this diode must be used together
with the OUT resistance suggested in the previous tip, because it must be avoided that
the diode brings the very large load current during low side recirculation, in order to
increase the clamping action of the diode itself (see following Figure 38).
Figure 38. Use of combination of OUT resistor and OUT diode to limit the below
ground voltage spike on OUT pin
V
BOOT
OUT
HB_OUT
0
t
VCC
DBOOT
RBOOT
BOOT
H.V.
CBOOT
RGH_ON
HS
HVG
DH
RGH_OFF
HB_OUT
OUT
ROUT
RGL_ON
LS
LVG
DL
GND
RGL_OFF
DOUT
Note that ROUT resistor is also in series with the charging path of the bootstrap capacitor. Its
effects may be more evident during the first charge of the bootstrap capacitor, when it is
completely discharged and a significant charging current may flow into the ROUT resistor,
producing an additional voltage drop between ground and OUT pin. Because during the
bootstrap charging the HVG pin is set to low level, the OUT pin and the HVG pin are shorted
together and have the same voltage, so the voltage drop due to ROUT results directly
transferred to the VGE (or VGS) of the high side IGBT (or MOSFET). Then the risk is that a
weak turn ON of the high side power switch, when the low side power switch is already ON,
could cause a cross-conduction in the power half bridge. Actually in most of cases this
doesn’t represent an issue for the proper working of the application, because the typical
intrinsic resistance RBOOT (~120 Ω) in series to the internal bootstrap diode is much higher
than the ROUT commonly used and the voltage drop on ROUT is negligible.
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AN2738
11.3
The below-ground voltage on the OUT pin
Issues related to the below-ground voltage phenomenon
Contrary to common belief, the issues resulting from significant below-ground voltages on
the OUT pin are not related to the maximum absolute voltage values that the OUT pin is
able to withstand, but are instead related mainly to the voltage value of BOOT pin, which is
indirectly bound to the OUT pin voltage. There are two main issues to be taken into
consideration, all related to the absolute maximum ratings of the IC:
11.3.1
●
VBOOT absolute minimum voltage
●
VBOOT-VOUT absolute maximum voltage
VBOOT voltage safe operating condition
Electrically, the OUT pin could safely tolerate even many volts below ground without
problems, but the BOOT pin cannot. The absolute value of the VBOOT voltage must not go
steadily below -0.3 V, in order to avoid the turn-on of the built-in junction between BOOT pin
and the IC gate driver substrate (connected to GND), which is normally in reverse condition.
The turn-on of this junction could, in fact, cause a current large enough to damage the
device.
11.3.2
Bootstrap capacitor over-charging
Another very important consideration is bootstrap capacitor over-charging. Actually, even
before the OUT pin approaches the zero voltage value, the bootstrap diode (internal or
external) tends to tie VBOOT close to the VCC supply voltage. Since the OUT pin is below
ground, the bootstrap capacitor is overcharged through the current coming from the
bootstrap diode, and the VBOOT - VOUT voltage increases. It is very important that the
bootstrap over-charge does not exceed the recommended maximum value for the VBOOT VOUT voltage (see the relevant datasheet), as the high side floating section of the gate driver
could be damaged.
Note that a significant over-charging of the bootstrap capacitor is only possible through the
static contribution of the below-ground voltage, because the dynamic below-ground voltage
is very brief and the over-charging transition is limited by the time constant RC of the
bootstrap capacitor and by the overall resistance in series with the bootstrap diode.
In Figure 39, the over-charging phenomenon is illustrated in detail.
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The below-ground voltage on the OUT pin
AN2738
Figure 39. Bootstrap over-charging due to below-ground voltage on OUT pin
VBOOT - VOUT
VBOOT
BOOTSTRAP NETWORK
&
BELOW GROUND VOLTAGE
VCC
DBOOT
RBOOT
BOOT
Time constant of this transition:
RBOOT * CBOOT >> Tspike typically
VCC
VOUT
VBOOT - VOUT
VBOOT - VOUT
overcharged
during static BGV
befor overcharge
OUT
0V
VBGV_static
t
VBGV_spike
VBOOT - VOUT
during BGV spike
VOUT
Tspike < 100ns
typically
Note that in the L6390 IC gate driver the internal DMOS in series with the integrated
bootstrap diode is fully turned on only when LVG is ON. In fact, when the LVG is OFF, the
gate of the bootstrap DMOS is biased at VCC voltage. This means that if the VBOOT is pulled
down by the bootstrap capacitor (due to below-ground voltage on OUT pin) at about 3 V
(typ) below VCC voltage, the DMOS turns on again. Figure 40 shows some characteristics
of different (internal and external) bootstrap networks.
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The below-ground voltage on the OUT pin
Figure 40. Different bootstrap network characteristics
2*
Regarding the below-ground voltage spike, typically this undershoot voltage has a very brief
duration (less than 100 ns) and is not enough to further over-charge the bootstrap capacitor,
because the charging time constant of the bootstrap is equal to the product of the CBOOT
and the RBOOT resistance in series with the diode. Moreover, the higher this resistance, the
lower the risk of bootstrap over-charging during BVG spikes.
For example, with a RBOOT of 120 Ω and a CBOOT of 100 nF, the associated time constant
would be about 12 µs, much higher than typical below-ground voltage spike duration. Using
an internal bootstrap diode it is very difficult to over-charge the floating section up to
dangerous voltage levels with the very short duration of the undershoot spike. More
attention should be given to the below-ground voltage of the VBOOT during this spike
because, as explained previously, the internal junction VBOOT -to-substrate could turn on;
typically for very short time (a few tenths of nanoseconds), this is not a problem.
In Figure 41, the main conditions to avoid issues related to the below-ground voltage
phenomenon in steady-state are resumed. As explained in paragraph 11.1, typically the
static below-ground voltage is hardly higher than 2 V, so these constraints are usually not a
limitation for most applications.
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The below-ground voltage on the OUT pin
AN2738
Figure 41. L6390 safe operating range when the OUT pin is below ground voltage (in
steady state)
Vcc
Vcc-Vboot
Vboot
Vboot > - 0.3V
0V
Vboot-Vout < 20V
below ground
voltage on Vout
Vout
11.4
Functionality of L6390 outputs in below-ground condition
The L6390 IC gate driver makes use of a level shifter to send the set-reset information to the
high side floating section. The level shifter, shown in Figure 42, is mainly composed of two
high voltage switches, for set/reset signals, driven by the low voltage section and linked to
two pull-up resistors connected to the BOOT pin.
Figure 42. Driver functionality in below-ground voltage condition on OUT pin
Vcc
Vcc
Vcc-Vboot
Vboot
Vboot
Vboot
Level
Shifter
Vboot > 5V
Vboot-Vout
0V
for full
functionality
of level shifter
High Side
floating
section
set
Low Side
section
gnd
HVG
reset
gnd
Vout
below ground voltage on Vout
Vout
The two level-shifted signals are then fed into a logic latch in the high side floating section,
providing the logic state for the high side gate driver (HVG output). The L6390 IC provides
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The below-ground voltage on the OUT pin
full functionality of the switching operation of the high side section until the level shifter is
operating. Because the two pull-up resistors are connected to the floating supply VBOOT of
the high side floating section, the driver still functions properly if the OUT voltage begins to
go below ground. This means that the operating limit for the level shifter structure must be
referred to the absolute value of the BOOT voltage relative to ground. In fact, the BOOT
voltage can be considered the supply of the level shifter block. The minimum operating value
of the level shifter supply is 5 V, so the VBOOT value must be at least 5 V. The VBOOT - VOUT
voltage can be any value in the 12.4 V to 20 V range. If the BOOT voltage is between 0 V
and 5 V, the functionality of the level shifter is not guaranteed, but internal structures should
not be damaged. Some examples are given in the following section.
11.4.1
Steady state (DC) conditions
When the LVG is off, the bootstrap diode turns on only when the BOOT voltage is about 2 V
below the VCC (worst case: minimum voltage drop value), so it could over-charge the
bootstrap capacitor up to a maximum voltage of about VCC - 2 V - VOUT. To avoid exceeding
the value of 20 V on VBOOT - VOUT voltage, the OUT pin could be forced permanently to the
value of VCC - 2 V- 20 V with the L6390 IC still operating safely, making the HVG output
switching as the HIN logic input (see the table below).
Table 3.
Minimum VOUT in DC condition providing safe and full operation of the
high side section
Example 1
Example 2
Example 3
VCC
12.5
15
17
VBOOT
10.5
13
15
VOUT (min)
-9.5
-7
-5
VBOOT - VOUT (max)
20
20
20
It must be emphasized, as explained in the previous paragraph, how in most applications
the static below-ground voltage of the OUT pin is very rarely lower than about - 2 V.
11.4.2
Transient conditions
If the time that the OUT pin goes below ground is limited, the bootstrap over-charge
probably will not exceed the safe operating range (20 V) of VBOOT - VOUT, even if the VOUT
voltage is lower than the limit of VCC - 2 V - 20 V. In this case the gate driver and the high
side section are fully operational if the VBOOT voltage remains above 5 V, as explained
previously, for proper functioning of the high side level shifter. The logic state is anyway held
if the VBOOT remains above ground.
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The below-ground voltage on the OUT pin
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Figure 43. OUT below-ground voltage in transient conditions: limited boot overcharging
VOUT
VBOOT
VBOOT - VOUT
VCC
VBOOT - VOUT
befor overcharge
VBOOT - VOUT
VBOOT - VOUT
safe operating
maximum range
(20V)
real
overcharge
VBOOT - VOUT
virtual final value
in steady state
0V
VBGV_static
t
VBGV_spike
VOUT
High side OFF time
HIN
11.4.3
Below-ground voltage spikes
As previously stated, the duration of the dynamic contribution of the below-ground voltage
on the OUT pin is not enough to over-charge the bootstrap capacitor. This fact usually
removes the risk of exceeding the recommended maximum value of 20 V for VBOOT -VOUT
voltage, but it increases the danger of pulling down the BOOT pin below ground, thus
violating its absolute minimum rating. As mentioned above, this phenomenon should be
avoided for safe IC gate driver operation. However, note that applicative bench tests have
shown the L6390 proper operation even with below-ground spikes on the OUT pin well
above - 50 V (see Figure 44).
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The below-ground voltage on the OUT pin
Figure 44. Example of below-ground voltage spike
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Layout suggestions
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Layout suggestions
Typically, for power applications using high voltages and large load currents, the board
layout of all circuits related to the power stage is important. Board layout includes several
aspects, such as track dimensions (length and width), circuit areas, but also the proper
routing of the traces and the optimized reciprocal arrangement of the various system
elements and power sources in the PCB area.
Reasons to give particular attention to the PCB layout include EMI issues (both induced and
perceived by the application) and over-voltage spikes due to parasitic inductances along the
PCB traces, the proper connection of the sense blocks, the logic inputs and the analog
outputs of the L6390 device. In fact, the L6390 IC not only has the function of driving the
power stage, but also embeds analog sensing blocks such as comparator and op-amps. For
example, especially regarding multi-phase power stages, it is important to keep the current
reading, performed through the integrated op-amp, from ground noise.
In Figure 45 some layout guidelines and suggestions for a 3-phase application are provided.
Figure 45. Layout suggestion for a 3-phase power system
TRACKS SWITCHING WITH HIGH VOLTAGE
TRANSITIONS SHOULD BE KEPT FAR FROM
THE LOGIC AND OPAMP ANALOG LINES
MINIMIZE
THIS AREA
MINIMIZE THE LENGHT
OF THESE PATHS
H.V.
L6390
L6390
L6390
HVG
inputs
control signals
OUT
to phase 1
LIN
HIN
LP1
to phase 2
to phase 3
LP1
LP1
+
LVG
BULK
CAPACITOR
uC
+
ADC
OPOUT
GND
GND
LP2
USE RSENSE
WITH LOW
PARASITIC
INDUCTANCE
LP3
DRIVER GROUND
MINIMIZE
TO LIMIT
THE BELOW
GROUND SPIKE
ON OUT PIN
LP4
LP2
LP2
LP3
LP3
LP4
LP4
SIGNAL GROUND
POWER GROUND
MINIMIZE TO LIMIT THE NOISE
ON THE INPUT LOGIC SIGNALS AND ON THE
ANALOG OPAMP OUTPUT
LP5
NOT CRITICAL
As explained in Section 8.9, the gate driving PCB traces should be designed to be as short
as possible and the area of the circuits should be minimized to avoid the sensitivity of such
structures to the surrounding noise. Typically, a good power system layout keeps the power
IGBTs (or MOSFETs) of each half-bridge as close as possible to the related gate driver.
In Figure 45 a set of parasitic inductances related to the different circuit tracks is shown. The
various groups of inductances may have undesired effects which should be limited as much
as possible. Moreover, note that Figure 45 emphasizes parasitic inductances located on the
lines usually managing high voltages and fast current transitions, which are very noisy.
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Layout suggestions
The group of LP1, LP2 and LP3 parasitic inductances is located along the low side path of
each half-bridge, between the OUT pin and the ground of the related driver and provides an
undesired contribution to the issue of below-ground voltage spike on each OUT pin of the
L6390 device (as described in Section 11.1). In fact, at the beginning of the current
recirculation on the low side switches, the current may experience a high dI/dt which may
produce, on those parasitic inductances, significant voltage spikes. These spikes are
summed with the voltage drop of the low side diode and with the RSENSE and have a
negative sign, so the overall voltage drop between OUT and GND may be significant. The
recommendation is to limit as much as possible each contribution to this phenomenon by
limiting the length of tracks LP1, LP2 and LP3 and using an RSENSE resistor with low
intrinsic inductance. More specifically, LP1 may be reduced by connecting the OUT line
directly to the collector (or drain) of the low side IGBT (or MOSFET). LP2 may be reduced by
placing the RSENSE resistor as close as possible to the emitter (or source) of the low side
IGBT (or MOSFET). The LP3 may be minimized by connecting the ground line (also called
driver ground) of the related gate driver directly on the RSENSE resistor.
LP4 represents the parasitic inductance located between the ground connections of each
gate driver (driver ground) and the ground connection of the application controller (also
called signal ground). Due to its location, this parasitic inductance introduces noise which is
experienced by the input logic signals and the op-amp output analog signals. In fact, each
phase of the bridge causes high currents (with high dI/dt) to flow on these paths, resulting in
voltage noise which drops between the gate driver ground and the controller ground. This
noise between the two grounds is directly added to all logic and analog voltage signals
between the gate driver and the microcontroller included the input logic signals and the
analog output of the related gate driver op-amp. It is recommended to minimize this noise by
reducing the distance between the signal ground and the driver ground (for each gate driver
in the system) as much as possible. Generally, it is recommended to connect the signal
ground to the three driver grounds through a star connection, in order to improve the
balancing and symmetry of the 3-phase driving topology.
Note:
It is very important to avoid any ground loop; only a single path must connect two different
ground nodes.
The LP5 parasitic inductance is usually not critical, because it stands between the negative
terminal of the bulk capacitor and the signal/power ground.The spikes on this parasitic
element minimally influence other nodes of the system.
Another useful suggestion is to ensure some distance between the lines switching with high
voltage transitions, and the signal lines sensitive to electrical noise. Specifically, the tracks of
each OUT phase bringing significant currents and high voltages should be separated from
the logic lines and analog sensing circuits of op-amps and comparators.
In Figure 46 a practical example of the layout is provided. The example represents the 3phase power stage section of the STEVAL-IHM021V1 demonstration board. It includes the
L6390 gate drivers, the six IGBTs (or MOSFETs) in the DPAK package and the bulk
capacitor. The layout suggestions described in this paragraph are implemented in the layout
example in Figure 46. As described in Figure 46, the fixed voltage tracks, such as GND or
HV lines, can be used to shield the logic and analog lines from the electrical noise produced
by the switching lines (e.g. OUT1, OUT2 and OUT3). Each half-bridge ground is connected
in a star configuration and the three RSENSE resistors are very close to each other and to the
power ground. Note also that the suggested 3-phase configuration occupies a modest
amount of surface area (7 cm x 6 cm), and thanks to the SMD packages and board vias, no
heat sinks are required.
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Layout suggestions
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Figure 46. Layout example from the STEVAL-IHM021V1 3-phase board
7 cm
L63
90
HS
L63
90
RSENSE
LS
RSENSE
T2LS
POWER GND
RSENSE
LK
90
BU
LS
HV
CA
OUT3
P
HS
TOP LAYER
Signal lines
(logic and analog)
kept far from
switching lines (OUTx)
HV
RSENSE
OUT1
RSENSE
Fixed potential tracks
(e.g. GND, HV) shield POWER GND
signal lines from electrical
noise due to switching lines
(OUTx)
T2
RSENSE
T3
BOTTOM LAYER
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HV
OU
OU
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HS
OU
OUT1
L63
6 cm
HV
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13
Revision history
Revision history
Table 4.
Document revision history
Date
Revision
Changes
02-Oct-2008
1
Initial release
03-Aug-2009
2
Removed chapter 11 The use of the resistor on the OUT line and
added Section 11.2: How to reduce the below ground spike voltage
on page 40.
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