AN2794 Application note 1 kW dual stage DC-AC converter based on the STP160N75F3 Introduction This application note provides design guidelines and performance characterization of the STEVAL-ISV001V1 demonstration board. This board implements a 1 kW dual stage DC-AC converter, suitable for use in batterypowered uninterruptible power supplies (UPS) or photovoltaic (PV) standalone systems. The converter is fed by a low DC input voltage varying from 20 V to 28 V, and is capable of supplying up to 1 kW of output power on a single-phase AC load. These features are possible thanks to a dual stage conversion topology that includes an efficient step-up pushpull DC-DC converter, which produces a regulated high-voltage DC bus and a sinusoidal HBridge PWM inverter to generate a 50 Hz, 230 Vrms output sine wave. Other key features of the system proposed are high power density, high switching frequency and efficiency greater than 90% over a wide output load range Figure 1. January 2012 1 kW DC-AC converter prototype Doc ID 14827 Rev 2 1/39 www.st.com Contents AN2794 Contents 1 System description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2 Design considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 2.1 Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 3 Schematic description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 4 Experimental results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 6 Bibliography . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 Appendix A Component list. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 Appendix B Product technical specification . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 7 2/39 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 Doc ID 14827 Rev 2 AN2794 List of tables List of tables Table 1. Table 2. Table 3. Table 4. Table 5. Table 6. Table 7. Table 8. System specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Push-pull converter specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 HF transformer design parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 Output inductor design parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 Power MOSFET . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Diode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Bill of material (BOM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 Doc ID 14827 Rev 2 3/39 List of figures AN2794 List of figures Figure 1. Figure 2. Figure 3. Figure 4. Figure 5. Figure 6. Figure 7. Figure 8. Figure 9. Figure 10. Figure 11. Figure 12. Figure 13. Figure 14. Figure 15. Figure 16. Figure 17. Figure 18. Figure 19. Figure 20. Figure 21. Figure 22. Figure 23. Figure 24. Figure 25. Figure 26. Figure 27. Figure 28. 4/39 1 kW DC-AC converter prototype . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Block diagram of an offline UPS system. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Possible use of a DC-AC converter in standalone PV conversion . . . . . . . . . . . . . . . . . . . . 5 Block diagram of the proposed conversion scheme . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Push-pull converter typical waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Distribution of converter losses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Distribution of losses with 3 STP160N75F3s paralleled . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Component placement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Top layer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Bottom layer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Characteristic waveforms (measured at 24 V input voltage and 280 W resistive load) . . . 26 Characteristic waveforms (measured at 28 V input voltage and 1000 W resistive load) . . 26 MOSFET voltage (ch4) and current (ch3) without RC snubber . . . . . . . . . . . . . . . . . . . . . 27 MOSFET voltage (ch4) and current (ch3) with RC snubber . . . . . . . . . . . . . . . . . . . . . . . . 27 Rectifier diode current (ch3) and voltage (ch4) without RDC snubber . . . . . . . . . . . . . . . . 27 Rectifier diode current (ch3) and voltage (ch4) with RDC snubber. . . . . . . . . . . . . . . . . . . 27 Ch1, ch3 MOSFETs drain current, ch2, ch4 MOSFET drain-source voltage . . . . . . . . . . . 28 Startup, ch2, ch3 inverter voltage and current, ch4 DC bus voltage . . . . . . . . . . . . . . . . . 28 DC-DC converter efficiency with 20 V input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 DC-DC converter efficiency with 22 V input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 DC-DC converter efficiency with 24 V input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 DC-DC converter efficiency with 26 V input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 DC-DC converter efficiency with 28 V input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 Converter efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 Technical specification for 1.5 mH 2.5 A inductor L4 (produced by MAGNETICA) . . . . . . 35 Technical specification for 1 kW, 100 kHz switch mode power transformer TX1 (produced by MAGNETICA) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 Dimensional drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37 Doc ID 14827 Rev 2 AN2794 1 System description System description In a UPS system, as shown in Figure 2, a DC-AC converter is always used to convert the DC power from the batteries to AC power used to supply the load. The basic scheme also includes a battery pack, a battery charger which converts AC power from the grid into DC power, and a transfer switch to supply the load from the mains or from the energy storage elements if a line voltage drop or failure occurs. Figure 2. Block diagram of an offline UPS system DC/AC AC/DC SWITCH Battery Another application where a DC-AC converter is always required is shown in the block diagram of Figure 3. In this case, the converter is part of a conversion scheme commonly used in standalone photovoltaic systems. An additional DC-DC converter operates as a battery charger while performing a maximum power point tracking algorithm (MPPT), which is necessary to maximize the energy yield from the PV array. The battery pack is always present to store energy when solar radiation is available and release it at night or during hours of low insolation. Figure 3. Possible use of a DC-AC converter in standalone PV conversion DC/DC Battery Charger + MPPT DC/AC Batteries LC Filter Load A possible implementation of an isolated DC-AC converter, which can be successfully used in both the above mentioned applications, is given in the block diagram of Figure 4. It consists of three main sections: 1. The DC-DC converter 2. The DC-AC converter 3. The power supply section Doc ID 14827 Rev 2 5/39 System description Figure 4. AN2794 Block diagram of the proposed conversion scheme 3TEPUPSTAGE0USH0ULL )NVERTER3TAGE("RIDGE 34'7.#7$ , 340.& 48 $ $ : 6 IN # $ ? - : , $ 6 OUT , : : - ? 3' 344(2 34&LITE . ,$ 6 0OWER 3UPPLY 6 3ECTION , 34..&, !-V The DC-DC section is a critical part of the converter design. In fact, the need for high overall efficiency (close to 90% or higher) together with the specifications for continuous power rating, low input voltage range leading to high input current, and the need for high switching frequency to minimize weight and size of passive components, makes it a quite challenging design. Due to the constraints given by the specifications given in Table 1, few topology solutions are suitable to meet the efficiency target. Actually, since the input voltage of the DC-AC converter must be at least equal to 350 V, it is not feasible to use non-isolated DC-DC converters. Moreover, the output power rating prevents the use of single switch topologies such as the flyback and the forward. Among the remaining isolated topologies, the half bridge and full bridge are more suitable for high DC input voltage applications and also characterized by the added complexity of gate drive circuitry of the high side switches. Table 1. System specifications Specification Value Nominal input voltage 24 V Output voltage 230 Vrms, 50 Hz Output power 1kW Efficiency 90% Switching frequency 100 kHz (DC-DC); 16 kHz (DC-AC) Due to such considerations, the push-pull represents the most suitable choice. This topology features two transistors on the primary side and a center tapped high frequency transformer, as shown in the step-up section in Figure 4. It is quite efficient at low input voltage making it widely used in battery powered UPS applications. Both power devices are ground referenced with consequent simple gate drive circuits. They are alternatively turned 6/39 Doc ID 14827 Rev 2 AN2794 System description on and off in order to transfer power to each primary of the center tapped transformer. Contemporary conduction of both devices must be avoided by limiting the duty cycle value of the constant frequency PWM modulator to less than 0.5. The PWM modulator should also prevent unequal ON times for the driving signals since this would result in transformer saturation caused by the "Flux Walking" phenomenon. The basic operation is similar to a forward converter. In fact, when a primary switch is active, the current flows through the rectifier diodes, charging the output inductor, while when both the switches are off, the output inductor discharges. It is important to point out that the operating frequency of the output inductor is twice the switching frequency. A transformer reset circuit is not needed thanks to the bipolar flux operation, which also means better transformer core utilization with respect to single-ended topologies. The main disadvantage of the push-pull converter is the breakdown voltage of primary power devices which has to be higher than twice the input voltage. In fact, when voltage is applied to one of the two transformer primary windings by the conduction of a transistor, the reflected voltage across the other primary winding puts the drain of the off state transistor at twice the input voltage with respect to ground. This is the reason why push-pull converters are not suitable for high input voltage applications. For the above mentioned reasons, the voltage fed push-pull converter, shown in Figure 4, is chosen to boost the input voltage from 24 V to a regulated 350 V, suitable for optimal inverter operation. The high voltage conversion ratio can be achieved by proper transformer turns ratio design, taking into account that the input to output voltage transfer function is given by: Equation 1 Vout = 2 N2 DVin N1 The duty cycle is set by a voltage mode PWM regulator (SG3525) to keep a constant output DC bus voltage. This voltage is then converted into AC using a standard H-bridge converter implemented with four ultrafast switching IGBTs in PowerMESH™ technology, switching at 16 kHz. The switching strategy, based on PWM sinusoidal modulation, is implemented on an 8-bit ST7lite39 microcontroller unit. This allows the use of a simple LC circuit to obtain a high quality sine wave in terms of harmonic content. The power supply section consists of a buck-boost converter to produce a regulated 15 V from a minimum input voltage of 4 V. The circuit can be simply implemented by means of a L5973 device, characterized by an internal P-channel DMOS transistor and few external components. In this way, it is possible to supply all the driving circuits and the PWM modulator. A standard linear regulator, L7805, provides 5 V supply to the microcontroller unit. Doc ID 14827 Rev 2 7/39 Design considerations 2 AN2794 Design considerations The basic operation of a voltage fed push-pull converter is shown in Figure 5, where theoretical converter waveforms are highlighted. In practice, significant overvoltages across devices M1, M2 and across the four rectifier diodes are observed in most cases due to the leakage inductance of the high frequency transformer. As a consequence, the breakdown voltage of primary devices must be greater than twice the input voltage, and the use of snubbing and/or clamping circuits is often helpful. Special attention has to be paid to transformer design, due to the difficulties in minimizing the leakage inductance and implementing low-voltage high-current terminations. Moreover, imbalance in the two primary inductance values must be avoided both by symmetrical windings and proper printed circuit board (PCB) layout. While transformer construction techniques guarantee good symmetry and low leakage inductance values, asymmetrical layout due to inappropriate component placement can be the source of different PCB trace inductances. Whatever the cause of a difference in peak current through the switching elements, transformer saturation in voltage mode push-pull converters can occur in a few switching cycles with catastrophic consequences. Figure 5. 8/39 Push-pull converter typical waveforms Doc ID 14827 Rev 2 AN2794 Design considerations Starting from the specifications in Table 2, a step-by-step design procedure and some design hints to obtain a symmetrical layout are given below. Table 2. Push-pull converter specifications Specification Symbol Value Nominal input voltage Vin 24 V Maximum input voltage Vinmax 28 V Minimum input voltage Vinmin 20 V Nominal output power Pout 1000 W Nominal output voltage Vout 350 V Target efficiency η > 90% Switching frequency f 100 kHz A switching frequency of f = 100 kHz was chosen to minimize passive components size and weight, then the following step-by-step calculation was done: ● Switching period: Equation 2 T= ● 1 1 = 5 = 10 μs f 10 Maximum duty cycle The theoretical maximum on time for each phase of the push-pull converter is: Equation 3 t * on = 0.5T = 5 μs Since deadtime has to be provided in order to avoid simultaneous device conduction, it is better to choose the maximum duty cycle of each phase as: Equation 4 Dmax = 0.9 t * on = 0.45 T This means a total deadtime of 1μs at maximum duty cycle, occurring for minimum input voltage operation. ● Input power Assuming 90% efficiency the input power is: Equation 5 Pin = Pout = 1111 W 0.9 Doc ID 14827 Rev 2 9/39 Design considerations ● AN2794 Maximum average input current: Equation 6 Pin 1111 = = 55.55 A Vinmin 20 Iin = ● Maximum equivalent flat topped input current: Equation 7 Ipft = ● Iin 55.55 = = 61.72 A 2Dmax 0.9 Maximum input RMS current: Equation 8 IinRMS = Ipft 2Dmax = 58.55A ● Maximum MOSFET RMS current: Equation 9 IMos RMS = Ipft D max = 41.4 A ● Minimum MOSFET breakdown voltage: Equation 10 VBrk Mos = 1 . 3 • 2 • VinMax = 72 .8 V ● Transformer turns ratio: Equation 11 N= ● Vout N2 = = 19 N1 2Vinmin Dmax Minimum duty cycle value: Equation 12 Dmin = ● Vout = 0.32 2NVinmax Duty cycle at nominal input voltage: Equation 13 Dmin = ● Vout = 0.38 2NVin Maximum average output current: Equation 14 Iout = 10/39 Pout = 2.86 A Vout Doc ID 14827 Rev 2 AN2794 Design considerations ● Secondary maximum RMS current Assuming that the secondary top flat current value is equal to the average output value the rms secondary current is: Equation 15 IsecRMS = Iout Dmax = 1.91 A ● Rectifier diode voltage: Equation 16 Vdiode = NVinMax = 532 V ● Output filter inductor value: Equation 17 Lmin ≥ ( t on N2 Vin - Vout ) Max N1 ΔI Assuming a ripple current value ΔI= 15% Iout = 0.43A, the minimum value for the output filter inductance is: Equation 18 L min = 1.109 mH With this value of inductance continuous current mode (CCM) operation is guaranteed for a minimum output current of: Equation 19 IoutMin = ΔI = 0.215 A 2 which means a minimum load of 75 W is required for CCM operation. The chosen value for this design is L=1.5 mH. ● Output filter capacitor value: Equation 20 C= 1 ΔIL Ts 8 ΔV0 Considering a maximum output ripple value equal to: Equation 21 Δ V0 = 0 .1 % Vout = 0 .35 V Doc ID 14827 Rev 2 11/39 Design considerations AN2794 the minimum value of capacitance is: Equation 22 Cmin = 1.53 μF and the equivalent series resistance (ESR) has to be lower than: Equation 23 ESRmax = ● ΔV0 = 0.81 Ω ΔIL Input capacitor: Equation 24 Cin = ICrms ΔTonMax ΔVin where Icrms is the RMS capacitor current value given by: Equation 25 2 2 ICrms = IIn - Iin = 19 A Rms and Equation 26 ΔVin = 0.1%VinMax = 0.028 V then Equation 27 Cin = ICrms 12/39 ΔTonMax ΔVin Doc ID 14827 Rev 2 = 3053 μF AN2794 Design considerations ● HF transformer design The design method is based on the Kg core geometry approach. The design can be done according to the specifications in Table 3. Table 3. HF transformer design parameters Specification Symbol Value Nominal input voltage Vin 24 V Maximum input voltage Vinmax 28 V Minimum input voltage Vinmin 20 V RMS input current Iin 41.4 A Nominal output voltage Vout 350 V Output current Iout 2.86 A Switching frequency f 100 kHz Efficiency η 98% Regulation α 0.05% Max operating flux density Bm 0.05T Window utilization Ku 0.3 Duty cycle Dmax 0.45 Temperature rise Tr 30 °C The first step is to compute the transformer apparent power given by: Equation 28 Pt = P0 1 + P0 = ( + 1)V0I0 = 2021 W η η The second step is the electrical condition parameter calculation Ke: Equation 29 ( ) 2 K e = 0.145 • K 2f • f 2 • Bm 10 -4 where Kf=4 is the waveform coefficient (for square waves). Equation 30 ( ) K e = 0.145(4)2 (100 .000 )2 (0.05)2 10 -4 = 5800 The next step is to calculate the core geometry parameter: Equation 31 Kg = Pt = 0.348 cm5 2Keα Doc ID 14827 Rev 2 13/39 Design considerations AN2794 The Kg constant is related to the core geometrical parameters by the following equation: Equation 32 Kg = Wa A 2cK u MLT where Wa is the core window area, Ac is the core cross sectional area and MLT is the mean length per turn. For example, choosing an E55/28/21 core with N27 ferrite, having ● Wa= 2.8 cm2 ● Ac= 3.5 cm2 ● MLT= 11.3 cm the resulting Kg factor is: ● Kg= 0.91 cm2 which is then suitable for this application. Once the core has been chosen, it is possible to calculate the number of primary turns as follows: Equation 33 N1 = Vinmin DmaxT ΔBAc = 2 turns The primary inductance value is: Equation 34 Lp = N2AL = 4 • 5800 nH = 23.2 μH and the number of secondary turns is: Equation 35 N2 = N • N1 = 38 turns At this point wires must be selected in order to implement primary and secondary windings. At 100 kHz the current penetration depth is: Equation 36 δ= 6.62 = 0.0209 cm f Then, the wire diameter can be selected as follows: Equation 37 d = 2δ = 0.0418 cm 14/39 Doc ID 14827 Rev 2 AN2794 Design considerations and the conductor section is: Equation 38 AW = π d2 = 0.00137cm2 4 Checking the wire table we notice that AWG26, having a wire area of AWAWG26 = 0.00128 cm2, can be used in this design. Considering a current density J = 500 A/cm2 the number of primary wires is given by: Equation 39 Snp = A wp A w AWG26 = 62 where: Equation 40 Awp = Iin = 0.08 cm2 J Since the AWG26 has a resistance of 1345 μΩ/cm, the primary resistance is: Equation 41 rp = 1345μΩ / cm = 21.69μΩ / cm 62 and so the value of resistance for the primary winding is: Equation 42 Rp = N1 • MLT • rp = 490 .1 μΩ Using the same procedure, the secondary winding is: Equation 43 A ws = Iout = 0.00572 cm2 J Equation 44 S ns = A ws =5 A w AWG 26 Equation 45 rs = 1345μΩ / cm = 269μΩ / cm 5 Equation 46 R s = N 2 • MLT • rs = 115 .5 m Ω Doc ID 14827 Rev 2 15/39 Design considerations AN2794 The total copper losses are: Equation 47 PCu = Pp + Ps = RpI2in + R sI2s = 1.78W And transformer regulation is: Equation 48 α= Pcu 100 = 0.178% Pout From the core loss curve of N27 material, at 55 °C, 50mT and 100 kHz, the selected core has the following losses: Equation 49 PV = 28.1 kW m3 • Ve = 1.23 W Where Ve= 43900 mm3 is the core volume. The transformer temperature rise is: Equation 50 Tr = R th • (PCu + PV ) = 33 o C with Equation 51 R th = 11 ● o C W Output inductor The output filter inductor can be made using powder cores to minimize eddy current losses and introduce a distributed air gap into the core. The design parameters are shown in Table 4: Table 4. Output inductor design parameters Specification Symbol Value Minimum inductance value Lmin 1.5 mH DC current I0 2.86 A AC current ΔI 0.41 A Output power P0 1000 W Ripple frequency fr 200 kHz Operating flux density Bm 0.3 T Core material 16/39 Kool µ Window utilization Ku 0.4 Temperature rise Tr 25 °C Doc ID 14827 Rev 2 AN2794 Design considerations The peak current value across the inductor is: Equation 52 Ipk = I0 + ΔI = 3.06A 2 To select a proper core we must compute the LI2pk value: Equation 53 2 LIpk = 10 .3 mH • A Knowing this parameter, from Magnetics’ core chart, a 46.7 mm x 28.7 mm x 12.2 mm Kool μ toroid, with μ=60 permeability and AL = 0.086 nH/turn can be selected. The required number of turns is then: Equation 54 L = 132 turns AL N= The resulting magnetizing force (DC bias) is: Equation 55 H = 0.4π NI = 84.2 oersteds Le The initial value of turns has to be increased by dividing it by 0.8 (as shown in the data catalog) to take into account the reduction of initial permeability (μe = 39 at full load) at nominal current value. Then, the adjusted number of turns is: Equation 56 N = 165 turns The wire table shows that at 3 A the AWG20 can be used. With this choice, the maximum number of turns per layer, for the selected core, is Nlayer= 96 and the resistance per single layer is rlayer= 0.166Ω. The total winding resistance is then: Equation 57 R= N Nlayer rlayer = 0.38Ω and the copper losses are: Equation 58 Pcu = RI 2o = 3 .1 W The core losses can be evaluated as follows: Doc ID 14827 Rev 2 17/39 Design considerations AN2794 Equation 59 PL = kB 2ac.12 f1.23 = 2.047mW / g Equation 60 B ac = ( ) ΔI μ e 10 -4 2 = 0.0137T MPL 0.4πN where MPL=11.8 cm is the magnetic path length. Since the core weight is 95.8 g, the core losses are: Equation 61 PL = 0 .2 W ● Analysis of the converter losses Once the transformer has been designed, the next step in performing the loss analysis is to choose the power devices both for the input and output stage of the push-pull converter. According to the calculations given above the following components have been selected: Table 5. Power MOSFET Device Type RDS(on) tr+tf Vbr Id at 100 °C STP160N75F3 Power MOSFET 4.5 mΩ 70 ns+15 ns 75 V 96 A Table 6. Diode Device Type VF at 175 °C trrMax VRRM IF at 100 °C STTH8R06 Ultrafast diode 1.4 V 25 ns 600 V 8A MOSFET and diode losses can be separated into conduction and switching losses which can be estimated, in the worst case operating condition (junction temperature of 100 °C), with the following equations: Equation 62 Pcond = 1.6R ds ON I2MosRMS = 12.5W Equation 63 Pgate = Q g Vgs f = 0.165 W Equation 64 Psw(ON+ OFF) = 18/39 1 VOffImos (tr + t f ) = 8.5 W 2 T Doc ID 14827 Rev 2 AN2794 Design considerations Equation 65 Pcond Diode = VFIsecRMS = 2.67 W Equation 66 Pdiode SW = VRMIRR t b f = 2 .4 W Note: Assuming: tB= trr/2, VRM= 350 V Converter losses are distributed according to the graphic in Figure 6, where PCB trace losses and control losses are not considered. What is important to note is that primary switch conduction accounts for 36% of total DC-DC converter losses. This contribution can be reduced by paralleling either two or three power devices. For example, by paralleling three STP160N75F3s, a reduction in MOSFET conduction losses of 33% is achieved. Thus MOSFET conduction losses account for 16% of total DC-DC converter losses, resulting in a 1.8% efficiency improvement. Figure 6. Distribution of converter losses 4% 5% 14% 36% 16% 25% MOSFET cond. Losses MOSFET sw. Losses Diode cond. Losses Diode sw. Losses Transformer Losses Inductor Losses Doc ID 14827 Rev 2 AM00627v1 19/39 Design considerations Figure 7. AN2794 Distribution of losses with 3 STP160N75F3s paralleled 6% 6% 16% 18% 33% 21% MOSFET cond. Losses MOSFET sw. Losses Diode cond. Losses Diode sw. Losses Transformer Losses Inductor Losses AM00628v1 2.1 Layout considerations Because of the high power level involved with this design, the parasitic elements must be reduced as much as possible. Proper operation of the push-pull converter can be assured through geometrical symmetry of the PCB board. In fact, geometrical symmetry leads to electrical symmetry, preventing a difference in the current values across the two primary windings of the transformer which can be the cause of core saturation. The output stage of the converter has also to be routed with a certain degree of symmetry even if in this case the impact of unwanted parasitic elements is lower because of lower current values with respect to the input stage. In Figure 8, Figure 9 and Figure 10, a symmetrical layout designed for the application is shown. 20/39 Doc ID 14827 Rev 2 AN2794 Design considerations Figure 8. Component placement AM00629v1 Figure 9. Top layer AM00630v1 Doc ID 14827 Rev 2 21/39 Design considerations AN2794 Figure 10. Bottom layer AM00631v1 To obtain geometrical symmetry the HF transformer has been placed at the center of the board, which has been developed using double-sided, 140 μm FR-4 substrate with 135 x 185 mm size. In addition, this placement of the transformer is the most suitable since it is the bulkiest part of the board. Both the primary and secondary AC current loops are placed very close to the transformer in order to reduce their area and consequently their parasitic inductances. For this reason the MOSFET and rectifier diodes lie at the edges of the PCB. Input loop PCB traces show identical shapes to guarantee the same values of resistance and parasitic inductance. Also the IGBTs of the inverter stage lie at one edge of the board. This gives the advantage of using a single heat sink for each group of power components. The output filter is placed on the right side of the transformer, between the bridge rectifier and the inverter stage. The power supply section lies on the left side of the transformer, simplifying the routing of the 15 V bus dedicated to supply all the control circuitry. 22/39 Doc ID 14827 Rev 2 AN2794 3 Schematic description Schematic description The schematic of the converter is shown in Figure 11. Three MOSFETs are paralleled in order to transfer power to each primary winding of the transformer. Both RC and RCD networks can be connected between the drain and source of the MOSFETs to reduce the overvoltages and voltage ringing caused by unclamped leakage inductance. The output of the transformer is rectified by a full bridge of ultrafast soft-recovery diodes. An RCD network is connected across the rectifier output to clamp the diode voltage to its steady state value and recover the reverse recovery energy stored in the leakage inductance. This energy is first transferred to the clamp capacitor and then partially diverted to the output through a resistor. The IGBT full bridge is connected to the output of the push-pull stage. Their control signals are generated by an SG3525 voltage mode PWM modulator. Its internal clock, necessary to generate the 100 kHz modulation, is set by an external RC network. The PWM output stage is capable of sourcing or sinking up to 100 mA which can be enough to directly drive the gate of the MOSFETs devices. The PWM controller power dissipation, given by the sum of its own power consumption and the power needed to drive six STP160N75F3s at 100 kHz, can be evaluated with the following equation: Equation 67 PContoller tot = 6Q g fVdrive + VsIs = 1.3W where Vs and Is are the supply voltage and current. Since this power dissipation would result in a high operating temperature of the IC, a totem pole driving circuit has been used to handle the power losses and peak currents, achieving a more favorable operating condition. This circuit was implemented by means of an NPNPNP complementary pair of BJT transistors. The control and driver stage schematic is shown in Figure 11. Doc ID 14827 Rev 2 23/39 & & X9 X9 5 N Doc ID 14827 Rev 2 6IN66 & X X)9HOHF & 5 N 5 N 287$ *1' 9& 287% /' & S &203 6'2:1 8 9&& 287 *1' 6<1& 95() ,1+ )% &203 6* 66 ',6&+$5 57 &7 26& 6<1& & Q) 5 N & Q 1 0 X & 3:0% 3:0$ : 5 : 5 ' & *1' 9287 /'A3DN 9,1 8 5 N & Q 9 5 N *$7(% & X9 & X) 5(6(7 5 N & Q9 &21 & Q 3$ 3$ 9287 3:0/2:+,*+ 3:0/2:+,*+ 9 *$7(,*%7/2: &21 6*1' &,1 ',$* 9&& +,1 6' /,1 ,& 6*1' &,1 ',$* 9&& +,1 6' /,1 67)/,7(B62,&B3 966 5(6(7 26&&/.,1 26& &V 9'' & 677+/ 9 9 677+/ 5 3$ 3$ 5(6(7 & Q ' ' %$7 ' %$7 5*$7(,*%7/2: 5 5 5*$7(,*%7+,*+ ' 5 &21 - 6285&(,*%7/2: *$7(,*%7/2: 6285&(,*%7+,*+ *$7(,*%7+,*+ 6285&(,*%7/2: *$7(,*%7/2: 6285&(,*%7+,*+ *$7(,*%7+,*+ %$7 ' %$7 5*$7(,*%7+,*+ ,*%7/2: 67*:1&:' &V ,*%7+,*+ 67*:1&:' 5*$7(,*%7/2: ' & 3%66$,1 3%6&.$,1 3%0,62$,1 3%026,$,1 3%&/.,1$,1 3%$,1 3%$,1 *1' /9* 1& 1& 287 +9* 9%227 /' *1' /9* 1& 1& 287 +9* 9%227 /' 6285&(,*%7/2: *$7(,*%7/2: &21 9287$& 6285&(,*%7+,*+ *$7(,*%7+,*+ ,*%7/2: 67*:1&:' ,& '5$,1,*%7+,*+ ,*%7+,*+ 67*:1&:' 3$/7,& 3$$7,& 3$$73:0 3$$73:0 3$$73:0 3$$73:0,&&'$7$ 3$0&2,&&&/.%5($. 3$ 8 9287$& 6285&(,*%7+,*+ *$7(,*%7+,*+ '5$,1,*%7+,*+ 6285&(,*%7/2: &21 9287 3:0/2:+,*+ & Q9 9287 9 3:0/2:+,*+ & X9 : 5 & X)9 : 5 P+ $ & X9 9 9R 9 4 6% 4 6' 9 9 677+5 ' *$7($ 4 6% 4 6' ' ' ' & & & X)9 X)9X)9 1 ' 4 6711)/ X+$ 0 & X9(/(& & X)9(/(& & Q 9 9 / 3:0$ ' 3:0% 9, 95() ,1 0 5 Q9 9LQ 5 Q9 5 N X9 & Q 6731) 5 5 ,1 5 N 9UHI 5 N & 5 N *$7(% 0 5 8 5 *$7($ : & S 6731) 0 6731) 6731) 5 5 . 6731) 5 /V : & Q 6731) 677+5 5 .: /S /S 7; /P X+ 11 677+5 677+5 &21 *1' & Q &21 9,1 677+5 X9 9287 75$)20$*1(7,&$ / Q9 0 24/39 Q9 9LQ Schematic description AN2794 Figure 11. Schematic !-V AN2794 Schematic description The PWM modulation of the H-bridge inverter is implemented on an ST7lite39 microcontroller connected to the gate drive circuit composed of two L6386, as shown in the schematic in Figure 11. The auxiliary power supply section consists of an L5973D and an L7805, used to implement a buck-boost converter to decrease the battery voltage from 24 V to 15 V and from 15 V to 5 V respectively. Doc ID 14827 Rev 2 25/39 Experimental results 4 AN2794 Experimental results Typical voltage and current waveforms of the DC-AC converter and the efficiency curves of the push-pull DC-DC stage, measured at different input voltages, are shown below. In particular, Figure 12 and Figure 13 show both input and output characteristic waveforms of the DC-DC converter both in light load and full load condition. The HF transformer leakage inductance, which is about 1% of the magnetizing inductance, is the cause of severe ringing across the input and the output power devices. MOSFETs voltage and current waveforms with and without the connection of a snubber network are shown in Figure 14 and 15, while Figure 16 and 17 show the effect of the RCD clamp circuit connected across the rectifier bridge output. In Figure 18 the current and the voltage across one of the three parallel-connected MOSFETs, powering each of the two windings of the transformer are shown, while in Figure 19 it is possible to observe the variation of the inverter output voltage and current together with the DC-DC converter bus voltage. In Figure 20, 21, 22, 23 and 24, the efficiency curves of the push-pull converter measured with an RL load are given. A maximum efficiency above 93% has been measured at nominal input voltage and 640 W output power. The minimum value of efficiency has been tested under low load and maximum input voltage. In Figure 25, the efficiency of the whole board is shown. The efficiency tests have been carried out connecting an RL load at the inverter output connectors, with 3 mH output inductor. 26/39 Figure 12. Characteristic waveforms (measured at 24 V input voltage and 280 W resistive load) Figure 13. Characteristic waveforms (measured at 28 V input voltage and 1000 W resistive load) Ch1 and Ch2: MOSFETs drain source voltage; Ch4: HF transformer output voltage; Ch3: filter inductor current Ch1 and Ch2: MOSFETs drain source voltage; Ch3: filter inductor current Doc ID 14827 Rev 2 AN2794 Experimental results Figure 14. MOSFET voltage (ch4) and current (ch3) without RC snubber Figure 15. MOSFET voltage (ch4) and current (ch3) with RC snubber Figure 16. Rectifier diode current (ch3) and voltage (ch4) without RDC snubber Figure 17. Rectifier diode current (ch3) and voltage (ch4) with RDC snubber Doc ID 14827 Rev 2 27/39 Experimental results AN2794 Figure 18. Ch1, ch3 MOSFETs drain current, ch2, ch4 MOSFET drain-source voltage Figure 19. Startup, ch2, ch3 inverter voltage and current, ch4 DC bus voltage Figure 20. DC-DC converter efficiency with 20 V input Figure 21. DC-DC converter efficiency with 22 V input 1 Efficiency 1 Efficiency 0.95 0.9 0.85 0.95 0.9 0.85 0.8 0.8 0 200 400 600 800 1000 0 1200 200 400 600 800 1 1 0.95 0.95 Efficiency Efficiency Figure 23. DC-DC converter efficiency with 26 V input 0.9 0.85 0.8 0.9 0.85 0.8 0 200 400 600 800 1000 1200 0 200 400 600 800 1000 1200 Output Power [W] Output Power [W] AM00638v1 28/39 1200 AM00637v1 AM00636v1 Figure 22. DC-DC converter efficiency with 24 V input 1000 Output Power [W] Output Power [W] Doc ID 14827 Rev 2 AM00639v1 AN2794 Experimental results Figure 24. DC-DC converter efficiency with 28 V input Figure 25. Converter efficiency 92 Effciency % Efficiency 93 0.95 0.9 0.85 0.8 91 90 89 88 0.75 0 200 400 600 800 1000 1200 Output Power [W] 87 0 200 400 600 800 1000 Output Power [W] AM00640v1 Doc ID 14827 Rev 2 AM00641v1 29/39 Conclusion 5 AN2794 Conclusion The theoretical analysis, design and implementation of a DC-AC converter, consisting of a push-pull DC-DC stage and a full-bridge inverter circuit, have been evaluated. Due to the use of the parallel connection of three STP160N75F3 MOSFETs the converter shows good performance in terms of efficiency. Moreover the use of an ST7lite39 8-bit microcontroller allows achieving simple control of the IGBTs used to implement the DC-AC stage. Any additional feature, such as regulation of the AC output voltage or protection requirements, can simply be achieved with firmware development. 6 30/39 Bibliography 1. Power Electronics: Converters, Applications and Design 2. Transformer and Inductor Design Handbook, Second Edition 3. Magnetic Core Selection for Transformers and Inductors, Second Edition 4. Switching Power Supply Design. New York. Doc ID 14827 Rev 2 AN2794 Component list Appendix A Table 7. Component list Bill of material (BOM) Component Part value Description Cs1 100 nF, 630 V Polip. cap., MKP series EPCOS Cs2 100 nF, 630 V Polip. cap., MKP series EPCOS C1 100 nF, 50 V X7R ceramic cap.., B37987 series EPCOS C2 100 nF, 50 V X7R ceramic cap., B37987 series EPCOS C57 100 nF, 50 V X7R ceramic cap., B37987 series EPCOS C59 100 nF, 50 V X7R ceramic cap., B37987 series EPCOS C10 47 µF, 35 V SMD tantalum capacitor TAJ series C11 4.7 nF, 25 V SMD multilayer ceramic capacitor C12 100 µF, 25 V SMD X7R ceramic cap. C3225 series; size 1210 TDK C14 47 µF, 35 V SMD tantalum capacitor TAJ series AVX C16 100 pF, 25 V SMD multilayer ceramic capacitor C41 100 pF, 50 V General purpose ceramic cap., radial C17 680 nF, 25 V SMD multilayer ceramic capacitor C18 22 µF, 25 V Electrolytic cap FC series Panasonic C19 22 µF, 25 V Electrolytic cap. FC series Panasonic C26 2.2 µF, 25 V X7R ceramic cap., B37984 series EPCOS C31 2.2 µF, 25 V X7R ceramic cap., B37984 series EPCOS C28 470 nF, 25 V X7R ceramic cap., B37984 series EPCOS C33 470 nF, 25 V X7R ceramic cap., B37984 series EPCOS C34 33 µF, 450 V Electrolytic cap. B43821 series EPCOS C35 33 µF, 450 V Electrolytic cap. B43821 series EPCOS C37 3900 µF, 35 V Elec. capacitor 0.012 Ω, YXH series Rubycon C38 3900 µF, 35 V Elec. capacitor 0.012 Ω, YXH series Rubycon C39 150 µF, 35 V Electrolytic cap. fc series C40 22 nF, 50 V General purpose ceramic cap., radial C42 100 µF, 25 V Electrolytic cap. fc series Panasonic C51 100 µF, 25 V Electrolytic cap.fc series Panasonic C52 100 µF, 25 V Electrolytic cap.fc series Panasonic C53 2.2 µF, 450 V Elcrolytic capactor B43851 series EPCOS C54 4.7 nF, 100 V Polip. cap., MKT series EPCOS C55 4.7 nF, 100 V Polip. cap., MKT series EPCOS C56 470 nF, 50 V X7R ceramic cap., B37984 series EPCOS Doc ID 14827 Rev 2 Supplier AVX muRata muRata AVX muRata Panasonic AVX 31/39 Component list Table 7. AN2794 Bill of material (BOM) (continued) Component Part value Description Supplier C58 0.33 µF, 50 V X7R ceramic cap., B37984 series EPCOS C60 150 nF, 50 V SMD multilayer ceramic capacitor muRata D1 STTH8R06D Ultrafast high voltage rectifier; TO-220AC STMicroelectronics D2 STTH8R06 D Ultrafast high voltage rectifier; TO-220AC STMicroelectronics D3 STTH8R06 D Ultrafast high voltage rectifier; TO-220AC STMicroelectronics D4 STTH8R06 D Ultrafast high voltage rectifier; TO-220AC STMicroelectronics D13 STTH8R06 D Ultrafast high voltage rectifier; TO-220AC STMicroelectronics D5 BAT46 Small signal Schottky diode; SOD-123 STMicroelectronics D6 BAT46 Small signal Schottky diode; SOD-123 STMicroelectronics D8 BAT46 Small signal Schottky diode; SOD-123 STMicroelectronics D7 BAT46 Small signal Schottky diode; SOD-123 STMicroelectronics D9 STTH1L06 Ultrafast high voltage rectifier; DO-41 STMicroelectronics D10 STTH1L06 Ultrafast high voltage rectifier; DO-41 STMicroelectronics D11 1N5821 Schottky rectifier; DO-221AD STMicroelectronics D12 1N5821 Schottky rectifier; DO-221AD STMicroelectronics VOUT AC 1 CON1 FASTON RS components VOUT AC 2 CON1 FASTON RS components VOUT - CON1 FASTON RS components VOUT + CON1 FASTON RS components VIN CON1 FASTON RS components GND CON1 FASTON RS components IC1 L6386D High-voltage high and low side driver; dip-14 STMicroelectronics IC2 L6386D High-voltage high and low side driver; dip-14 STMicroelectronics IGBT LOW 1 STGW19NC60WD N-channel 19 A - 600 V TO-247 PowerMESH™ IGBT STMicroelectronics IGBT HIGH 1 STGW19NC60WD N-channel 19 A - 600 V TO-247 PowerMESH™ IGBT STMicroelectronics IGBT LOW 2 STGW19NC60WD N-channel 19 A - 600 V TO-247 PowerMESH™ IGBT STMicroelectronics IGBT HIGH 2 STGW19NC60WD N-channel 19 A - 600 V TO-247 PowerMESH™ IGBT STMicroelectronics J1 CON10 L3 150 µH, 3 A L4(1) 1174.0018 ST04 M1 STP160N75F3 N-channel 75 V - 3.5 mΩ 120 A TO-220 STripFET™ Power MOSFET STMicroelectronics M2 STP160N75F3 N-channel 75 V - 3.5 mΩ 120 A TO-220 STripFET™ Power MOSFET STMicroelectronics M3 STP160N75F3 N-channel 75 V - 3.5 mΩ 120 A TO-220 STripFET™ Power MOSFET STMicroelectronics 32/39 10-way idc connector commercial box header series Power use SMD inductor; SLF12575T series 1.5 mH, filter inductor Doc ID 14827 Rev 2 Tyco Electronics TDK MAGNETICA AN2794 Table 7. Component list Bill of material (BOM) (continued) Component Part value M4 STP160N75F3 N-channel 75 V - 3.5 mΩ 120 A TO-220 STripFET™ Power MOSFET STMicroelectronics M5 STP160N75F3 N-channel 75 V - 3.5 mΩ 120 A TO-220 STripFET™ Power MOSFET STMicroelectronics M6 STP160N75F3 N-channel 75 V - 3.5 mΩ 120 A TO-220 STripFET™ Power MOSFET STMicroelectronics Q8 STN4NF03L N-channel 30 V , 6.5 A SOT-223 STripFET™ II Power MOSFET STMicroelectronics Q9 2SD882 NPN Power BJT 30 V, 3 A transistor- SOT-32 STMicroelectronics Q10 2SD882 NPN Power BJT 30 V, 3 A transistor- SOT-32 STMicroelectronics Q11 2SB772 NPN Power BJT 30 V, 3 A transistor - SOT-32 STMicroelectronics Q12 2SB772 NPN Power BJT 30 V, 3 A transistor - SOT-32 STMicroelectronics RGATE IGBT LOW 1 100 SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components RGATE IGBT HIGH 1 100 SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components RGATE IGBT LOW 2 100 SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components RGATE IGBT HIGH 2 100 SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components R7 390 kΩ SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components R9 5.6 kΩ SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components 12 Ω SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components 10 Ω SMD standard film res - 1/8 W - 1% - 100 ppm/°C BC components R81 22 kΩ Standard film res - 1/4 W 5%, axial 05 T-Ohm R82 3.3 kΩ Standard film res - 1/4 W 5%, axial 05 T-Ohm R83 39 kΩ Standard film res - 1/4 W 5%, axial 05 T-Ohm R87 10 kΩ SMD standard film res - 1/8 W - 1% - 100ppm/°C R20 Description Supplier R21 R22 R23 R24 R25 R99 R100 R101 R102 R103 R104 Doc ID 14827 Rev 2 BC components 33/39 Component list Table 7. Component AN2794 Bill of material (BOM) (continued) Part value Description Supplier R88 R89 R90 10 kΩ SMD standard film res - 1/8 W - 1% - 100ppm/°C BC components R93 1.5 kΩ SMD standard film res - 1/8 W – 1% - 100ppm/°C BC components R94 470 Ω High voltage 17 W ceramic resistor sbcv type Meggit CGS R95 470 Ω High voltage 17 W ceramic resistor sbcv type Meggit CGS 10 Ω Standard film res – 2 W 5%, axial 05 T-Ohm 47 kΩ Standard film res - 1/4 W 5%, axial 05 T-Ohm R91 R92 R96 R97 R98 (2) TX1 1356.0004 rev.01 Power transformer MAGNETICA U1 SG3525 Pulse width modulator SO-16 (narrow) STMicroelectronics U16 L5973D 2.5 A switch step down regulator; HSOP8 STMicroelectronics U17 ST7FLITE39F2 8-bit microcontroller; SO-20 STMicroelectronics U20 L7805 124 HEAT SINK Part n. 78185, S562 cooled package TO-220; thermal res. 7.52 °C/W at length 70 mm width 40 mm height 57 mm Aavid Thermalloy HEAT SINK Part n. 78350, SA36 cooled package TO-220; thermal res. 1.2°C/W at length 135 mm width 49.5 mm height 85.5 mm Aavid Thermalloy 125 126 Positive voltage regulator; D2PAK 1. The technical specification for this component is provided in Figure 26. 2. The technical specification for this component is provided in Figure 27. 34/39 Doc ID 14827 Rev 2 STMicroelectronics AN2794 Product technical specification Appendix B Product technical specification Figure 26. Technical specification for 1.5 mH 2.5 A inductor L4 (produced by MAGNETICA) TYPICAL APPLICATION TECHNICAL DATA INDUCTOR FOR DC/DC CONVERTERS AS BUCK, BOOST E INDUCTANCE BUCK-BOOST CONVERTERS. ALSO SUITABLE IN HALF (MEASURE 1KHZ, TA 20°C) BRIDGE, PUSH-PULL AND FULL-BRIDGE APPLICATIONS RESISTANCE SCHEMATIC 1.5mH ±15% 0.52 max (MEASURE DC, TA 20°C) 800 VP MAX OPERATING VOLTAGE (F 100K HZ, IR 2.5A, TA 20°C) 1 2.5 A MAX OPERATING VOLTAGE (MEASURE DC 800 VP, TA 20°C) 4.5 A NOM SATURATION CURRENT (MEASURE DC, L 50%NOM, TA 20°C) SELF -RESONANT FREQUENY 1MHZ NOM (TA 20°C) -10°C÷+45°C OPERATING TEMPERATURE RANGE 3 (IR 2.5 A MAX) 45X20 H46mm 78g CIRCA DIMENSIONS WEIGHT INDUCTANCE VS CURRENT INDUCTANCE VS FREQUENCY 250% L/L(1kHz) 100% L 200% 150% 100% 50% 10% 0% 0 1 2 3 4 5 I [A] 6 0 DIMENSIONAL DRAWING 200 400 600 800 1000 f [kHz] BOTTOM VIEW (PIN SIDE) 20 max 45 max 12.7 2 3 30.48 46 max 4 3 min 1 10.16 1 DIMENSIONS IN MM, DRAWING NOT IN SCALE 2 2 3 0.8 (X4), RECOMMENDED PCB HOLE Doc ID 14827 Rev 2 1.2 (X4) 35/39 Product technical specification AN2794 Figure 27. Technical specification for 1 kW, 100 kHz switch mode power transformer TX1 (produced by MAGNETICA) T YPICAL APPLICATION T ECHNICAL DATA - TRANSFORMER TO POWER APPLICATIONS WITH HALF BRIDGE , PUSH -PULL E FULL -BRIDGE TYPOLOGY . I NDUCTANCE ( MEASURE 1 K H Z , T A 20°C) SCHEMATIC PIN PIN PIN 1 2 ) 17.2 uH MIN 17.2 uH MIN 5.7 mH MIN ) 6 mΩ MAX 6 mΩ MAX 90 mΩ MAX IN CC R ESISTANCE 13 ( MEASURE D . C , T A 20°C) PIN PIN PIN 3 4 5 1,2 – 3,4,5 3,4,5 – 6,7 9 – 13 (10-12 12 10 1,2 – 3 , 4 , 5 3,4,5 – 6 , 7 9 – 13 (10-12 IN CC TRANSFORMER RATIO ( MEASURE 10 K H Z , 10-12 IN CC , T A 20°C) PIN PIN 13 – 9 ⇔ 1,2 – 3,4,5 13 – 9 ⇔ 3,4,5 – 6,7 L EAKAGE I NDUCTANCE 6 7 ( MEASURE 9-13, 1-2-3-4-5-6-7 9 AND 10-12 IN C . C , F 18 ± 5% 18 ± 5% 0.11 % NOM 10 K H Z , T A 20°C) OPERATING VOLTAGE 8 0 0 V P MAX ( MEASURE 13-9, 10-12 IN CC , F 100 K H Z , D UTY C YCLE 0.8,T A 20°C) OPERATING CURRENT ( MEASURE 13-9 WITH 1-2-3-4-5-6-7 P MAX 1 K W ,F 100 K H Z , T A 20°C) PRODUCT PICTURE 2 . 5 A MAX IN CC , OPERATING FREQUENCY 100 K H Z NOM OPERATING TEMPERATURE RANGE -10°C ÷+45°C (P MAX 1 K W , T A 20°C) (P MAX 1 K W, F 100 K H Z ) I INSULATION CLASS ( P MAX 1 K W, T A 20°C ) P RIMARY TO SECONDARY INSULATION (F 50H Z , DURATION TEST 2500V 2”, T A 20°C) MAXIMUM DIMENSIONS 57X57H45 mm W EIGHT PIN (*) 1A 2A 3B 4B 5B 6C 7C FUNCTION 36/39 PIN DESCRIPTION PIN (*) P RIMARY DRAIN A P RIMARY DRAIN A 8 9 10 D 11 12D 13 14 P RIMARY +V B 24V P RIMARY DRAIN B P RIMARY DRAIN B (*)P IN WITH THE SAME SUBSCRIPT MU 2 9 2 g CIRCA ST BE CONNECTED TOGETHER ON PCB Doc ID 14827 Rev 2 FUNCTION NOT USED SECONDARY GROUND INTERMEDIARY S ECONDARY ACCESS MISSING , REFERENCE TO PCB ASSEMBLING S ECONDARY ACCESS S ECONDARY 400V 2.5A INTERMEDIARY NOT USED AN2794 Product technical specification Figure 28. Dimensional drawing 56.5 max 1356.0004 SMT 1kW 100kHz MAGNETICA 08149 1.0, Recommended PCB hole 1.4 14 13 12 4 10 9 8 55.5 max 8 7 3 min 40 5 8 7 MISSING PIN REFERENCE AS PCB ASSEMBLING 1 14 BOTTOM VIEW ( PIN SIDE ) Doc ID 14827 Rev 2 37/39 Revision history 7 AN2794 Revision history Table 8. 38/39 Document revision history Date Revision Changes 16-Feb-2009 1 Initial release 13-Jan-2012 2 – Introduction modified – Section 3 modified Doc ID 14827 Rev 2 AN2794 Please Read Carefully: Information in this document is provided solely in connection with ST products. STMicroelectronics NV and its subsidiaries (“ST”) reserve the right to make changes, corrections, modifications or improvements, to this document, and the products and services described herein at any time, without notice. All ST products are sold pursuant to ST’s terms and conditions of sale. Purchasers are solely responsible for the choice, selection and use of the ST products and services described herein, and ST assumes no liability whatsoever relating to the choice, selection or use of the ST products and services described herein. No license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted under this document. If any part of this document refers to any third party products or services it shall not be deemed a license grant by ST for the use of such third party products or services, or any intellectual property contained therein or considered as a warranty covering the use in any manner whatsoever of such third party products or services or any intellectual property contained therein. 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Resale of ST products with provisions different from the statements and/or technical features set forth in this document shall immediately void any warranty granted by ST for the ST product or service described herein and shall not create or extend in any manner whatsoever, any liability of ST. ST and the ST logo are trademarks or registered trademarks of ST in various countries. Information in this document supersedes and replaces all information previously supplied. The ST logo is a registered trademark of STMicroelectronics. All other names are the property of their respective owners. © 2012 STMicroelectronics - All rights reserved STMicroelectronics group of companies Australia - Belgium - Brazil - Canada - China - Czech Republic - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan Malaysia - Malta - Morocco - Philippines - Singapore - Spain - Sweden - Switzerland - United Kingdom - United States of America www.st.com Doc ID 14827 Rev 2 39/39

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