AN3410 Application note A 93% efficient LED driver solution for the US market Introduction This application note describes an LED driver that meets present requirements for the US market. It utilizes ST’s L6564 power factor controller in an unconventional circuit to regulate the input power to a step-down switching regulator. The circuit also compensates for variations in LED voltage drop, to maintain the average output current in a tight band over a wide range of line voltage and LED characteristics. While the input current waveform is not perfectly sinusoidal, power factor and harmonic content are well within the requirements for the US commercial market. The form factor was designed to fit into the PAR38 envelope the driver and LEDs can be used to replace 65 W incandescent floodlamps. Figure 1. ■ Specifications: – Output current 350 mA +/-3% over 90 V-138 V line range – Load: 18 series-connected 1 W LEDs – Efficiency > 93% – Power factor > 0.97 – Dimmer safe – Non-isolated ■ ST devices: – L6564 transition-mode PFC controller – STD5NM50 FET – STTH1R04A fast recovery diode – TS321AILT low power op amp Physical envelope !-V Figure 2. STEVAL-ILL041V1 demonstration board picture September 2011 Doc ID 018890 Rev 1 1/28 www.st.com Contents AN3410 Contents 1 Schematic diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 2 Circuit description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.1 Power components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.2 Power factor controller operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.3 Controlling the LED average current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 2.4 Setting the “DC” operating point . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.5 Designing the “DC” control loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 3 Control loop dynamics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 4 Performance with LED loads . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 5 Graphical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 6 Waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 7 Startup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 8 Component stress . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 9 Thermal stress . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 10 Conducted EMI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 11 PC layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 12 Bill of materials . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 13 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 14 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 2/28 Doc ID 018890 Rev 1 AN3410 List of figures List of figures Figure 1. Figure 2. Figure 3. Figure 4. Figure 5. Figure 6. Figure 7. Figure 8. Figure 9. Figure 10. Figure 11. Figure 12. Figure 13. Figure 14. Figure 15. Figure 16. Figure 17. Figure 18. Figure 19. Figure 20. Figure 21. Figure 22. Figure 23. Physical envelope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 STEVAL-ILL041V1 demonstration board picture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Circuit schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Constant-power V-I curve and linear approximation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Linear approximation error . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Input current compensated for 3 levels of output power . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Error for 3 levels of load power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 LED current vs. line voltage for 18 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 Power factor vs. line voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 Efficiency vs. line voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 Power loss vs. line voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 Waveforms at 96 V line. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 Waveforms at 108 V line. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 Waveforms at 120 V line. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 Waveforms at 132 V line. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 Startup waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Component electrical stress . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Conducted EMI, line 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Conducted EMI, line 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Top side foil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Top side placement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Bottom side layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Bottom side placement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Doc ID 018890 Rev 1 3/28 4/28 , M(Y! & ! Doc ID 018890 Rev 1 I 2 + 2 + 2 + 0&#?/+ ).6 #-0 0EAKSAMPLEHOLD &" 6FF # U& 6IN6 6IN6 6FF 3OURCE 'ATEDRV 2 /HM*UMPER 'ATE 0EAK,INE6OLTAGE MINUS,%$VOLTAGE --3$ $ 2 :#$ 2 + 2 34$.- 1 $RAIN + 2 # U&6 2 + /! + 2 /! 2 + + 6 43!),4 2 /54 + 5 43!),4 + N& # -5,4 0&#?/+ #3 6 6 /P!MP 6MINSTART '.$ 6MAXKILL 6CC U& '$ , # :#$ U& #-0 # U&6 + 2 # (6 ).6 5 )NSTANTANEOUS,INE6OLTAGEMINUS,%$6OLTAGE 7 7 6!#$IMMABLE I (6 -ULT 2 + / # U&6 7 ,%$STRINGS M! Circuit schematic N&6 $ --3$ 2EFLECTED,%$6OLTAGE (6 I 2 + Figure 3. # 6CC "OOTSTRAP0OWER /54.%' #RAMER% )NDUCTOR , # U&6 7 Schematic diagram I , M(Y! + 2 --3$ 2 U(!PK + I(6 4/03)$%'2/5.$0,!.% $ I N&6 # + 2 (6 344(2! $ 2 I ).'.$ 2 + I 1 (6 (6 6 !# + 2 I (6 (6 "RIDGE! 6 !# "2 2 + -OSTPOS Schematic diagram AN3410 !-V AN3410 Circuit description 2 Circuit description 2.1 Power components C7, L2, and L3 provide filtering for conducted EMI. Bridge rectifier BR1 feeds the step-down (buck) switching regulator. The regulator appears inverted – the flywheel diode, D1, is connected to the positive rail instead of the negative. Q2 pulls the inductor input negative, rather than positive. Inductor L1 filters the PWM voltage into a triangle wave of current. C2 removes the high-frequency ripple and attenuates the 120 Hz component in the LED load. Note that the buck regulator is not capable of supplying power to the load if the load voltage is greater than the input voltage. There are “flat spots” in the input current waveform around the input voltage zero crossings. Power factor remains excellent, even with this distortion. 2.2 Power factor controller operation Startup The circuit starts up with a trickle of current into C8 through R7. It takes about ¼ second to charge C8 to U1’s startup voltage. The trickle current adds to LED current, slightly improving circuit efficiency. The startup timer in U1 starts the switching cycle by turning on Q2. Current in Q2 and L1 increases from zero to about 1400 mA at the peaks of the input sinewave. This current appears on R22 which drops about 1 volt max. L1’s current continues to flow after Q2 turns off, instead flowing in D1. The current ramps toward zero, at which time D1 turns off. The FET drain voltage then begins to fall. Quasi-resonant FET turn-on L1 and stray capacitance then ring the voltage at D1’s anode down to about twice the LED voltage below the positive rail. When the ringing voltage turns up, U1 senses the end of L1’s discharge and turns on Q2 very close to the minimum ringing voltage, starting the next cycle. Current in L1’s upper winding therefore ramps between zero and twice the load current. When Q2 turns on, D1 has already turned off, so Q2 never sees D1’s reverse recovery current. Bootstrap power Housekeeping power is supplied by the auxiliary (lower) winding on L1. The winding is connected through D4 so that the transformed LED voltage (positive) is applied to C9, which powers U2, and C8 which powers U1. R2 and C9 form a filter to remove ringing spikes due to leakage inductance. The auxiliary (lower) winding on L1 has a turns ratio that puts about 15 V on C9 with the AC line applied. The voltage on C9 is proportional to the LED voltage. This will limit the number of series LEDs in the load to a relatively narrow range, set by the acceptable Vcc for U1 and U2. The auxiliary winding also provides U1 with timing for the zero-current sensing function, through R5. Doc ID 018890 Rev 1 5/28 Circuit description 2.3 AN3410 Controlling the LED average current The control circuit works by controlling average input power. For this explanation, it is assumed that the power converter efficiency is constant over the range of line voltage and LED voltage. Therefore, average output power is also controlled. Linear approximation of input power Over a narrow range of line voltage, the sum of scaled average line voltage and scaled average line current closely approximates a constant power curve. If the sum is held constant, the input power can be held approximately constant by a feedback circuit. Figure 4. Constant-power V-I curve and linear approximation Figure 5 below shows the approximation’s error over a typical line voltage range. Figure 5. Linear approximation error This will be explained in more detail later. Obtaining the line voltage reference In previous work it was found that the reference waveform for the inverted step-down switching regulator should be taken from the negative output terminal for best power factor. This point gives a line current waveform that goes to zero around the zero crossings and rises (and falls) more rapidly than the line voltage sine wave. (The converter input current goes to zero when line voltage falls below the DC output voltage.) The resulting line current can be seen in the scope photos. It’s ugly, but the power factor is excellent and THD is acceptable. Average line voltage (minus the LED voltage) is derived from the peak voltage at the bottom of the LED string by R6, R15, and R20. U1 contains a precision peak detector, which places 6/28 Doc ID 018890 Rev 1 AN3410 Circuit description the peak from this divider on C6. (Normally this voltage is used internally by the L6564 to adjust its multiplier gain to accommodate a wide line voltage range.) Obtaining the line current (the controlled variable) The vast majority of input current flows through R22, the sense resistor for the PFC-flyback converter. The average of the current in R22 and the average of the line voltage (minus the LED voltage) will be used in the power calculation. Calculating the average power Scaling and addition of voltage and current is done by R17 and R14. The AC noise present at their junction is removed by C12 - the DC voltage on C12 now represents the input power as calculated by the linear approximation. This voltage will be regulated by the slow PFC feedback loop. Op amp U2 is wired as a non-inverting amplifier. The feedback loop requires only one inversion, supplied by the op amp in U1. U2 performs three different functions: – Derives a reference voltage from U1. – Provides gain for the relatively low voltage on C12. – Provides a point in the circuit to compensate for different LED voltages. A DC reference voltage is derived from U1’s inverting input. This point will always be at 2.5 V if the control loop is in steady-state, because there is no DC current path to any other voltage source. The reference voltage is delivered to U2’s inverting input by divider R18R21. The voltage divider R18-R21 also sets the DC gain for U2. If this circuit acted alone, the input power would be (approximately) regulated to a fixed value, and the LED current would inversely track the LED voltage. 2.4 Setting the “DC” operating point The control loop is to set the average current through R22 to deliver slightly more than the desired LED current when both the line voltage and LED voltage are at design center. Deviations of line and LED voltage from this point will then cause smaller deviations of LED current. The input current required is ILED x VLED / (VLINE x efficiency). The straight-line approximation (of the constant-power curve) should give equal voltage from the average line voltage and the average input current. The value of R22 is determined from the usual calculations (see ST’s excellent application note AN1059, reference 1). The average input current (in R22) can now be calculated from the design center line voltage, output power, and efficiency. At design center line voltage, LED current, and LED voltage, the average voltage appearing across R17 due to current from R14 must match the average voltage on R22. Doc ID 018890 Rev 1 7/28 Circuit description AN3410 Stirring in the LED voltage Compensation for LED voltage changes follows two paths in this design: – by direct subtraction of the LED voltage from the line voltage (path 1) – from the transformed LED voltage on C9 through R12 (path 2). Path 1 An increase of LED voltage will reduce the voltage at U1 pin 5. This reduces current through R14. The feedback loop will then call for more line current to compensate. Path 2 The voltage compensation obtained from the 18-LED load is not quite sufficient to flatten the LED current over the expected range of LED voltage, so the second path through R12 is also implemented. The LED voltage (multiplied by L1’s turns ratio) is available on D4’s cathode, filtered by C9. Current proportional to this voltage is delivered to U2’s inverting input by R12. The reference for the operating point is thus compensated by the LED voltage. The figures below show the results of compensating the input current setting for load voltage. Figure 6. Input current compensated for 3 levels of output power In Figure 6 the reference has been compensated by adding a current proportional to the LED voltage. Three levels of input power are shown, corresponding to LED voltages of 10% below nominal, nominal, and 10% above nominal. Figure 7 below shows the error between the approximations and the ideal values. 8/28 Doc ID 018890 Rev 1 AN3410 Circuit description Figure 7. Error for 3 levels of load power Note that the curves coincide at the nominal line voltage, where LED voltage compensation is perfect. Also, for nominal LED voltage, line voltage compensation is at its best, +/-1.5%. Over the entire range of line voltage (120 V +10%,-20%) and LED (+/-10%) voltage, the variation of LED current is less than 4.5% (+1.5%, -3%). 2.5 Designing the “DC” control loop Finding values for the resistors in this loop is surprisingly easy. (For this procedure, diode drops will be ignored for simplicity.) Let us assume that the average LED current is to be 350 mA, and a string of 18 LEDs drops about 54.6 V. Output power is 18 watts. Input power will be assumed to be 20 W (90% efficiency, actually conservative). Assume the input current and voltage waveforms are sine waves (power factor is high, so this is a good assumption). The converter output current waveform is a triangle wave with an (assumed) rectified sinusoidal upper envelope, and a lower envelope at zero. The short-term average of the triangle wave is half its height. The rectified sinusoid has an average current of 63% of its peak current. At the peak of the input voltage, the peak current delivered to the LEDs is therefore. Equation 1 We will allow some margin for the relatively slow fall time of the drain voltage due to the zero current detection, so the peak current will be slightly higher. We will design the inductor for 1.4 A peak current in the main winding. The L6564 control chip has an upper voltage limit on its current sense input of 1.08 V for linear operation. So the maximum value of R22 is Equation 2 Doc ID 018890 Rev 1 9/28 Circuit description AN3410 Select 0.681 Ω. Average input current will be slightly lower than expected - the input voltage and current are assumed sinusoidal, and the average current will be 0.9 times the input rms current. So at the design center of 115 V, average input current is Equation 3 The bulk of this current flows in R22, so the average voltage on R22 is Equation 4 0.156A * 0.681 Ohms = 0.106V R17 serves two functions - it is part of the linear approximation calculation, and part of a low-pass filter that cleans up both the triangular FET drain current pulses and the 120 Hz envelope. A value of 20 kΩ has been selected somewhat arbitrarily. At our nominal operating point, the average voltage developed on R17 due to line voltage must be equal to the average voltage at its lower end from R22. The current required is Equation 5 The L6564 Vff terminal connects internally to a precision peak detector. Its normal function, which will not be used, is to reduce the loop gain at high line voltages. However, the peak detector function gives us a buffered peak line voltage (minus LED voltage) reference. If LED voltage is zero, the entire line voltage appears on divider R6-R15-R20. For nominal line, the peak voltage at U1's MULT input is Equation 6 =3.61 V To provide the 5.3 µA needed by the linear compensator, Equation 7 We will use the nearest standard value, 649 kΩ. We will now examine the LED voltage compensation, to determine the values of R12, R18, and R21. If there is no LED voltage, there will be no reflected voltage from L1's auxiliary winding. (We will assume the circuit runs anyway) Therefore, R12 contributes no current. Since there is no output voltage there will be no output power, and since efficiency is assumed constant, no input power. Therefore, there will be no voltage on R22, and the only voltage on C12 will be the divided line peak voltage from C6. Divider R17-R14 will deliver 0.106 V to the noninverting input of U2. If the loop is balanced, the op amp output will be at exactly 2.5 V (so that the COMP terminal on U1 does not move). The op amp inputs will be at exactly the same voltage, 0.106 V. 10/28 Doc ID 018890 Rev 1 AN3410 Circuit description To find the value of R21, we first arbitrarily select 25.2 kΩ for R18. R21’s value is then Equation 8 =1.115 kΩ Use 1.10 kΩ. Now we examine the operating points when the LEDs are at nominal voltage. The peak voltage at U1's Vff terminal will now be Equation 9 So at nominal input power (20 W) the average voltage applied to C12 and U2's non-inverting input is the voltage from R22 plus the divided voltage from R14 and R17: Equation 10 R12 must be selected to deliver this voltage at U2's inverting input. The equivalent voltage and resistance presented by U2's 2.50 V output and the R18-R21 divider is: Equation 11 R12 must bring the voltage on U2's inverting input up to 0.175 V from 1.06 V when the reflected LED voltage is at its nominal value. Neglecting diode and R2 drops, the voltage at the top of R12 is the LED voltage times the turns ratio of L1's auxiliary winding: Equation 12 R12 must provide enough current to bring the voltage up to 0.175V. The added voltage due to R12 must be 0.175 V - 0.104 V = 0.071 V. R12's value is then Equation 13 We will use 240 kΩ. Some trimming may be required to hit the 350 mA * 1.012 target to center the operating point in the error band. Doc ID 018890 Rev 1 11/28 Circuit description AN3410 In summary, the parts selected are: 12/28 – R22 0.681 Ω – R17 20 kΩ – R14 649 kΩ – R18 25.2 kΩ – R21 1.10 kΩ – R12 240 kΩ Doc ID 018890 Rev 1 AN3410 3 Control loop dynamics Control loop dynamics The control loop is intentionally very slow, much slower than a normal PFC loop. The reason is that the eye perceives abrupt changes of light intensity as flashes. The eye has an intensity control loop of its own, which adjusts the light falling on the retina by controlling the iris opening. The loop bandwidth is about ¼ Hz. If light level changes occur slower than this, they are not perceived as flashes. the iris can keep up with the change. The LED driver control loop is set up as a continuous -20 dB/decade / 90 degrees lag system, which gives a response to perturbations having no overshoot. The current control loop breakpoints are: – A pole at zero frequency – A pole at 0.35 Hz (R17-C12) – A zero at 0.35 Hz (R25-C11) A third pole at high frequency due to noise filter C5 (gain is well below unity at this corner the pole is inconsequential). The pole due to R17 and C12 is essential to the system – switching ripple and the 120 Hz envelope must be filtered. The R25-C11 zero cancels this pole, keeping the gain slope at 20 dB/decade. There are two problems that affect the control loop: – There is a second path in the system that gives positive feedback, the compensation for LED voltage. Fortunately, the voltage loop gain is low due to the low dynamic resistance of the LEDs, typically 1 Ohm per LED. Changes in the input power can only cause very small changes in the LED voltage. – The R18-R21 voltage divider, necessary for the DC control scheme, gives considerable unnecessary loop gain. Note that R23 provides a very handy single-point gain adjustment. Increasing its value reduces the gain of the entire loop. This is a good point for stability testing - it should be possible to reduce the value of R23 by a factor of 2 to 4 (6-12dB) before sustained (slow!) oscillation results. Control loop transient response can be observed at startup (see Figure 16). The LED current rises rapidly (a nice transient), and the control loop takes over very smoothly. The LED current should not overshoot, and it should settle to its final level within 2 or 3 seconds. Either excessive gain (overshoot) or unmatched time constants for the breakpoint networks R17-C12 and R25-C11 (lumps or dips in the startup waveforms) can cause strange behavior during startup. Excessive gain can result in ringing or sustained oscillation - both are quite annoying. Mismatched breakpoints can result in a dip or rise of output current about 1.5 seconds after the unit starts. The result is not too annoying, but it should be corrected - customers demand smooth operation. Doc ID 018890 Rev 1 13/28 Performance with LED loads 4 AN3410 Performance with LED loads The unit was designed to source 350 mA into a string of 18 1 W LEDs. The project goal was to obtain the highest possible efficiency. Table 1. Numeric data for 18-LED load Line voltage 90 V 96 V 102 V 108 V 114 V 120 V 126 V 132 V 138 V Input power, watts 20.5 20.77 20.82 20.61 20.51 20.46 20.31 19.95 19.48 Power factor 0.973 0.978 0.982 0.985 0.987 0.988 0.990 0.991 0.991 Output voltage 54.8 54.8 54.7 54.7 54.6 54.6 54.5 54.4 54.3 Output current, amps 0.348 0.353 0.355 0.352 0.351 0.35 0.348 0.342 0.334 Output power, watts 19.070 19.344 19.418 19.254 19.164 19.110 18.966 18.604 18.136 Efficiency% 93.03 93.14 93.27 93.42 93.44 93.40 93.38 93.26 93.10 Power loss, watts 1.429 1.425 1.401 1.355 1.345 1.350 1.344 1.345 1.343 14/28 Doc ID 018890 Rev 1 AN3410 5 Graphical data Graphical data Figure 8. LED current vs. line voltage for 18 Figure 9. Power factor vs. line voltage Figure 10. Efficiency vs. line voltage Doc ID 018890 Rev 1 15/28 Graphical data AN3410 Figure 11. Power loss vs. line voltage 16/28 Doc ID 018890 Rev 1 AN3410 6 Waveforms Waveforms Trace colors: – Yellow = line voltage, 50 V/div ref 0 – Magenta = line current, 100 mA/div ref 0 – Blue = LED voltage, 10 V/div ref -3 div – Green = LED current, 100 mA/div ref -3 div Figure 12. Waveforms at 96 V line Figure 13. Waveforms at 108 V line Figure 14. Waveforms at 120 V line Figure 15. Waveforms at 132 V line Note that the flat spot near the zero crossing becomes longer as the load voltage becomes a larger percentage of the line voltage (low line is worst). This will place an upper limit on the LED output voltage as waveform distortion reduces power factor. The design can be pushed to higher numbers of LEDs, up to the point where power factor or THD become limiting factors. Doc ID 018890 Rev 1 17/28 Startup 7 AN3410 Startup Startup scope photo - cold start, everything discharged, 7 LEDs, 120 V input. Trace colors: – Yellow AC line, 100 V/div, ref +2 div – Blue LED voltage, 10 V/div, ref -3 div – Green LED current, 100 mA/div, ref -3 div Figure 16. Startup waveforms This startup time is too long for some applications. Some circuitry should be added to increase the current limit during startup. An R-C series network could be added from U1 pin 1 to ground (pin 8). The network would have no effect once the feedback loop takes over – the voltage on pin 1 is stable at 2.5 V. The time can also be reduced by reducing the value of R7, but this will reduce efficiency. 18/28 Doc ID 018890 Rev 1 AN3410 8 Component stress Component stress This plot was taken near the peak of line voltage, where stresses are greatest. Trace colors: – Yellow Q2 drain voltage, 100 V/div, ref -3 div – Blue Q2 drain current, 0.735 A/div, ref -3 div – Magenta D1 diode voltage, 100 mA/div, ref -3 div Figure 17. Component electrical stress Doc ID 018890 Rev 1 19/28 Thermal stress 9 AN3410 Thermal stress It is not likely that the layout will be used as is by the customer. The temperatures below are for guidance only – measurements were taken after 30 minutes operation, in open air, on the workbench, in a 25.3 °C ambient, 120 V input, 18 LED load. The board axis was horizontal, board position vertical. L3 was above L2. Table 2. 20/28 Thermal stress Component Temperature L2 40.3 BR1 62.4 Q2 52.5 L1 51.4 L3 46.1 U1 42.3 D1 59.3 C2 32.2 Doc ID 018890 Rev 1 AN3410 10 Conducted EMI Conducted EMI The following plots each show the maximum of 3 successive sweeps (peak hold). Figure 18. Conducted EMI, line 1 Figure 19. Conducted EMI, line 2 This unit should pass average and quasi-peak testing as is. If difficulties are encountered, the values of C7 and C1 can be increased at the expense of reduced power factor. Doc ID 018890 Rev 1 21/28 PC layout 11 AN3410 PC layout Figure 20. Top side foil !-V Figure 21. Top side placement !-V Figure 22. Bottom side layout !-V 22/28 Doc ID 018890 Rev 1 AN3410 PC layout Figure 23. Bottom side placement !-V Doc ID 018890 Rev 1 23/28 Bill of materials AN3410 12 Bill of materials Table 3. BOM Designator Comment Description Footprint Manufacturer Vendor BR1 Bridge 0.8 A 1A DIP bridge RH0x Diodes Inc HD06 Digi-Key HD06DICT-ND C1, C7 220 nF 250 V Capacitor BOXSDLWH7.5 -0.6-10.3-6-10.8 Panasonic ECQE2224KB Digi-Key P10971-ND C2 470 µF 63 V Capacitor CEH16x20 mm Panasonic EEUFC1J471 Digi-Key P10352-ND C5 0.1 µF Capacitor 0805 0805 X7R C6 1.0 µF Capacitor 0805 Murata GRM21BR71C105M A01L Digi-Key 490-1692-1ND C8, C9, C12 22 µF 25 V Capacitor CEV5MMP Panasonic EEAFC1E220 Digi-Key FC series P11213-ND C11 2.2 µF Capacitor 0805 0805 Z5V C13 2.2 nF Capacitor 0805 0805 X7R D1 STTH1R04A Diode SMA ST STTH1R04A D4, D8, D10 MMSD4148 Diode SOD-123 Fairchild MMSD4148 Digi-Key MMSD4148ND F1 1A Fuse RES0.5 Littelfuse 0251001.MXL Digi-Key F2313-ND L1 Cramer E35882 inductor Shulin EF-20 bobbin Cramer E35882 Inductor L2, L3 1 mHy 0.5 A Inductor INDUC9MMVE RT Wurth 744 772 102 Q2 STD5NM50 FET N-CHAN D-PAK ST STD5NM50 R1, R7 47 kΩ Resistor 1206 1206 5% R2 47 Ω Resistor 0805 0805 5% R3, R6, R15 220 kΩ Resistor 1206 1206 5% R5 47 kΩ Resistor 0805 0805 5% R8, R9 10 kΩ Resistor 1206 0805 5% R11 150 kΩ 1% Resistor 0805 0805 1% R12 453 kΩ 1% Resistor 0805 0805 1% R13 0 Ω jumper Resistor 0805 0805 000 R14 681 kΩ 1% Resistor 0805 0805 1% R16 200 Ω Resistor 0805 0805 5% R17 20.0 kΩ 1% Resistor 0805 0805 1% R18 25.2 kΩ 1% Resistor 0805 0805 1% 24/28 Doc ID 018890 Rev 1 AN3410 Table 3. Bill of materials BOM (continued) Designator Comment Description Footprint Manufacturer R19 10 kΩ 1% Resistor 0805 0805 1% R20 10 kΩ Resistor 0805 0805 1% R21 1.40 kΩ 1% Resistor 0805 0805 1% R22 0.681 Ω 1% Resistor 0805 0805 1% R23 100 kΩ Resistor 0805 0805 5% R25 180 kΩ Resistor 0805 0805 5% U1 L6564 Transition mode PFC controller SSOP10 ST L6564 U2 TS321AILT Low power op amp in SOT23-5L SOT-23-5L ST TS321AILT Doc ID 018890 Rev 1 Vendor Digi-Key 497-8093-1ND 25/28 References 13 26/28 AN3410 References 1. “Design equations of high-power-factor flyback converters based on the L6561” (AN1059) 2. L6561 PFC controller datasheet 3. L6562 PFC controller datasheet 4. L6562A PFC controller datasheet 5. L6564 PFC controller datasheet 6. STD5NM50 datasheet 7. STTH1R04A datasheet 8. “Low-cost LED driver for an A19 lamp” (AN3256) Doc ID 018890 Rev 1 AN3410 14 Revision history Revision history Table 4. Document revision history Date Revision 05-Sep-2011 1 Changes Initial release. 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