cd00004062

AN1088
APPLICATION NOTE
®
L6234 THREE PHASE MOTOR DRIVER
by Domenico Arrigo
INTRODUCTION
The L6234 is a DMOSs triple half-bridge driver with input supply voltage up 52V and output current of
5A. It can be used in a very wide range of applications.
It has been realized in Multipower BCD60II technology which allows the combination of isolated DMOS
transistors with CMOS and Bipolar circuits on the same chip. It is available in Power DIP 20 (16+2+2)
and in Power SO 20 packages.
All the inputs are TTL/CMOS compatible and each half bridge can be driven by its own dedicated input
and enable.
The DMOS structure has an intrinsic free wheeling body diode so the use of external diodes, which are
necessary in the bipolar configuration, can be avoided. The DMOS structure allows a very low quiescent
current of 6.5 mA typ. at Vs=42V , irrespective of the load.
DEVICE DESCRIPTION
The device is composed of three channels. Each channel is composed of a half bridge with two power
DMOS switches ( typ. Rdson of 300mW @ 25°C) and intrinsic free wheeling diodes. Each channel includes two TTL/CMOS and uP compatible comparators, and a logic block to interface the inputs with the
drivers. The device includes an internal bandgap reference of 1.22V, a 10V voltage reference to supply
the internal circuitry of the device, a central charge pump to drive the upper DMOS switch, thermal shutdown protection and an internal hysteretic function which turns off the device when the junction temperature exceeds approximately 160 °C. Hysteresys is about 20 °C.
Figure 1. L6234 Block Diagram
C4 220nF
C3
10nF
VCP
C5
1µF
VREF
D2
1N4148
D1
1N4148
VBOOT
Vs
VREF
10V
CHARGE
PUMP
Vs
IN1
T1
Vs
C2
100nF
C1
100µF
OUT1
EN1
T2
IN2
BRUSHLESS
MOTOR
WINDINGS
T3
OUT2
EN2
T4
SENSE1
THERMAL
PROTECTION
IN3
T5
OUT3
EN3
T6
SENSE2
RSENSE
GND
D98IN940A
April 2001
1/14
AN1088 APPLICATION NOTE
PIN DESCRIPTION.
Figure 2.
Vs ( INPUT SUPPLY VOLTAGE PINS).
VS
These are the two input supply voltage pins. The unregulated
input DC voltage can range from 7V to 52V.
T1
T3
ON/OFF
ON
VF
With inductive loads the recommended operating maximum
supply voltage is 42V to prevent overvoltage applied to the
L
B
C
OFF
-VF
(VS+VF)
DMosfets. In fact considering a full bridge configuration (see
fig. 2), when the bridge is switched off (ENABLE CHOPPING)
the current recirculation produces a negative voltage to the
ON/OFF
T2
T4
source of the lower DMOS switches (point A). In this condiA
S
tion the drain-source voltage of T1 and T4 is VS + VF + Vsense .
-VSENSE
Dinamically VF can be same Volts depending on the current
Rsense
slope, dI/dt, and also Vsense, depending on the parasitic inD98IN938A
ductance and current slope can be some Volts. So the drainsource voltage of T1 and T4 DMOS switches can reach more
than 10V over the VS voltage. The input capacitors C1 and C2 are chosen in order to reduce overvoltage
due to current decay and to parasitic inductance. For this reason C2 has to be placed as closed as possible to VS and GND pins.
The device can sustain a 4A DC input current for each of the two Vs pins, in accordance with the
power dissipation.
Figure 3. Reference Voltage vs.
Junction Temperature.
OUT1, OUT2, OUT3 (OUTPUTS). These are the output pins
that correspond to the mid point of each half bridge. They are Vref [V]
designed to sustain a DC current of 4A.
11
Vs = 52V
10
Vs = 24V
9
SENSE1, SENSE2.
Vs = 10V
SENSE1 is the common source of the lower DMOS of the half 8
bridge 1 and 2.
7
SENSE2 is the source of the lower DMOS of the half bridge 6
Vs = 7V
3.
5
Each of these pins can handle a current of 5A.
4
A resistance, Rsense, connected to these pins provides feed- 3
back for motor current control.
2
Care must be taken with the negative voltage applied to these 1
pins : negative DC voltage lower than -1V could damage the
device. For duration lower than 300ns the device can sustain 0-50 -25 0 25 50 75 100 125
pulsed negative voltage up to -4V.
Tj [°C]
For example, if enable chopping current control method is
used, negative voltage pulses appear to these pins, due to the Figure 4. Reference Voltage vs.
Supply Voltage.
current recirculation through the sensing resistor.
Vref [V]
12
Vref ( Voltage Reference).
This is the internal 10V voltage reference pin to bias the logic
and the low voltage circuitry of the device. A 1µF electrolytic capacitor connected from this pin to GND ensures the stability of the
DMOS drive circuit. This pin can be externally loaded up to 5mA .
Figure 3 and 4 show the typical behavior of the Vref pin.
10
8
6
4
Tj = 25°C
Vcp ( CHARGE PUMP ).
This is the internal oscillator output pin for the charge pump.
The oscillator supplied by the
10V Voltage Reference
switches from GND to 10V with a typical frequency of
2/14
2
0
0
10
20
30
Vs [V]
40
50
150
AN1088 APPLICATION NOTE
1.2MHz (see fig 4). When the oscillator output is at ground , C3 is charged by Vs through D1. When it
rises to 10V, D1 is reverse biased and the charge flows from C3 to C4 through D2, so the Vboot pin after a few cycles reaches the maximum voltage of Vs + 10V - VD1- VD2.
Vboot ( BOOTSTRAP).
This is the input bootstrap pin which gives the overvoltage necessary to drive all the upper DMOS of the
three half bridges (see fig 5).
Figure 5. Charge Pump Circuit.
Vs
C2
0.1µF
Vs+Vref-VD1-VD2
Vs+Vref-VD1
C1
100µF
D1
1N4148
Vs-VD1
f=1.2 MHz
VCP
C3
10nF
D2
1N4148
C4
0.22µF
VBOOT
Vs
Vref
f=1.2 MHz
CHARGE
HIGH
SIDE
DRIVER
OUT
SENSE
Vref
10V
PUMP
LOGIC INPUTS PINS.
EN1, EN2, EN3 (ENABLES).
These pins are TTL/CMOS and µP compatible. Each half bridge can be enabled by its own dedicated
pin with a logic HIGH. The logic LOW on these pins switches off the related half bridge (see Fig. 6). The
maximum switching frequency is 50kHz.
Figure 6. Control logic for each half bridge.
Figure 7. Cross Conduction Protection.
high level
INPUT
high level
high level
INPUT
PIN
low level
low level
time
high level
ENABLE
low level
low level
high level
time
low level
low level
DMOS ON
time
UPPER
DMOS
DMOS OFF
DMOS OFF
DMOS ON
UPPER
DMOS
DMOS OFF
DMOS OFF
DMOS OFF
300ns
LOWER
DMOS
LOWER
DMOS
time
tdelay
300ns
DMOS ON
time
DMOS ON
tdelay
DMOS ON
DMOS OFF
DMOS OFF DMOS OFF DMOS OFF
time
time
IN1, IN2, IN3 (INPUTS).
These pins are TTL/CMOS and µP compatible. They allow switching on the upper DMOS ( INPUT at
high logic level) or the lower Dmos (INPUT at low logic level) in each half bridge (see Fig. 6).
3/14
AN1088 APPLICATION NOTE
Cross conduction protection (see Fig. 7) avoids simultaneously turning on both the upper and lower
DMOS of each half bridge. There is a fixed delay time of 300ns between the turn on and the turn off of
the two DMOS switches in each half bridge. The switching operating frequency is up 50kHz. High commutation frequency permits the reduction of ripple of the output current but increases the device’s power
dissipation, however low commutation frequency causes high ripple of the output current. The switching
frequency should be higher than 16kHz to avoid acoustic noises.
The sink current at the INPUTS and ENABLES pins is approximately 30µA if the voltage to these pins is
at least 1V less than the Vref voltage (see Fig. 3 and Fig. 4). To avoid overload of the logic INPUTS and
ENABLES , voltage should be applied to Vs prior to the logic signal inputs.
POWER DISSIPATION
An evaluation of the power dissipation of the IC driving a three phase motor in a chopping current control application follows.
With a simplified approach it can be distinguished three periods (see Fig. 8) :
Figure 8.
Rise Time, Tr, period.
This is the rise time period, Tr, in which the current switches from one winding to another. In this
time a DMOS is switched on and the current increases up to the peak value Ipk with the law i(t)
= (Ipk/Tr) t. The energy lost for the rise time in
the period T is :
Tchop
Ipk
Iload
Ival
Tr
Erise = ∫ Rdson ⋅ i2(t)dt = Rdson ⋅ I2pk ⋅
0
Trise
Tload
Tr
3
Fall Time,Tf, period.
When the current switches from one winding to
another, there is a fall time in which the current
that flows in the intrisic diode of the DMOS decreases from Ipk to zero. If VD is the voltage fall
of the diode, the energy lost is :
Tfall
tf
Efall = ∫ VD(t) ⋅ i(t)dt
0
Tload
During this time the current that flows in the winding is limited by the chopping current control. The energy dissipated due to the ON resistance of the DMOS is :
Eload = Rdson ⋅ (Irms)2 ⋅ Tload
In the formula, Irms is the RMS load current, given by :
Irms =
 Ipk − Ival 2

√
(Iload)2 + 

3
√


and Iload is the average load current.
When the switch is ON, the energy dissipated due to the commutation of the chopping current control in
the DMOS can be assumed to be:
tcom
Eon = Vs ⋅ Ival ⋅
2
where tcom is the commutation time of the DMOS switch.
4/14
AN1088 APPLICATION NOTE
When the switch is OFF :
Eoff = Vs ⋅ Ipk ⋅
tcom
2
The energy lost by commutation in a chopping period, given by Eon + Eoff, is :
Ecom = Vs ⋅ Iload ⋅ tcom
The energy lost by commutation during the Tload time is given by :
Ecom = Vs ⋅ Iload ⋅ tcom ⋅ Tload ⋅ fchop
Quiescent Power Dissipation, Pq.
The power dissipation due to the quiescent current is Pq = Vs ⋅ Iq , in which Iq is the quiescent current
at the chopping frequency, fchop = 1/Tchop.
Total Power Dissipation.
Let’s evaluate the power dissipation of the device driving a three phase brushless motor in chopping current control. In the driving sequence only one upper DMOS and a lower one are on at the same time
(see fig. 9 and 10). The total power dissipation is given by :
Ptot =
2 ⋅ (Erise + Efall + Eload + Ecom)
+ Pq
T
Figure 11 shows the total power dissipation, Pd, of the L6234 driving a three phase brushless motor in
input chopping current control at different chopping frequency.
EVALUATION BOARD.
The L6234 Power SO20 board has been realized to evaluate the device driving, in closed loop control, a
three phase brushless motor with open collector Hall effect sensors.
Figure 9. Input chopping current circulation.
_
PHASE 12
CHOPPING INPUT
I1A
half bridge 1
Vs
half bridge 2
ILOAD
I1A
ON/OFF
OFF
OUT1
OUT2
ILOAD
OFF/ON
I1B
ON
I2B
I1B
IOFF
I2B
5/14
AN1088 APPLICATION NOTE
Figure 10. Three Phase Brushless motor control sequence.
IOUT1
BRUSHLESS MOTOR
OUT1
OUT2
L6234
OUT3
IOUT2
T
IOUT3
The device soldered on the copper heat dissipating Figure 11. L6234 Power Dissipation in Three
area on the board ,without any additional heat sink,
Phase Brushless Motor Control.
can sustain a DC load current of 2.3 A at Tamb of
Pd [W]
approximately 40 °C.
INPUT CHOPPING
Vs=36V
fchop=30kHz
The board provides a closed loop speed and torque
15
L=2mH
fchop=50kHz
control, with a constant TOFF chopping current conT=2ms
Tj=100C
trol method. It allows the user to change the direction and brake the motor.
DC
10
Constant tOFF Chopping Current Control.
When the current through the motor exceeds the
threshold, fixed by the ratio between the control
5
voltage Vcontrol and the sensing resistor, Rsense,
an error signal is generated, the output of the
LM393 comparator switches to ground. This state is
maintained by the monostable (M74HC123) for a
0
constant delay time ( tOFF ) generating a PWM sig2
0
1
3
4
5
nal that achieves the chopping current control. The
ILOAD [A]
PWM signal is used for chopping the INPUT pattern. During the toff in chopping current control, the
current flows in the low side loop ( see fig. 9 ) and does not flow through the sensing resistor.
The tOFF value can be set by the R9 and C11 to values shown in the table 1.
A suitable value of toff for the majority of applications is 30µs. The larger the tOFF, the higher is the current ripple. If the tOFF is too large the ripple current becomes excessive . On the other hand if the tOFF is
too small the winding current cannot decrease under the threshold and current regulation is not guaranteed.
6/14
AN1088 APPLICATION NOTE
Figure 12. Application board Schematic Circuit.
Vs=8V to 42
+5V
OUT
3
T1
1
L7805
IN
J7
C1
100uF
60V
2
C7
10uF
GND
C6
220nF
Z1
18V
10k
R5
1N4148
1N4148
D1
D2
C3
10nF
C2
100nF
Vs
HALL
EFFECT
SIGNALS
IN1
IN2
IN2
IN3
IN3
14
EN1
8
EN2
3
EN3
13
CONTROL EN1
LOGIC
EN2
BRAKE
DIR
9 12
IN1
EN3
PWM
Vcp
17
18
CONSTANT toff
CHOPPING
CURRENT
CONTROL
R11
10k
B
PWM
C10
100nF
16
_
Q
2
3
1 10 11 20
16
2 19
C5
1uF
TORQUE &
SPEED
CONTROL
A
M74HC123
4 monostable
15 14
R2
R3
R4
1Ω
1Ω
1Ω
REFERENCE
SPEED
Reference
Speed
Table 1. toff selection
2
+
4
3
R6 1K
Vsense
C8
470pF
R7 11 k
8
LM393
R9
R1
1Ω
Hall effect signal
1
BRUSHLESS MOTOR
SENSE
Vref
8
1
OUT3
15
R10 10K each
Vcontrol
C9
100nF
5
POWER SO20
+5V
+5V
OUT2
L6234
Figure 13. Constant toff current control.
+5V
OUT1
6
+5V
PWM
HALL
EFFECT
SENSORS
Vboot
7
4
GND
Vsense
C4
220nF
toff
20µs
30µs
R9
100k
100k
C11
270pF
330pF
45µs
70µs
100k
100k
560pF
1nF
+5V
J1
+5V
100k
Vcontrol
R8
1K
C11
330pF
Torque & Speed Closed Loop Control.
The motor’s rotational speed is determined by the frequency of the Hall effect signals. The speed control
loop has been achieved by comparing this frequency with a frequency of a reference oscillator (see fig.
14) that corresponds to a desired speed limit.
Figure 14. PLL Motor Control.
REFERENCE
FEEDBACK
PHASE/
FREQUENCY
DETECTOR
Amp.
Vcontrol
PWM
MOTOR
COMPENSATION
NETWORK
HALL
SENSORS
D01IN1209
7/14
AN1088 APPLICATION NOTE
Figure 15. Oscillator for Reference Speed.
When the hall effect signal frequency is lower than the reference
frequency, the control voltage is
maintained to a value that sets the
motor current limit and therefore the
torque control limit. The peak current limit is given by Ipeak = Vcontrol/Rsense.
When the frequency from the Hall
Effect sensors exceeds the reference frequency and an error signal
is generated by the PLL (see Fig.
14). An LM358 comparator, a loop
amplifier and an auxiliary OP-AMP
ensure the right gain and filtering to
guarantee the stability (see fig.16).
The error signal causes Vcontrol
decrease to a value that sets the
PWM chopping current control in order to reduce the torque and set
the desired speed. The motor
speed is regulated to within ± 0.02
% of the desired speed.
Reference Speed
+5V
4
8
3
NE555
1
R26
36K
R27
7
C21
100nF
16K
6
5 2
C19
100nF
C20
100nF
Figure 16. Phase Locked Loop and filtering.
+5V
+5V
R17
10M
LM358
C12
100nF
8
R14
47K
3
Vcontrol
+
-
1
2
BAT47
R13
47K
+5V
4
R12
47K
33K
+5V
R20
Output
11
Aux.
OP-AMP
2.5V
8
9
Loop
Amplifier
+VIN 13
C14
100nF
C13
1uF
R21
91K
C17
47nF
270K
R15
47K
P2
1K
R19
91K
R16
P1
5K
+5V
GND
C16
220nF
C15
100nF
R18
14
HALL1 (Speed feedback)
6
12 10
5
3635
Phase/ Frequency
Detector
3
2
7
15
1
Reference Speed
TP8
Figure 17. Control Logic Circuit.
+5V
+5V
R22 R23
10k 10k
MOTOR
HALL
EFFECT
SIGNALS
R29
10k
SW2
R24
10k
HALL1
1
7
19
2
18
HALL2
HALL3
DIR
BRAKE
PWM
3
17
4
GAL 16V8 16
5
15
6
14
10
20
EN1
EN2
EN3
IN1
IN2
IN3
EN1
EN2
EN3
IN1
IN2
IN3
PWM
+5V
R25
10k
R26
10k
C18
100nF
DIRECTION CHANGE
DIR =0 GND : BACK ROTATION
DIR = 5V : FORWARD ROTATION
BRAKE
GND
SW1
DIR
J1
BRAKE FUNCTION
BRAKE = GND : BRAKE
BRAKE = 5V : GO
8/14
4
16
GND
Control Logic Circuit.
The logic sequence to the motor is
generated by a GAL16V8, which
decodes the Hall Effect signals and
generates the INPUT and ENABLE
pattern shown in Fig. 18.
The brake function is obtained by
setting the input pattern to logic low
and thus turning on the lower
DMOS switches of the enabled halfbridges.
The PWM signal is used for chopping the INPUT pattern.
The control logic circuit decodes
Hall effect sensors having different
phasing.
With the DIR jumper opened the
application achieves forward rotation for motors having 60° and 120°
Hall Effect sensor electrical phasing
and the reverse rotation for motors
having 300° and 240° Hall Effect
sensor phasing.
Connecting the DIR jumper to
ground sets the reverse rotation for
motors having 60° and 120° Hall
sensors phasing and the forward
rotation for motors having 300° and
240° Hall sensor phasing.
The SW2 switch performs the startstop function.
AN1088 APPLICATION NOTE
Figure 17.
0˚
ELECTRICAL DEGREES
360˚
HALL1
SENSOR
SIGNALS
HALL2
HALL3
EN1
ENABLE
EN2
EN3
IN1
FORWARD
ROTATION
IN2
IN3
IN1
REVERSE
ROTATION
IN2
IN3
IOUT1
0
MOTOR
DRIVE
CURRENT
IN
FORWARD
ROTATION
IOUT2
0
IOUT3
0
NO PWM
PWM CONSTANT tOFF
D98IN912
9/14
AN1088 APPLICATION NOTE
Layout Considerations.
Special attention must be taken to avoid overvoltages at Vs and additional negative voltages to the
SENSE pins and noise due to distributed inductance. Thus the input capacitor must be connected close
to the Vs pins with symmetrical paths. The paths between the SENSE pins and the input capacitor
ground have to be minimized and symmetrical . The sensing resistors must be non-inductive. The device GND has to be connected with a separate path to the input capacitor ground.
Figure 19. Application Board Layout.
Figure 20. Component side.
10/14
AN1088 APPLICATION NOTE
Figure 21. Copper side.
APPLICATION IDEAS.
The L6234 can be used in many different applications. Typical examples are a half bridge driver using
one channel and a full bridge driver using two channels. In addition, the bridges can be paralleled to reduce the RDSon and the device dissipation.
The paralleled configuration can also be used to increase output current capability. Channel 1 can be
paralleled with Channel 3 or Channel 2 can be paralleled with Channel 3. Channel 1 should not be paralleled with Channel 2 because the sources of their low side DMOSs are connected to the same SENSE1
pin .
Application ideas for the L6234 follow.
Figure 22. Constant frequency current control
Vs
1N4148
1N4148
100uF
220nF
100nF
CONTROL
LOGIC
EN1
EN1
EN2
EN2
EN3
EN3
IN1
IN1
10nF
Vs
Vcp
Vboot
OUT1
OUT2
L6234
POWER SO20
IN2
OUT3
IN3
IN2
GND
Vref
SENSE1
IN3
SENSE2
Reset
1uF
S
Vsense
Q
R
RSENSE
S
+5V Rx
Cx
OSC
Q
+5V
R
Vcontrol
+5V
100nF
L6506
Constant frequency
Current Control
1
Fchop= __________
0.69 Rx Cx
for Rx>10kOhm
11/14
AN1088 APPLICATION NOTE
Low Cost Application with Speed and Torque Control Loops.
Figure 23. Complete three phase brushless motor application with speed and torque control.
VS
IN1
HALL
EFFECT
SIGNALS
7
IN2
17
18
L6234
8
EN2
+5V
16
2
19
GND
1
7
1/2
M74HC123
MONOSTABLE
Q
12
11
6
+
RSENSE
0.3W
VSENSE
5
V5=+5V
1/4
TSM221
6
7
BRUSHLESS MOTOR
SENSE
V5=+5V
A
OUT3
15
13
1,10,11,20
11
OUT2
5
POWER SO20
3
EN3
10
OUT1
6
14
EN1
PWM
12
4
IN3
CONTROL
LOGIC
9
HALL
EFFECT
SENSORS
1/fe
C1 200nF
V5
3
Vm
+5V
Rx2
R2 1M
1
Cx2
VCONTROL
R3 4K
R4
1K
1/4
TSM221
4
+
2
R1 100K
3
Ton
1Q
Vref
(Reference
Speed Voltage)
16
2
1/2
M74HC123
MONOSTABLE
13
15
1
14
1B
HALL EFFECT SIGNAL
8
+5V
Rx1
100K
Cx1
D01IN1210
SPEED LOOP
A low cost solution to obtain a complete three phase brushless motor control application with speed and
torque closed control loop is shown in Fig. 23. This simple low cost solution is useful when high dynamic
performances and accuracy of the speed loop are not required.
The current regulation limit, which determines the torque , is given by Vcontrol/Rsense. The constant
toff of the PWM is fixed by Rx2 and Cx2.
The speed loop is realised using a Hall effect signal, whose frequency is proportional to the motor
speed. At each positive transition of the Hall effect sensors the monostable maintains the pulse for a
constant time , Ton, with a fixed amplitude, V5. The average value of this signal is proportional to the frequency of the Hall effect signal and the motor speed . An OP-AMP configured as an integrator , filters
this signal and compares it with a reference voltage, Vref, which sets the speed . The generated error
signal is the control voltage, Vcontrol, of the currrent loop. Therefore the current loop modifies the produced torque in order to regulate the speed at the desired value.
The values of Cf and R2 should be chosen to obtain a nearly ripple free op-amp output, even at low motor speed. This constrain limits the system bandwidth and so limits the response time of the loop.
The regulated speed, for a rotor with n pairs of permanent magnetic poles , is given by :
R1 

1 + R2 


ωm =
⋅ Vref ⋅ 60
1
V5 ⋅ Ton ⋅ n +
KG
with
KG =
[RPM]
R4
1
1 R2
⋅ Kt ⋅
⋅ ⋅
Rsense B R1
R3 + R4
in which Kt, expressed in [Nm/A] , is the Motor Torque constant and B, expressed in [Nms], is the Total
Viscous Friction.
In most cases 1/KG can be neglected.
12/14
AN1088 APPLICATION NOTE
The Ton values, given by KCx1Rx1, must be less than the period of the Hall effect electrical signal at
the desired motor speed , so Ton must meet the requirement of 1.1 :
( 1.1 ) Ton <
60
n ⋅ ωm
For the motor and the load used in this application, which have the following parameters :
Jt = 10-4 [Kg ⋅ m2] (Motor plus Load Inertia Moment); Kt = 10-2 [Nm/A] ;
n=4 ;
R1=100k +/- 10% [kΩ]
;
R2=1M ±1[kW]
;
B = 10-5 [Nms]
Cf=220n [F]
A regulated speed of 6000RPM can be obtained with an accuracy of around +/-3%, considering Ton accuracy of +/-1% , the V5 and Vref mismatch of +/-1% .
If the speed is 6000RPM, there are 100 rotor revolution for second, with n=4, the Hall effect frequency
is 400Hz. Therefore Ton has to be lower than 2.5ms (according to equation 1.1).
The phase margin is about 45° and the response time of the speed loop for a speed step variation is
around 200ms .
6X6 BRUSHLESS APPLICATION
Figure 24. 6x6 Three Phase Brushlees Application Circuit
Vs
100uF
100nF
SPEED AND POSITION
FEEDBACK
IN1
IN2
IN3
EN1
EN2
EN3
EN2
Vcp
Vref
THREE PHASE
BRUSHLESS
MOTOR
Vboot
OUT1
OUT1A
OUT2
OUT1B
L6234
GND
CONTROL
LOGIC
220nF
10nF
Vs
IN1A
IN1B
IN2A
EN1
1N4148
1N4148
OUT3
OUT2A
SENSE1
SENSE2
OUT2B
1uF
Vs
100nF
100uF
1N4148
1N4148
220nF
10nF
Vs
IN3A
IN1
IN2
IN3
EN1
EN2
EN3
IN3B
IN2B
EN3
Constant Toff PWM
Current Control. Two M74HC124 plus an LM339
Vcp
Vboot
OUT1
L6234
OUT3
GND
Vref
OUT3A
OUT2
OUT3B
SENSE1
SENSE2
Comparator & monostable
+5V 100nF
+5V
8
B
100nF
PWM3
16 2 3 1
A
M74HC123
monostable 15
_ 4
14 8
Q
1
1uF
+5V
LM339
+
1K
Vsense3
470pF
4
Vcontrol
+5V
Comparator & monostable
Vsense2
PWM2
Comparator & monostable
Vsense1
PWM1
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AN1088 APPLICATION NOTE
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences
of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is
granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specification mentioned in this publication are
subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products
are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics.
The ST logo is a registered trademark of STMicroelectronics
© 2001 STMicroelectronics – Printed in Italy – All Rights Reserved
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