cd00004396

AN1625
APPLICATION NOTE
L6235 THREE PHASE BRUSHLESS DC MOTOR DRIVER
by Vincenzo Marano
Modern motion control applications need more flexibility that can be addressed only with specialized
ICs products. The L6235 is a fully integrated motor driver IC specifically developed to drive a wide range
of BLDC motors with Hall effect sensors. This IC is a one-chip cost effective solution that includes
several unique circuit design features. These features, including a universal decoding logic that allows
the device to be used with most common Hall effect spacing, will be described. The principal aim of this
development project was to produce an easy to use, fully protected power IC. In addition several key
functions as protection circuit and high speed PWM current control allow to drastically reduce the
external components count to meet requirements for many different applications.
1
INTRODUCTION
For small-motor applications many appliance designers favor modern three phase brushless DC motors because of the high efficiency (as great as 95%) and small size for a given delivered power. Designers have to
handle control logic, torque and speed control, power-delivery issues and ensure safe operation in every load
condition. The L6235 is a highly integrated, mixed-signal power IC that allows to easily design a complete motor
control system for BLDC motor. Figure 1 shows the L6235 block diagram. The IC integrates six Power DMOS,
a centralized logic circuit to decode hall effect sensors and a constant tOFF PWM current control technique (Synchronous mode) plus other added features for safe operation and flexibility.
Figure 1. L6235 Block Diagram.
VBOOT
VCP
VBOOT
VBOOT
CHARGE
PUMP
VSA
THERMAL
PROTECTION
OCD1
DIAG
OCD
OUT1
10V
OCD1
OCD2
OCD
OCD3
VBOOT
EN
BRAKE
FWD/REV
OCD2
H3
HALL-EFFECT
SENSORS
DECODING
LOGIC
H2
GATE
LOGIC
SENSEA
VBOOT
H1
RCPULSE
OUT2
10V
TACHO
MONOSTABLE
VSB
OCD3
OUT3
10V
TACHO
10V
5V
SENSEB
PWM
VOLTAGE
REGULATOR
ONE SHOT
MONOSTABLE
MASKING
TIME
+
SENSE
COMPARATOR
VREF
RCOFF
D99IN1095B
October 2003
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AN1625 APPLICATION NOTE
Table of Contents
1
2
3
4
5
2/39
INTRODUCTION ................................................................................................................................1
DESIGNING AN APPLICATION WITH L6235 ...................................................................................3
2.1 Current Ratings ........................................................................................................................3
2.2 Voltage Ratings and Operating Range ....................................................................................3
2.3 Choosing the Bulk Capacitor....................................................................................................5
2.4 Layout Considerations .............................................................................................................5
2.5 Sensing Resistor ......................................................................................................................7
2.6 Charge pump external components .........................................................................................8
2.7 Sharing the Charge Pump Circuitry .........................................................................................9
2.8 Reference Voltage for PWM Current Control.........................................................................10
2.9 Input Logic pins ......................................................................................................................11
2.10 DIAG pin.................................................................................................................................11
2.11 Programmable off-time Monostable .......................................................................................12
2.11.1 Off-time Selection and minimum on-time ........................................................................14
2.11.2 Slow Decay Mode ...........................................................................................................14
2.12 Over Current Detection .........................................................................................................16
2.13 Power Management ...............................................................................................................20
2.13.1 Maximum output current vs. selectable devices..............................................................20
2.13.2 Power Dissipation Formulae ...........................................................................................21
2.14 The decoding logic ................................................................................................................24
2.15 Tacho Output and Speed Loop .............................................................................................25
2.15.1 Static performance - Speed Regulation vs. Resistant Torque: .......................................28
2.15.2 Dynamic performance: ....................................................................................................29
2.15.3 Loop Stability: ..................................................................................................................30
2.15.4 Reference voltage ripple: ................................................................................................30
2.16 Brake.....................................................................................................................................31
APPLICATION EXAMPLE................................................................................................................32
APPENDIX - EVALUATION BOARDS .............................................................................................34
4.1 PractiSPIN..............................................................................................................................34
4.2 EVAL6235N ...........................................................................................................................35
4.2.1 Important Notes ................................................................................................................36
REFERENCES.................................................................................................................................39
AN1625 APPLICATION NOTE
2
DESIGNING AN APPLICATION WITH L6235
2.1 Current Ratings
With MOSFET (DMOS) devices, unlike bipolar transistors, current under short circuit conditions is, at first approximation, limited by the RDS(ON) of the DMOS themselves and could reach very high values. L6235 Out pins
and the two VSA and VSB pins are rated for a maximum of 2.8 A r.m.s. and 5.6 A peak (typical values). These
values are meant to avoid damaging metal structures, including the metallization on the die and bond wires. In
practical applications, though, maximum allowable current is less than these values, due to power dissipation
limits (see Power Management section).
The device has a built-in Over Current Detection (OCD) that allows protection against short circuits between the
outputs and between an output and ground (see Over Current Detection Section).
2.2 Voltage Ratings and Operating Range
The L6235 requires a single supply voltage (VS), for the motor supply. Internal voltage regulators provide the
5V and 10 V required for the internal circuitry. The operating range for VS is 8 to 52 V. To prevent working into
undesirable low supply voltage an Under Voltage Lock Out (UVLO) circuit shuts down the device when supply
voltage falls below 6 V; to resume normal operating conditions, VS must then exceed 7 V. The hysteresis is provided to avoid false intervention of the UVLO function during fast VS ringings. It should be noted, however, that
DMOS's RDS(ON) is a function of the VS supply voltage. Actually, when VS is less than 10V, RDS(ON) is adversely
affected, and this is particularly true for the High Side DMOS that are driven from VBOOT supply. This supply is
obtained through a charge pump from the internal 10V supply, which will tend to reduce its output voltage when
VS goes below 10V. Figure 2 shows the supply voltage of the high side gate drivers (VBOOT - VS) versus the
supply voltage (VS).
Figure 2. High side gate drivers supply voltage versus supply voltage.
8
7 .6
VBOOT - VS
[V]
7 .2
6 .8
6 .4
6
8
8 .5
9
9 .5
10
1 0 .5
VS [V]
Note that VS must be connected to both VSA and VSB because the bootstrap voltage (at VBOOT pin) is the same
for the two H-bridges. The integrated DMOS have a rated Drain-Source breakdown voltage of 60 V. However
VS should be kept below 52 V, since in normal working conditions the DMOS see a Vds voltage that will exceed
VS supply. In particular when a high-side DMOS turns off due to a phase change (OUT1 in Figure 3), if one of
the other outputs (OUT2 in Figure 3) is high (during the off-time all active bridges turn their high-side on) the
load current starts flowing in the low-side freewheeling diode and the SENSE pin sees a negative spike due to
a not negligible parasitic inductance of the PCB path from the pin to GND. This spike is followed by a stable
negative voltage due to the drop on RSENSE. The output pin sees a similar behavior, but with a slightly larger
voltage due to the forward recovery time of the integrated freewheeling diode and the forward voltage drop
across it. Typical duration of this spike is 30 ns. At the same time, the OUT2 pin (in the example of Figure 3)
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AN1625 APPLICATION NOTE
sees a voltage above VS, due to voltage drop across the high-side (integrated) freewheeling diode, as the current reverses direction and flows into the bulk capacitor. It turns out that the highest differential voltage is observed between two OUT pins when a phase change turns a high-side off during an off-time, and this must
always be kept below 60 V [2].
Figure 3. Currents and voltages if a phase change turns a high-side off during off-time.
Bulk
Capacitor
Equivalent
Circuit
ESR
Current starts
flowing in the
third half bridge
PCB Parasitic
Inductance
VS
ESL
RSENSE*I+ V F(Diode)
RSENSE*I
PCB Parasitic
Inductance
on-time
off-time
during off-time
a phase change can occur
Figure 4 shows the voltage waveforms at the OUT pins referring to a possible practical situation, with a peak
output current of 2.8 A, VS = 52 V, RSENSE = 0.33 Ω, TJ = 25 °C (approximately) and a good PCB layout. Below
ground spike amplitude is -2.64 V for one output; the other OUT pin is at about 55 V. In these conditions, total
differential voltage reaches almost 60 V, which is the absolute maximum rating for the DMOS. Keeping differential voltage between two Output pins within rated values is a must that can be accomplished with proper selection of Bulk capacitor value and equivalent series resistance (ESR), according to current peaks and adopting
good layout practices to minimize PCB parasitic inductances (see below) [2].
Figure 4. Voltage at the two outputs if a phase change turns a high-side off during off-time.
OUT1
OUT2
SENSE
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AN1625 APPLICATION NOTE
2.3 Choosing the Bulk Capacitor
Since the bulk capacitor, placed between VS and GND pins, is charged and discharged during IC operation, its
AC current capability must be greater than the r.m.s. value of the charge/discharge current. This current flows
from the capacitor to the IC during the on-time (tON) and from the IC (during some phase changes; from the
power supply during off-time) to the capacitor during the off-time (tOFF). The r.m.s. value of the current flowing
into the bulk capacitor depends on peak output current, output current ripple, switching frequency, duty-cycle.
It also depends on power supply characteristics. A power supply with poor high frequency performances (or
long, inductive connections to the IC) will cause the bulk capacitor to be recharged slowly: the higher the current
control switching frequency, the higher the current ripple in the capacitor; r.m.s. current in the capacitor, however, does not exceed the r.m.s. output current. Bulk capacitor value (C) and the ESR determine the amount of
voltage ripple on the capacitor itself and on the IC. Neglecting the output current ripple and assuming that during
the on-time the capacitor is not recharged by the power supply, the voltage at the end of the on-time is
tO N
V S – I O UT ⋅  ESR + -------C 
where IOUT is the output current. Usually (if C>100 µF) the capacitance role is much less than the ESR, then
supply voltage ripple can be estimated as
IO UT ⋅ ESR
For Example, if a maximum ripple of 500 mV is allowed and IOUT = 2 A, the capacitor ESR should be lower than
0.5V
ESR < ------------ = 250mΩ
2A
Note that additional ripple is due to parasitic inductances on VS PCB tracks (see Voltage Ratings and Operating
Range section).
Actually, current sunk by VSA and VSB pins of the device is subject to higher peaks due to reverse recovery
charge of internal freewheeling diodes. Duration of these peaks is, tough, very short (100÷200 ns) and can be
filtered using a small value (100÷200 nF), good quality ceramic capacitor, connected as close as possible to the
VSA, VSB and GND pins of the IC. Bulk capacitor will be chosen with maximum operating voltage 25% greater
than the maximum supply voltage, considering also power supply tolerances. For example, with a 48 V nominal
power supply, with 5% tolerance, maximum voltage is 50.4 V, then operating voltage for the capacitor should
be at least 63 V.
2.4 Layout Considerations
Working with devices that combine high power switches and control logic in the same IC careful attention has
to be paid to the PCB layout. In extreme cases, Power DMOS commutation can induce noises that could cause
improper operation in the logic section of the device. Noise can be radiated by high dv/dt nodes or high di/dt
paths, or conducted through GND or Supply connections. Logic connections, especially high-impedance nodes
(actually all logic inputs, see further), must be kept far from switching nodes and paths. With the L6235, in particular, external components for the charge pump circuitry should be connected together through short paths,
since these components are subject to voltage and current switching at relatively high frequency (600 kHz). Primary mean in minimizing conducted noise is working on a good GND layout (see Figure 5).
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AN1625 APPLICATION NOTE
Figure 5. Typical Application and Layout suggestions.
3-Phase
Brushless
DC Motor
D1
D2
C8
C5
R4
OUT1 OUT2 OUT3
VBOOT
CP
VSA
VSB
H1
+
Logic Supply
3.3 ÷ 5 V
SENSE A
H2
R5
FWD/REV
or
Custom Logic
+
SENSE B
H3
µC
L6235
BRAKE
C6
C7
VS = 8 ÷ 52 V
TACHO
-
DIAG
-
EN
R1
RCOFF
Vr ef
RCPULSE
GND GND GND GND
C1
C3
C2
R2
C4
R3
Vref = 0 ÷1V
High current GND tracks (i.e. the tracks connected to the sensing resistor) must be connected directly to the negative terminal of the bulk capacitor. A good quality, high-frequency bypass capacitor is also required (typically a
100 nF÷200 nF ceramic would suffice), since electrolytic capacitors show a poor high frequency performance.
Both bulk electrolytic and high frequency bypass capacitors have to be connected with short tracks to VSA, VSB
and GND. On the L6235 GND pins are the Logic GND, since only the quiescent current flows through them. Logic
GND and Power GND should be connected together in a single point, the bulk capacitor, to keep noise in the
Power GND from affecting Logic GND. Specific care should be paid layouting the path from the SENSE pins
through the sensing resistor to the negative terminal of the bulk capacitor (Power Ground). These tracks must be
as short as possible in order to minimize parasitic inductances that can cause dangerous voltage spikes on
SENSE and OUT pins (see the Voltage Ratings and Operating Range section); for the same reason the capacitors on VSA, VSB and GND pins should be very close to the GND and supply pins. Refer to the Sensing Resistors
section for information on selecting the sense resistors. Traces that connect to VSA, VSB, SENSEA, SENSEB, and
the three OUT pins must be designed with adequate width, since high currents are flowing through these traces,
and layer changes should be avoided. Should a layer change prove necessary, multiple and large via holes have
to be used. A wide GND copper area can be used to improve power dissipation for the device.
Figure 6 shows two typical situations that must be avoided. An important consideration about the location of the
bulk capacitor is the ability to absorb the inductive energy from the load, without allowing the supply voltage to
exceed the maximum rating. The diode shown in Figure 6 prevents the recirculation current from reaching the
capacitors and will result in a high voltage on the IC pins that can destroy the device. Having a switch or a power
connection that can disconnect the capacitors from the IC, while there is still current in the motor, will also result
in a high voltage transient since there is no capacitance to absorb the recirculation current.
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AN1625 APPLICATION NOTE
Figure 6. Two situations that must be avoided.
VSA
DON’T put a diode here!
Recirculating current cannot flow into the
bulk capacitor and causes a high voltage
spike that can destroy the IC.
VSB
SENSE A
SENSE B
L6235
+
R5
C6
C7
VS = 8 ÷ 52 V
GND GND GND GND
-
DON’T connect the Logic GND here
Voltage drop due to current in sense
path can disturb logic GND.
2.5 Sensing Resistor
Motor winding current flows through the sensing resistor, causing a voltage drop that is used, by the logic, to
control the peak value of the load current. Two issues must be taken into account when choosing the RSENSE
value:
■
The sensing resistor dissipates energy and provides dangerous negative voltages on the SENSE pins
during the current recirculation. For this reason the resistance of this component should be kept low.
■
The voltage drop across RSENSE is compared to the reference voltage (on Vref pin) by the internal comparator. The lower is the RSENSE value, the higher is the peak current error due to noise on Vref pin and
to the input offset of the current sense comparator: too small values of RSENSE must be avoided.
A good compromise is calculating the sensing resistor value so that the voltage drop, corresponding to the peak
current in the load (Ipeak), is about 0.5 V: RSENSE = 0.5 V / Ipeak.
It should be clear that sensing resistor must absolutely be non-inductive type in order to avoid dangerous negative spikes on SENSE pins. Wire wounded resistors cannot be used here, while Metallic film resistors are recommended for their high peak current capability and low inductance. For the same reason the connections
between the SENSE pins, C6, C7, VSA, VSB and GND pins (see Figure 5) must be taken as short as possible
(see also the Layout Considerations section).
The average power dissipated by the sensing resistor is:
2
PR ≈ I rms ⋅ R S ENSE ⋅ D ;
D is the duty-cycle of the PWM current control, Irms is the r.m.s. value of the load current.
Nevertheless, sensing resistor power rating should be chosen taking into account the peak value of the dissipated power:
2
P R ≈ Ipk ⋅ R S ENSE
,
where Ipk is the peak value of the load current.
Using multiple resistors in parallel will help obtaining the required power rating with standard resistors, and re-
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AN1625 APPLICATION NOTE
duce the inductance.
RSENSE tolerance reflects on the peak current error: 1% resistors should be preferred.
The following table shows RSENSE recommended values (to have 0.5V drop on it) and power ratings for typical
examples of current peak values.
Ipk
RSENSE Value [Ω]
RSENSE Power Rating [W]
0.5
1
0.25
1
0.5
0.5
1.5
0.33
0.75
3 X 1Ω, 0.25W paralleled
2
0.25
1
4 X 1Ω, 0.25W paralleled
Alternatives
2 X 1Ω, 0.25W paralleled
2.6 Charge pump external components
An internal oscillator, with its output at CP pin, switches from GND to 10 V with a typical frequency of 600 kHz
(see Figure 7).
Figure 7. Charge Pump.
VS + 10 V - VD1 - VD2
VS + 10V - VD1
VS - VD1
f = 600 kHz
C8
D1
C5
D2
R4
VBOOT
To High-Side
Gate Drivers
CP
VSA
VSB
10 V
RDS(ON) = 70 Ω
Charge Pump
Oscillator
10 V
5V
10 V
RDS (ON) = 70 Ω
f = 600 kHz
L6235
When the oscillator output is at ground, C5 is charged by VS through D2. When it rises to 10 V, D2 is reverse
biased and the charge flows from C5 to C8 through D1, so the VBOOT pin, after a few cycles, reaches the maximum voltage of VS + 10 V - VD1 - VD2, which supplies the high-side gate drivers.
With a differential voltage between VS and VBOOT of about 9V and the bridges switching at 50 kHz, the typical
current drawn by the VBOOT pin is 1.85mA.
Resistor R4 is added to reduce the maximum current in the external components and to reduce the slew rate of
the rising and falling edges of the voltage at the CP pin, in order to minimize interferences with the rest of the
circuit. For the same reason care must be taken in realizing the PCB layout of R4, C5, D1, D2 connections (see
also the Layout Considerations section). Recommended values for the charge pump circuitry are:
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AN1625 APPLICATION NOTE
D1, D2
: 1N4148
R4
: 100 Ω
C5
: 10 nF 100V ceramic
C8
: 220 nF 35V ceramic
(1/8 W)
Due to the high charge pump frequency, fast diodes are required. Connecting the cold side of the bulk capacitor
(C8) to VS instead of GND the average current in the external diodes during operation is less than 10 mA (with R4
= 100 Ω); at startup (when VS is provided to the IC) is less than 200 mA while the reverse voltage is about 10 V
in all conditions. 1N4148 diodes withstand about 200 mA DC (1 A peak), and the maximum reverse voltage is 75
V, so they should fit for the majority of applications.
2.7 Sharing the Charge Pump Circuitry
If more than one device is used in the application, it's possible to use the charge pump from one L6235 to supply
the VBOOT pins of several ICs. The unused CP pins on the slaved devices are left unconnected, as shown in
Figure 8. A 100 nF capacitor (C8) should be connected to the VBOOT pin of each device.
Supply voltage pins (VS) of the devices sharing the charge pump must be connected together.
The higher the number of devices sharing the same charge pump, the lower will be the differential voltage available for gate drive (VBOOT - VS), causing a higher RDS(ON) for the high side DMOS, so higher dissipating power.
In this case it's recommended to omit the resistor on the CP pin, obtaining a higher current capability of the
charge pump circuitry.
Better performance can also be obtained using a 33 nF capacitor for C5 and using schottky diodes (for example
BAT47).
Sharing the same charge pump circuitry for more than 3÷4 devices is not recommended, since it will reduce the
VBOOT voltage increasing the high-side MOS on-resistance and thus power dissipation.
Figure 8. Sharing the charge pump circuitry.
To other Devices
D1 = BAT47
D2 = BAT47
C8 = 100nF
C18 = 100 nF
C5 = 33nF
VBOOT
CP
VSA
VSB
VBOOT
CP
VSA
VSB
To High-Side
Gate Drivers
To High-Side
Gate Drivers
L6235
L6235
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AN1625 APPLICATION NOTE
2.8 Reference Voltage for PWM Current Control
The device has an analog input, Vref, connected to the internal sense comparator, to control the peak value of
the motor current through the integrated PWM circuitry. A fixed reference voltage can be easily obtained through
a resistive divider from an available 5 V voltage rail (maybe the one supplying the µC or the rest of the application) and GND.
A very simple way to obtain a variable voltage without using a DAC is to low-pass filter a PWM output of a µC
(see Figure 9).
Figure 9. Obtaining a variable voltage through a PWM output of a µC.
PWM Output
of a µC
RLP
Vref
RDIV
C LP
GND
Assuming that the PWM output swings from 0 to 5V, the resulting average voltage will be
5V ⋅ Dµ C ⋅ RDIV
Vref = ---------------------------------------R L P + R DIV
where DµC is the duty-cycle of the PWM output of the µC.
Assuming that the µC output impedance is lower than 1kΩ, with RLP = 56kΩ, RDIV = 15kΩ, CLP = 10nF and a
µC PWM switching from 0 to 5V at 100kHz, the low pass filter time constant is about 0.12 ms and the remaining
ripple on the Vref voltage will be about 20 mV. Using higher values for RLP, RDIV and CLP will reduce the ripple,
but the reference voltage will take more time to vary after changing the duty-cycle of the µC PWM, and too high
values of RLP will also increase the impedance of the Vref net at low frequencies, causing a poor noise immunity.
As sensing resistor value is typically kept small, a small noise on Vref input pins might cause a considerable error
in the output current. It's then recommended to decouple this pin with a ceramic capacitor of some tens of nF,
placed very close to Vref and GND pins. Note that Vref pin cannot be left unconnected, while, if connected to
GND, zero current is not guaranteed due to voltage offset in the sense comparator. The best way to cut down
(IC) power consumption and clear the load current is pulling down the EN pin. With very small reference voltage,
PWM integrated circuitry can loose control of the current due to the minimum allowed duration of tON (see the
Programmable off-time Monostable section).
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AN1625 APPLICATION NOTE
2.9 Input Logic pins
H1, H2, H3, FWD/REV, BRAKE, ENABLE, are CMOS/TTL compatible logic input pins. The input comparator
has been realized with hysteresis to ensure the required noise immunity. Typical values for turn-on and turn-off
thresholds are VTH(ON) = 1.8 V and VTH(OFF) = 1.3 V. Pins are ESD protected (see Figure 10) (2kV human-body
electro-static discharge), and can be directly connected to the logic outputs of a µC; a series resistor is generally not
recommended, as it could help inducted noise to disturb the inputs. All logic pins enforce a specific behavior and cannot be left unconnected. If connected to the DIAG pin, EN pin must be driven through a series resistor of 2.2 kΩ minimum (for 5 V logic), to allow the voltage at the pin to be pulled below the turn-off threshold (see below).
Figure 10. Logic input pins.
5V
H1, H2 H3,
FWD/REV , BRAKE,
ENABLE
ESD
Protection
2.10 DIAG pin
DIAG pin is an open-drain output pulled to GND in case of overcurrent or over temperature conditions. Connecting this pin to EN will allow the internal open drain to disable all the power DMOS of the L6235, provided that
the EN pin is driven through a resistor (see Input Logic pins).
A capacitor (C1 in Figure 5 and Figure 11) connected between EN and DIAG pins and GND is also recommended, to reduce the r.m.s. value of the output current when overcurrent conditions persist (see Over Current Protection section).
Figure 11. DIAG pin.
µC or Logic
Output
EN
R1
DIAG
C1
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AN1625 APPLICATION NOTE
2.11 Programmable off-time Monostable
The L6235 includes a constant off time PWM Current Controller. The current control circuit senses the bridge
current by sensing the voltage drop across an external sense resistor connected between the source of the
three lower power MOS transistors and ground, as shown in Figure 12. As the current in the motor increases
the voltage across the sense resistor increases proportionally. When the voltage drop across the sense resistor
becomes greater than the voltage at the reference input pin VREF the sense comparator triggers the
monostable switching the bridge off. The power MOS remain off for the time set by the monostable and the motor current recirculates around the upper half of the bridge in Slow Decay Mode as described in the next section.
When the monostable times out, the bridge will again turn on. Since the internal dead time, used to prevent
cross conduction in the bridge, delays the turn on of the power MOS, the effective Off Time tOFF is the sum of
the monostable time plus the dead time.
Figure 13 shows the typical operating waveforms of the output current, the voltage drop across the sensing resistor, the pin RC voltage and the status of the bridge. More details regarding the Synchronous Rectification and
the output stage configuration are included in the next section.
Immediately after the Power MOS turn on, a high peak current flows through the sense resistor due to the reverse recovery of the freewheeling diodes. The L6235 provides a 1µs Blanking Time tBLANK that inhibits the
comparator output so that the current spike cannot prematurely retrigger the monostable.
Figure 12. PWM Current Controller Simplified Schematic
VSB
VSA
BLANKING TIME
MONOSTABLE
TO GATE
LOGIC
1µs
5mA
FROM THE
LOW-SIDE
GATE DRIVERS
MONOSTABLE
SET
S
(0)
BLANKER
OUT2
Q
(1)
OUT3
R
DRIVERS
+
DEAD TIME
-
DRIVERS
+
DEAD TIME
+
5V
2.5V
OUT1
DRIVERS
+
DEAD TIME
+
SENSE
COMPARATOR
COFF
-
RCOFF
VREF
ROFF
RSENSE
SENSEB
SENSEA
D02IN1380
12/39
VS
AN1625 APPLICATION NOTE
Figure 13. Output Current Regulation Waveforms
IOUT
VREF
RSENSE
tON
tOFF
tOFF
1µs tBLANK
VSENSE
1µs tBLANK
VREF
0
VRC
Slow Decay
Slow Decay
tRCRISE
tRCRISE
5V
2.5V
tRCFALL
tRCFALL
1µs tDT
1µs tDT
ON
OFF
SYNCHRONOUS RECTIFICATION
D02IN1351
B
C
D
A
B
C
D
Figure 14 shows the magnitude of the Off Time tOFF versus COFF and ROFF values. It can be approximately
calculated from the equations:
tRCFALL = 0.6 · ROFF · COFF
tOFF = tRCFALL + tDT = 0.6 · ROFF · COFF + tDT
where ROFF and COFF are the external component values and tDT is the internally generated Dead Time with:
20KΩ ≤ ROFF ≤ 100KΩ
0.47nF ≤ COFF ≤ 100nF
tDT = 1µs (typical value)
Therefore:
tOFF(MIN) = 6.6µs
tOFF(MAX) = 6ms
These values allow a sufficient range of tOFF to implement the drive circuit for most motors.
The capacitor value chosen for COFF also affects the Rise Time tRCRISE of the voltage at the pin RCOFF. The
Rise Time tRCRISE will only be an issue if the capacitor is not completely charged before the next time the
monostable is triggered. Therefore, the On Time tON, which depends by motors and supply parameters, has to
be bigger than tRCRISE for allowing a good current regulation by the PWM stage. Furthermore, the On Time tON
can not be smaller than the minimum on time tON(MIN).
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AN1625 APPLICATION NOTE
 t O N > t O N ( MIN ) = 1.5 µ s (typ. value)

 t O N > t RCRISE – t DT
tRCRISE = 600 · COFF
2.11.1 Off-time Selection and minimum on-time
Figure 14 also shows the lower limit for the On Time tON for having a good PWM current regulation capacity. It
has to be said that tON is always bigger than tON(MIN) because the device imposes this condition, but it can be
smaller than tRCRISE - tDT. In this last case the device continues to work but the Off Time tOFF is not more constant.
So, small COFF value gives more flexibility for the applications (allows smaller On Time and, therefore, higher
switching frequency), but, the smaller is the value for COFF, the more influential will be the noises on the circuit
performance.
Figure 14. Off-time selection and minimum on-time.
4
1 . 10
R = 100 kΩ
R = 47 kΩ
to f f [ u s]
1 . 10
R = 20 kΩ
3
100
10
1
0.1
1
10
100
10
100
Coff [nF]
to n ( m in ) [ u s]
100
10
1
0.1
1
Coff [nF]
2.11.2 Slow Decay Mode
Figure 15 shows the operation of the bridge in the Slow Decay mode during the Off Time. At any time only two
legs of the three-phase bridge are active, therefore only the two active legs of the bridge are shown in the figure
and the third leg will be off. At the start of the Off Time, the lower power MOS is switched off and the current
14/39
AN1625 APPLICATION NOTE
recirculates around the upper half of the bridge. Since the voltage across the coil is low, the current decays slowly. After the Dead Time the upper power MOS is operated in the synchronous rectification mode reducing the
impedance of the freewheeling diode and the related conducting losses. When the monostable times out, upper
MOS that was operating the synchronous mode turns off and the lower power MOS is turned on again after
some delay set by the Dead Time to prevent cross conduction.
Figure 15. Slow Decay Mode Output Stage Configurations
A) ON TIME
B) 1µs DEAD TIME
D01IN1336
C) SYNCHRONOUS
RECTIFICATION
D) 1µs DEAD TIME
In some conditions (short off-time, very low regulated current, high motor winding L / R) the system may need
an on-time shorter than 1.5 µs. In these cases the PWM current controller can loose the regulation.
Figure 16 shows the operation of the circuit in this condition. When the current first reaches the threshold, both
the high-side are turned on for a fixed time and the current decays.
During the following on-time current increases above the threshold, but the bridge cannot be turned off until the
minimum on-time expires. Since current increases more in each on-time than it decays during the off-time, it
keeps growing during each cycle, with steady state asymptotic value set by duty-cycle and load DC resistance:
the resulting peak current will be Ipk = VS × D / RLOAD, where D = tON / (tON+tOFF) is the duty-cycle and RLOAD
is the load resistance.
Figure 16. Minimum on-time can cause the PWM controller to loose the regulation.
minimum tON
is about 1.5 µs
Vref / RSENSE
needed tON is
less than 1.5 µs
15/39
AN1625 APPLICATION NOTE
2.12 Over Current Detection
To implement an Over Current (i.e. short circuit) Protection, a dedicated Over Current Detection (OCD) circuitry
(see Figure 17 for a simplified schematic) senses the current in each high side. Power DMOS are actually made
up with thousands of individual identical cells, each carrying a fraction of the total current flowing. The current
sensing element, connected in parallel to the Power DMOS, is made only with few such cells, having a 1:N ratio
compared to the power DMOS. The total drain current is split between the output and the sense element according to the cell ratio. Sensed current is, then, a small fraction of the output current and will not contribute
significantly to power dissipation.
Figure 17. Over Current Detection simplified circuitry.
OUT1
VSA
HIGH SIDE DMOS
µC or LOGIC
VDD
REN
VSB
HIGH SIDE DMOS
I2
POWER DMOS
n cells
POWER DMOS
n cells
I3
POWER SENSE
1 cell
POWER DMOS
n cells
POWER SENSE
1 cell
+
OCD
COMPARATOR
EN
OUT3
HIGH SIDE DMOS
I1
POWER SENSE
1 cell
TO GATE
LOGIC
OUT2
I1 / n
I2/ n
I1+I2 / n
CEN
INTERNAL
OPEN-DRAIN
DIAG
RDS(ON)
40Ω TYP.
IREF
OVER TEMPERATURE
I3/ n
IREF
D02IN1381
This sensed current is compared to an internally generated reference to detect an over current condition. An
internal open drain mosfet turns on when the sum of the currents in the bridges 1 and 2 or the current in the
bridge 3 reaches the threshold (5.6A typical value); the open drain is available at the DIAG pin for diagnostic
purposes or to ensure an over current protection, connecting EN and DIAG together and using an RC network
(see Figure 17).
16/39
AN1625 APPLICATION NOTE
Figure 18. Over Current Operation after a short circuit between an OUT pin and GND. EN and DIAG
pins are connected together.
tDELAY
tDISABLE
tOCD(ON)
Output Current
Output Current
IS OVER
tEN(FALL)
tD(OFF)EN
VTH(OFF)
tOCD(OFF)
tD(ON)EN
tDISABLE
EN = DIAG
VTH(ON)
VEN(LOW)
EN = DIAG
tEN(RISE)
Figure 18 shows the device operating in overcurrent condition (short to ground). When an over current is detected the internal open drain mosfet pulls the EN pin to GND switching off all 6 power DMOS of the device and
allowing the current to decay. Under a persistent over current condition, like a short to ground or a short between
two output pins, the external RC network on the EN pin (see Figure 17) reduces the r.m.s. value of the output
current by imposing a fixed disable-time after each over current occurrence. The values of REN and CEN are
selected to ensure proper operation of the device under a short circuit condition. When the current flowing
through the high side DMOS reaches the OCD threshold (5.6 A typ.), after an internal propagation delay
(tOCD(ON)) the open drain starts discharging CEN. When the EN pin voltage falls below the turn-off threshold
(VTH(OFF)) all the Power DMOS turn off after the internal propagation delay (tD(OFF)EN). The current begins to
decay as it circulates through the freewheeling diodes. Since the DMOS are off, there is no current flowing
through them and no current to sense so the OCD circuit, after a short delay (tOCD(OFF)), switches the internal
open drain device off, and REN can charge CEN. When the voltage at EN pin reaches the turn-on threshold
(VTH(ON)), after the tD(ON)EN delay, the DMOS turn on and the current restarts. Even if the maximum output current can be very high, the external RC network provides a disable time (tDISABLE) to ensure a safe r.m.s. value
(see Figure 18).
The maximum value reached by the current depends on its slew-rate, so on the short circuit nature and supply
voltage, and on the total intervention delay (tDELAY). It can be noticed that after the first current peak, the maximum value reached by the output current becomes lower, because the capacitor on EN and DIAG pins is discharged starting from a lower voltage, resulting in a shorter tDELAY.
The following approximate relations estimate the disable time and the first OCD intervention delay after the short
circuit (worst case).
The time the device remains disabled is:
t DIS ABLE = tOCD ( OF F ) + tEN ( RISE ) + tD ( O N )E N
VDD – V EN ( LOW )
tE N ( RISE ) = REN ⋅ CEN ⋅ ln  ------------------------------------------ V

DD – V T H ( O N )
VEN(LOW) is the minimum voltage reached by the EN pin, and can be estimated with the relation:
17/39
AN1625 APPLICATION NOTE
t
V EN ( LO W ) = V T H ( O FF ) ⋅ e
+t
D ( O FF )
O CD ( O FF )
 ------------------------------------------------

R
⋅C
O PD R
EN
The total intervention time is
t DE LAY = t OCD ( O N ) + tE N ( F ALL ) + t D ( O FF )EN
where
VDD
tE N ( F ALL ) = RO PDR ⋅ CE N ⋅ ln  -------------------------
V

T H ( OF F )
tOCD(OFF), tOCD(ON), tD(ON)EN, tD(OFF)EN, and ROPDR are device intrinsic parameters, VDD is the pull-up voltage
applied to REN.
The external RC network, CEN in particular, must be chosen obtaining a reasonable fast OCD intervention (short
tDELAY) and a safe disable time (long tDISABLE). Figure 19 shows both tDISABLE and tDELAY as a function of CEN:
at least 100µs for tDISABLE are recommended, keeping the delay time below 1÷2µs at the same time.
The internal open drain can also be turned on if the device experiences an over temperature (OVT) condition.
The OVT will cause the device to shut down when the die temperature exceeds the OVT threshold (TJ>165 °C
typ.).
Since the OVT is also connected directly to the gate drive circuit (see Figure 1), all the Power DMOS will shut
down, even if EN pin voltage is still over Vth(OFF). When the junction temperature falls below the OVT turn-off
threshold (150 °C typ.), the open drain turns off, CEN is recharged up to VTH(ON) and then the PowerDMOS are
turned on back.
18/39
AN1625 APPLICATION NOTE
Figure 19. Typical disable and delay time as a function of C EN, for several values of REN.
R
3
1 .1 0
EN
= 220 kΩ
R
EN
= 100 kΩ
R
R
tDISABLE [µs]
R
EN
= 47 kΩ
= 33 kΩ
EN
= 10 kΩ
EN
100
10
1
1
10
C
EN
C
EN
100
[n F ]
tDELAY [µs]
10
1
0 .1
1
10
100
[n F ]
19/39
AN1625 APPLICATION NOTE
2.13 Power Management
Even when operating at current levels well below the maximum ratings of the device, the operating junction temperature must be kept below 125 °C.
Figure 20 shows the IC dissipated power versus the r.m.s. load current, in 4 different driving sequences, assuming the supply voltage is 24V.
Figure 20. IC Dissipated Power versus Output Current.
I1
IOUT
10
I2
8
PD [W]
6
IOUT
I3
IOUT
4
Test Condition s:
Supply Voltage = 24 V
2
0
0
0.5
1
1.5
2
2.5
3
IOUT [A]
No PWM
fSW = 30 kHz (slow decay)
2.13.1 Maximum output current vs. selectable devices
Figure 21 reports a performance comparison between L6229 (std. power) and L6235 (high power) for different
packages, with the following assumptions:
- Supply voltage: 24 V; Switching frequency: 30 kHz.
- Tamb = 25 °C, TJ = 125 °C.
- Maximum RDS(ON) (taking into account process spread) has been considered, @ 125 °C.
- Maximum quiescent current IQ (taking into account process spread) has been considered.
- PCB is a FR4 with a dissipating copper surface on the top side of 6 cm2 (with a thickness of 35 µm) for
SO and PowerDIP packages (D, N suffixes).
- PCB is a FR4 with a dissipating copper surface on the top side of 6 cm2 (with a thickness of 35 µm), 16
via holes and a ground layer for the PowerSO package (PD suffix).
- For each device (on the x axis) y axis reports the maximum output current.
20/39
AN1625 APPLICATION NOTE
Figure 21. Maximum output current vs. selectable devices.
2.30
2.10
1.90
1.70
Load Current 1 .50
[A]
1 .30
1 .1 0
0 .9 0
0 .7 0
5 PD
L62 3
5N
L 623
5D
L62 3
9 PD
L62 2
9N
L 622
L 622
9D
0 .5 0
2.13.2 Power Dissipation Formulae
Figure 22 to Figure 25 are screenshots of a spreadsheet that helps calculating power dissipation in specified
conditions (application and motor data), and estimates the resulting junction temperature for a given package
and copper area available on the PCB [3].
The model considers power dissipation during the on-time and the off-time, taking into account the selected decay, rise and fall time (when a phase change occurs), the switching losses and the quiescent current power dissipation.
21/39
AN1625 APPLICATION NOTE
Figure 22. Current in the three phases and the signal of one of the hall effect sensors.
( f HALL = fEL = n * fmech )
1 / fHALL = T
Hall
T = 1 / f EL
I1
∆I
I pk
I
trise
I2
tfall
∆I
Ipk
I
trise
I3
tfall
∆I
I pk
I
t rise
tfall
Figure 23. Input Data.
Input D ata
Device Input Values
Maximum Drain-Source ON Resistance
Ron
=
5.60E-01
[Ω]
Max imum diode voltage
Vd
=
1.20E+00
[V]
Quiescent Current
Iq
=
5.50E-03
[A]
Motor Input Values
Maximum BEMF Voltage
Vb
=
1.00E+01
[V]
Motor Inductance
Lm
=
8.00E-04
[H]
Motor Resistance
Rm
=
2.10E+00
[Ω]
n
=
1
Polar Couples
-
Application Input Values
Supply Voltage
Peak Current
=
2.40E+01
[V]
Ipk
=
1.50E+00
[A]
tOFF
=
8.00E-06
[s]
Sensing Resistance
Rs
=
3.30E-01
[Ω]
Motor Speed
sp
=
1.00E+04
[rpm]
Off-Time
22/39
Vs
Average Value
between
High-Side and
Low-Side
AN1625 APPLICATION NOTE
Figure 24. Power Dissipation formulae and results.
Result
PowerDMOS
Commutation Time
Electrical frequency
Tcom = 9.60E-08
[s]
Vs / (250V/µs)
Fel = 1.67E+02 [Hz]
n*sp/60
Rise Time
Trise =
5.65E-5
[s]
( – Ipk ⋅ Rm – 2 ⋅ 2Ipk ⋅ Ron – Ipk ⋅ R s + V s )
Lm
– ln ------------------------------------------------------------------------------------------------------------------- ⋅ ---------------------------------------------Vs
Rm + Rs + 2Ron
Fall Time
Tfall = 5.13E-05
[s]
Vs – 2 ⋅ Vd
Lm
– ln ------------------------------------------------------------------------------------------------ ⋅ ---------------------------( Ipk ⋅ Rm + Ipk ⋅ Rs + Vs – ( 2 ⋅ Vd ) ) ( Rm + Rs )
-
Vb+I(2*Ron+Rm))/(Vs -I*Rs)
Duty Cycle
Switching
Frequency
D = 6.08E-01
fSW = 4.90E+04 [Hz]
(1-D) / tOFF
Current Ripple
∆I = 3.19E-01
[A]
2.1*((2Ron+Rm)*Ipk+Vb)*toff/Lm
Period
T = 6.00E-03
[s]
1 / fel
Tload = 5.66E-03
[s]
T-6Trise
I = 1.34E+00
[A]
Ipk – ∆I
----2
r.m.s. Current
during Load
Time
Irms = 1.34E+00
[A]
Rise Time
Dissipating
power
Prise = 1.58E-02
[W]
Fall Time
Dissipating
power
Pfall = 3.00E-02
[W]
Load Time
Average Current
during Load
Time
2
∆I Ipk ⋅ ( Ipk – ∆I ) + ------3
 2Ron

2 Trise
2
⋅ Ipk ⋅ --------------- ⋅ --3  T
– Vs + 2 ⋅ Vd )
--2- ⋅ 2 ⋅ V d Tfall ⋅ (-------------------------------------+
T
( Rm + Rs )
– T fall
1 – exp ---------------- ⋅ ( Rm + Rs )
Lm
Lm ⋅ ( Ipk ⋅ Rm + Ipk ⋅ Rs + Vs – 2 ⋅ Vd ) ⋅ --------------------------------------------------------------------------------2
( Rm + Rs )
Load Time
Diss. Power
Pload = 1.91E-00
[W]
(2Ron · Irms2 · Tload) / T
Commutation
Dissipating Pw
Pcom = 2.86E-01
[W]
(2Vs · I · Tcom · Tload · fSW) / T
Quiescent
Dissipating Pw
Pq = 1.32E-01
[W]
Vs · Iq
Total Dissipating Power
P = 2.37E+00
[W]
Pq + Pcom + Pload + Pfall + Prise
23/39
AN1625 APPLICATION NOTE
Figure 25. Thermal Data inputs and results
Input Data
Package
PowerSO36
Copper Area
5.0
Copper Area is on
Same side of the device
Ground Layer
YES
Ambient Temperature
50
1÷10 sq. cm
-25 ÷ 100 ºC
Results
Thermal Resistance
Junction to Ambient
20.81
ºC / W
Thermal Resistance
Junction to Pins / Slug
1.00
ºC / W
Estimated Junction
Temperature
99.35
ºC
Estimated
Pins / Slug Temperature
96.98
ºC
2.14 The decoding logic
The L6235 integrated decoding logic provides the correct sequence on the three outputs for motors with both
60° and 120° spaced hall effect sensors signals. The sensors' outputs are directly connected to the H1, H2, H3
inputs of the device. The table below reports the output configurations for all possible hall effect input signals.
Hall 120°
1
2
3a
-
4
5
6a
-
Hall 60°
1
2
-
3b
4
5
-
6b
H1
H
H
L
H
L
L
H
L
H2
L
H
H
H
H
L
L
L
H3
L
L
L
H
H
H
H
L
OUT1
Vs
High Z
GND
GND
GND
High Z
Vs
Vs
OUT2
High Z
Vs
Vs
Vs
High Z
GND
GND
GND
OUT3
GND
GND
High Z
High Z
Vs
Vs
High Z
High Z
Phasing
1->3
2->3
2->1
2->1
3->1
3->2
1->2
1->2
24/39
AN1625 APPLICATION NOTE
2.15 Tacho Output and Speed Loop
H1 input is internally connected to a monostable that provides, through an open drain mosfet, a fixed width pulse
on the tacho output (see Figure 26). Through this output realizing a speed loop is very easy and inexpensive.
Providing an external pull-up resistor on this open drain output, the resulting waveform at the pin will be a
square-wave whose frequency is proportional to the motor rotation speed, with a fixed on-time (tPULSE) set by
an external RC network connected at the RCPULSE pin.
Figure 26. Tacho monostable.
H1
Hall effect sensor 1
from the motor
to the Decoding Logic
RCPULSE
Tacho
Monostable
GND
tPULSE
+5V
TACHO
Tacho monostable is identical to RCoff monostable, and the fixed pulse time is defined by:
t p uls e ≅ 0.6R PULSE C PULS E
Figure 27. tacho pulses selection.
4
1 .10
R PUL = 100kΩ
R PUL = 47kΩ
3
1 .10
tpulse [µs]
RPUL = 20kΩ
100
10
1
10
Cpul [nF]
100
25/39
AN1625 APPLICATION NOTE
Also the duty cycle of this signal, so its average value is proportional to the motor rotation speed. Simply integrating the square-wave a voltage proportional to the motor speed will be available to realize a speed loop, as
in Figure 28: RPULSE and CPULSE define the fixed on-time (tPULSE) of the tacho output, integrated and compared
to a voltage proportional to the desired speed (Vspeed) by the op-amp; the output of the op-amp represents the
speed error signal. Providing this signal to the Vref input of the L6235, which sets the current in the motor windings, the speed error will act on the motor modifying its torque, in order to maintain the speed at a constant value.
R1 and R2 set the maximum current in the motor by limiting the voltage at the Vref pin.
Figure 28. Tacho output allows easy implementation of a speed loop.
BLDC Motor
H1
H2
H3
RCPULSE
OUT1 OUT2 OUT 3
V REF
V ref
L6235
TAC HO SENSEA SENSEB
RSENSE
CPULSE
RPULSE
Cfb
V pullup
V OP-AM P
Rpullup
Rfb2
V TACHO
Rfb1
R1
tPULSE
V speed
R2
Defining
KT
Motor torque constant
[Nm/A]
D
Dynamic friction torque
[Nms/rad]
J
Motor + Load inertia moment
[kgm2]
TR
Load resistant torque
[Nm]
τM = J/D
Mechanical time constant
[s]
n
Number of polar couples
ωm
Motor rotation speed
[rad/s]
ftacho = nωm / 2π
Tacho output frequency
[Hz]
τI = Cfb · Rfb2
Integrator time constant
[s]
Gop-amp = Rfb2 /Rfb1
Gw-v =Vpullup · n · tpulse / 2π
[Vs/rad]
G1=R2 / RSENSE · (R1+R2)
[1/Ω]
and neglecting the current ripple due to PWM control, the expression of the control loop transfer function (see
Figure 29) is:
26/39
AN1625 APPLICATION NOTE
– G op – amp ⋅ Gω – v ⋅ G 1 ⋅ k T
G loop ( s ) = ---------------------------------------------------------------------D ⋅ ( 1 + sτ I ) ⋅ ( 1 + sτ M )
Figure 29. Control loop block diagram.
V speed [V]
+
Rfb2
_
Rfb1
1
1+ sτI
+
+
<vOp-Amp>
[V]
R2
R1 + R2
1
<vref > [V]
Gop-amp
RSENSE
G1
<I pk, Motor> [A]
<vTACHO> [V]
Vpullup t pulse n
2π
G
ω-v
ω M [rad/s]
1/ D
1+ sτM
Gmotor
+
_
T [Nm]
kT
TR [Nm]
Can be noticed that since the motor is current controlled, the electrical time constant of the motor (L/R) does not
appear in any transfer function.
With the following values, module and phase of Gloop are shown in Figure 30.
KT = 9.8 mNm/A
D = 3.34 µNms/rad
J = 6.5 µkgm2
TR = 4 mNm
τM = J/D = 1.95 s (mechanical pole at 0.08 Hz)
n=2
ωm = 2618 rad/s (25000 r.p.m.)
ftacho = nωm / 2π = 833 Hz
Vpullup = 5V
tpulse = 1 ms
Rfb1 = 100 kΩ
Rfb2 = 1 MΩ
C = 33 nF
R1 = 5.6 kΩ
R2 = 1.8 kΩ
RSENSE = 0.33Ω
27/39
AN1625 APPLICATION NOTE
Figure 30. Gloop module and phase.
50
fc
|Gloop| [dB]
25
0
25
50
1 . 10
3
0.01
0.1
f [Hz]
1
10
100
10
100
phase of Gloop [ ° ]
0
fc
45
90
135
180
1 . 10
3
0.01
0.1
1
f [Hz]
The relation between the speed reference voltage (Vspeed), the load resistant torque (Tb) and the motor speed
(ωM) is given by the expression:
ωM
G op – a mp
V spee d ⋅  1 + ------------------------ ⋅ G 1 ⋅ kT – T b

1 + sτ I 
= ----------------------------------------------------------------------------------------------------------G op – amp
D + ( 1 ⋅ sτ M ) + Gω – v ⋅ G 1 ⋅ k T ⋅ -----------------------1 + sτ I
for a given speed, the speed reference voltage to apply is:
ω M ⋅ ( D + G ω – v ⋅ G1 ⋅ kT ⋅ G op – a mp ) + Tb
Vspeed = -----------------------------------------------------------------------------------------------------------( 1 + G op – amp ) ⋅ G1 ⋅ kT
Designing the speed loop, care must be taken choosing the values of Rpullup, Rfb1, Rfb2, Cfb, R1 and R2 obtaining
a good compromise between static performance, dynamic performance, stability and torque ripple:
2.15.1 Static performance - Speed Regulation vs. Resistant Torque:
The relation between Vspeed, Tb and ωM shows that for a fixed speed reference, the load torque (Tb) affects the
speed. To minimize the resistant torque effect the term (D + Gω-v · G1 · kT · Gop-amp) must be kept as high as
possible.
Figure 31 shows how the speed changes with the load torque, and percentage regulation error.
Due to the op-amp output voltage saturation, beyond a certain load torque value the system cannot produce
further torque, then the motor speed drastically decreases.
28/39
AN1625 APPLICATION NOTE
Figure 31. Regulated speed variations versus mechanical load.
4
3.5.10
0
2
Vref=4.67 V
% Speed Regulation Error
Regulated Speed
4
3 .10
Vref=4.17 V
4
2.5 .10
Vref=3.67 V
4
2 .10
4
1.5.10
Vref=4.17 V
4
6
8
10
0.005
0.01
0.015
0.02
0.025
0.03
0
0.005
Resistant Torque (Tb) [Nm]
0.01
0.015 0.02
0.025
Resistant Torque (Tb) [Nm]
0.03
2.15.2 Dynamic performance:
The loop bandwidth is the frequency range in which the loop gain is greater than 1 (0 dB). In the example it's
about 2.5 Hz. It expresses how fast will the loop regulate the speed after load changes.
The transfer function between the resistant torque and the motor speed is
–( 1 + sτ 1 )
-------------------------ωM ( s )
D
---------------------- = ------------------------------------------------------------------------------------------------------------------------------Tb ( s )
1
( 1 + sτ M ) ⋅ ( 1 + sτ I ) + ---- ⋅ G ω – v ⋅ G op – amp ⋅ G1 ⋅ kT
D
,
while between speed reference voltage and speed we have:
G o p – amp
 1 + ------------------------ ⋅ G1 ⋅ kT

ωM ( s )
1 + sτ I 
-------------------------- = ----------------------------------------------------------------------------------------------------------G op – amp
Vspe ed ( s )
D ⋅ ( 1 + sτ M ) + Gω – v ⋅ G 1 ⋅ k T ⋅ ------------------------1 + sτ I
Figure 32 and Figure 33 show how the speed changes after applying a 1mN resistant torque step and a 1V
speed voltage step, respectively
29/39
AN1625 APPLICATION NOTE
Figure 32. Speed response to a 1mNm resistant torque step
0
Speed Variation [r.
p.m.]
20
40
60
80
100
0
0.1
0.2
0.3
0.4
t [s]
0.5
0.6
0.7
0.5
0.6
0.7
Figure 33. Speed response to a 1V speed voltage step.
8000
Speed Variation [r.p.m. ]
6400
4800
3200
1600
0
0
0.1
0.2
0.3
0.4
t [s]
2.15.3 Loop Stability:
The phase margin is defined as 180° minus the phase of Gloop at the cut frequency (where Gloop=1). It should
be at least 45° to guarantee the stability. In the example it's about 65°.
2.15.4 Reference voltage ripple:
Due to the tacho waveform integration, the reference voltage provided to the L6235 Vref input is a triangular
wave (see Figure 27). The ripple can be calculated through the approximate relation:
R2
speedref ⋅ ( 1 – DC )
∆Vre f = ------------------------------------------------------ ⋅ -------------------- ,
Rfb1 ⋅ ftacho ⋅ C fb
R1 + R 2
where DC is the duty-cycle of the square wave at the tacho output:
n ⋅ tpulse ⋅ ω m
DC = ---------------------------------2π
Since this reference voltage is the torque control voltage, the ripple should be kept as low as possible,
accordingly to others main parameters. In the example the ripple is about 60 mV.
30/39
AN1625 APPLICATION NOTE
The limits of this system depend on the fact that the speed information is obtained by an analog integration of
the TACHO output. This operation introduces a low frequency pole in the Gloop function, and a ripple in the reference voltage. The introduced pole, in conjunction with the very low frequency mechanical pole of the motor,
strongly affects loop stability, system bandwidth and static speed regulation error: to preserve stability a DC loop
gain (Gloop(0)) diminution may be needed. At the same time, decreasing the reference voltage ripple requires
to reduce the frequency of the integrator pole, cutting the bandwidth and reducing the phase margin.
A full digital approach to convert the tacho frequency in a voltage can give further improvement in static and
dynamic speed regulation: a microcontroller can measure the frequency of the tacho output and provide the reference voltage to the L6235 through a D/A converter or a low-pass filtered (see Reference Voltage section)
PWM output (whose frequency can be much higher than the hall effect signals frequency, resulting in a strongly
reduced Vref ripple).
Another possibility is using a PLL to generate a voltage proportional to the speed (or used directly, taking a frequency input as the command).
2.16 Brake
In general, motor braking can be achieved making a short circuit across the windings: the BEMF forces a current, proportional to the braking torque, that flows in the opposite direction than in normal running mode. For
high BEMF and inertia moment the current may reach very high values: a power resistor is often used to reduce
the maximum braking current and dissipate the motor energy.
L6235 Brake pin can be used to quickly stop the motor while it is running: providing a low logic level to this pin
all the high-side DMOS switch on, making a short-circuit across the motor windings.
A power resistor is not used: while the motor is braking, both Thermal and Over Current protections still work,
avoiding BEMF to cause a current exceeding the device's maximum ratings.
Connecting EN and DIAG pins together and using a RC network (see Over Current Detection section) a disable
time between each over current event can be set, reducing the maximum r.m.s. value of the current.
Figure 34 and Figure 35 show what happens if the current exceeds the OCD threshold while the motor is braking: as soon as the current in one of the three motor phases reaches the OCD threshold (5.6 A typ.) the open
drain mosfet internally connected to the DIAG pin discharges the external capacity; the EN pin voltage falls to
GND and all the bridges of the device are disabled for a time that depends on the RC network values. During
this disable time the current forced by the BEMF decreases, and so the braking torque; when the current becomes zero (because the motor inductances have been fully discharged), if the BEMF is less than the supply
voltage there is no braking effect (since the freewheeling diodes cannot be turned on) until the disable time expires and all the high side PowerDMOS turn on again
31/39
AN1625 APPLICATION NOTE
Figure 34. Overcurrent during motor braking.
on-time
braking
overcurrent – disable time
Figure 35. Overcurrent during motor braking.
I1
I2
“Brake” Command
I3
EN = DIAG
3
APPLICATION EXAMPLE
Application Data
Rotation Speed:
10000 rpm (fEL=167Hz)
Winding peak Current:
1.5 A
Maximum Ripple:
350mA
Maximum BEMF at 10000rpm:
10 V
32/39
AN1625 APPLICATION NOTE
Motor Data
Winding Resistance (2 phases):
2Ω
Winding Inductance (2 phases):
800 µH
Supply Voltage:
24 V +/- 5%
Polar Couples:
1
Referring to approximated formulae in Figure 24, it's possible to calculate the Duty Cycle (D), the Switching Frequency (fSW), the Current Ripple (∆I). With a 8 µs off-time, we will have:
D ≅ 61%, fSW ≅ 49 kHz, ∆I ≅ 320 mA.
The on-time is tON = D / fSW ≅ 12.5 µs, which is far from the minimum allowed (1.5 µs).
The bulk capacitor needs to withstand at least 24 V + 5% + 25% ≅ 32 V. A 50 V capacitor will be used. Allowing
a voltage ripple of 200 mV, the capacitor ESR should be lower than 200 mV / 1A = 200 mΩ; the AC current
capability should be about 1.5 A (worst case).
Providing a reference voltage of 0.5 V, 0.33 Ω sensing resistor are needed. The resistors power rating is about
PR ≅ Irms2 · RSENSE · D ≅ 0.37 W. Three 1 Ω / 0.25 W - 1% resistors in parallel are used. The charge pump
uses recommended components (1N4148 diodes, ceramic capacitors and a 100 Ω resistor to reduce EMI).
R = 24 kΩ, C = 470 pF are connected to the RC pins, obtaining tOFF ≅ 7.8 µs. On the EN pin a 5.6 nF has been
placed, and the pin is driven by the µC through a 100 kΩ resistor.
With these values, in case of short circuit between two OUT pins or an OUT pin and GND, the PowerDMOS
turns off after about 1 µs, and tDISABLE ≅ 240 µs.
Figure 36. Application Example.
3-Phase
Brushless DC Motor
1N4148
1N4148
10nF 50V
Ceramic
220nF 35V
Ceramic
100Ω
0.25W
OUT1 OUT2 OUT3
+
µC
VBOOT
CP
VSA
100µF 50V
ESR<200mΩ
VSB
H1
SENSE A
H2
SENSE B
+
H3
Logic Supply
or
5V
Custom Logic
CW/CCW
100nF 50V
Ceramic
L6235
BRAKE
-
VS = 24 V
EN
3X
1 Ω, 0.25 W, 1%
100 k Ω
DIAG
18 kΩ 0.25 W
1%
Vref
Vref = 0.5 V
2 kΩ 0.25 W
1%
5.6 nF
Ceramic
RCPULSE TACHO
RCOFF
-
GND GND GND GND
470 pF
Ceramic
47nF
Ceramic
24 kΩ
1%
Referring to Figure 23, Figure 24, Figure 25, the dissipating power is about 2.37 W. If the ambient temperature
is lower than 50 °C, with 5 cm2 of copper area on the PCB, a ground layer and a PowerSO36 package, the
estimated junction temperature is about 97 °C.
33/39
AN1625 APPLICATION NOTE
4
APPENDIX - EVALUATION BOARDS
4.1 PractiSPIN
PractiSPIN is an evaluation and demonstration system that can be used with the PowerSPIN family (L62XX) of
devices. A Graphical User Interface (GUI) (see Figure 37) program runs on an IBM-PC under windows and communicates with a common ST7 based interface board (see Figure 38) through the RS232 serial port. The ST7
interface board connects to a device specific evaluation board (target board) via a standard 34 pin ribbon cable
interface.
Depending on the target device the PractiSPIN can drive a stepper motor, 1 or 2 DC motors or a brushless DC
(BLDC) motor, operating significant parameters such as SPEED, CURRENT, VOLTAGE, DIRECTION, ACCELERATION and DECELERATION RATES from a user friendly graphic interface, and programming a sequence
of movements.
The software also allows evaluating the power dissipated by the selected device and, for a given package and
dissipating copper area on the PCB, estimates the device's junction temperature.
Figure 37. PractiSPIN PC Software
34/39
AN1625 APPLICATION NOTE
Figure 38. PractiSPIN ST7 Evaluation Board
4.2 EVAL6235N
An evaluation board has been produced to help the evaluation of the device in PowerDIP package. It implements a typical application with several added components. Figure 40 shows the electrical schematic of the
board; in the table below the part list is reported.
CN1, CN2
2-poles connector
JP1, JP2
2-pin jumper
CN3
3-poles connector
R1
700Ω 0.6W resistor
CN4
34-poles connector
R2,R3,R4,R7,R8,R9
10kΩ resistor
C1
220nF/100V Ceramic or Polyester capacitor
R5
100Ω resistor
C2
220nF/100V Ceramic or Polyester capacitor
R15,R6
1kΩ resistor
C3
100µF/63V capacitor
R11,R10
100kΩ trimmer
C4
10nF/100V Ceramic capacitor
R12,R13,R14
1Ω 1% resistor
C5
10µF/16V Capacitor
R16
1MΩ resistor
C6
33nF Capacitor
R17
20kΩ resistor
C7
1nF Capacitor
R18
4.7kΩ resistor
C8
820pF Capacitor
R19
5.6kΩ resistor
C9
10nF Capacitor
R20
2.2kΩ resistor
C10
220nF Capacitor
R21
1.8kΩ resistor
C11
68nF Capacitor
R22
5kΩ trimmer
C12
100nF Capacitor
S1
quad switch
D1, D2
1N4148 Diode
U1
L6235N
D3
BZX79C5V1 5.1V Zener Diode
U2
LM358
The Evaluation Board provides external connectors for the supply voltage, an external 5 V reference for the logic
inputs, three outputs for the motor and a 34-pin connector to control the main functions of the board through an
external µC board.
Running the evaluation board in stand-alone mode, instead, four switches (S1) allow enabling the device, setting the direction of the rotation, braking the motor, choosing to run in torque or speed mode. R17 and R22 set
35/39
AN1625 APPLICATION NOTE
the reference voltage provided to the Vref pin of the L6235 (in torque mode) or to the error amplifier, U2 (in
speed mode); R20, C11 make up a low-pass filters to provide an external reference voltage by a PWM output
of a µC (see also the Reference Voltage section). R10, C8 are used to set the off-time and R11, C9 set the
duration of the TACHO output pulses.
The 5V voltage for logic inputs and for the reference voltage is obtained from R1, D3. For supply voltages greater than 20V, R1 must be replaced with a higher value resistor. The jumper JP1 and JP2 allow disconnecting the
internal zener diode network, in case the 5V voltage is provided through pin 11 of CN5 (for example an external
µC board can provide 5 V to the evaluation board). Also CN2 connector can be used to provide an external 5 V
voltage to the board. CN2, or pin 1 of CN5, can also be used to provide a 5 V voltage to external circuits. In this
case only a small current can be drawn form the board, depending on the supply voltage and R1 value. Figure
41, Figure 42, Figure 43 show the component placement and the two layers layout of the L6235N Evaluation
Board.
A large GND area has been used, to guarantee minimal noise and good power dissipation for the device.
Figure 39. EVAL6235N.
R22
1st switch
of S1
JP1
R1
R2
R6
C6
JP2
4.2.1 Important Notes
JP1 : close for use with PractiSPIN ST7 board
JP2 : close for use with PractiSPIN ST7 board
C6 : recommended change to 5.6 nF for safe Overcurrent protection
R2 : recommended change to 100 k for safe Overcurrent protection
R6 : recommended change to 100 k (and remove R2) if EN pin is driven from the CN4 connector (for example
with PractiSPIN ST7 board) for safe Overcurrent protection
R22 : set the maximum current obtainable through PractiSPIN (see PractiSPIN documentation)
R1 : recommended change to adequate value (depending on supply voltage) to obtain 5V across D3
S1 : move first switch in TRQ position for use with PractiSPIN ST7 board
36/39
AN1625 APPLICATION NOTE
H2
H3
FRW/REW
ENABLE
1
2
3
1
2
2
1
H1
BRAKE
CN3
CN1
DIAG
CN2
RC/INH
Figure 40. L6235N Evaluation Board Electrical schematic.
U2B
LM358
7
P2.4
8
4
5
2
4
6
8
10
12
14
16
18
20
22
24
26
28
30
32
34
+
+5V
-
C5
6
P2.5
P2.6
P2.1
P2.7
JP2
R1
P2.2
INT3
INT2
JP1
TINA0 P2.0
TP
D3
CN4
+5V
P3.1
PullUp
VCCREF
TP
OCMPA1 P4.2
ADC_REF
A0IN6 P7.6
1
3
5
7
9
11
13
15
17
19
21
23
25
27
29
31
33
VCCREF
TP
H1
C3
SENSE
VREF
+5V
16
5
21
R21
U1
L6235
OUT3
U2A
LM358
1
10
SENSE2
R14
L6235 N
R11
C4
R5
22
VCP
D1
9
RCPULSE
R18
C9
8
TACHO
CW
VSA
D2
C12
CW
SENSE1
20
+
R16
R12
3
VSB
2
17
+5V
8
4
C10
R13
3
OUT2
OUT1
R19
R10
15
C11
R15
Pullup
4
VBOOT
RCOFF
CW
C2
C8
19
PullUp
13
GND
R22
R17
VREF
18
GND
C1
7
C7
GND
6
R20
H1H2H3
2
H3
H2
H1
BRAKE
PullUp
FWR/REW
DIAG
ENABLE
GND
VREF
FW
10
24
23
1
11
8
7
R4
R9
11
6
12
13
1
2
3
4
5
REV
5
BRAKE
EN
14
R8
R3
CN5
HALL CON
9
14
12
+5V
S1
R7
PullUp
4
3
R2
C6
TRQ
16
2
R6
DIAG
1
FRW/REV
BRAKE
SPEED
ENABLE
15
H1
H2
H3
37/39
AN1625 APPLICATION NOTE
Figure 41. L6235 Evaluation Board Component placement.
Figure 42. L6235 Evaluation Board Top Layer Layout.
Signal GND
Power GND
Bulk Capacitor
38/39
AN1625 APPLICATION NOTE
Figure 43. L6235 Evaluation Board Bottom Layer Layout.
SENSE path
5
REFERENCES
1
D. Arrigo, A. Genova, T. Hopkins, V. Marano, A. Novelli, "A New Fully Integrated Stepper Motor
Driver IC", Proceedings of PCIM 2001, September 2001, Intertech Communication.
2
T. Hopkins, "Controlling Voltage Transients in Full Bridge Driver Applications" (AN280).
3
P. Casati and C. Cognetti, "A New High Power IC Surface Mount Package Family" (AN668).
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences
of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted
by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject
to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not
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