dm00024579

AN3358
Application note
Turbo2 600 V diodes:
optimized solutions for PFC and other applications
Introduction
In a switched mode power supply, there are a great number of electronic functions where
600 V ultrafast diodes are used. Each diode has a specific function. In one application a
parameter can be critical but secondary in another.
A rectifier manufacturer who wants to propose an optimized solution for each function needs
to develop several families with different trade-offs (mainly between the forward voltage VF
and reverse recovery charge Qrr).
STMicroelectronics’ Turbo2 600 V ultrafast diodes offer three different families in order to
offer an optimal solution for each application.
After some general information about this new technology, a discussion of the PFC
application, working in continuous mode, transition mode and fixed-off-time, is presented. In
the case of continuous mode operation, hard switching and soft switching conditions are
considered. Some other conventional functions are also touched upon.
September 2011
Doc ID 018581 Rev 1
1/18
www.st.com
Contents
AN3358
Contents
1
2
General information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.1
Technology information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.2
VF, Qrr trade-off for the three families . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.3
Platinum doping and low leakage current . . . . . . . . . . . . . . . . . . . . . . . . . 3
Main applications of 600 V ultrafast diodes . . . . . . . . . . . . . . . . . . . . . . 5
2.1
2.2
Power factor corrector applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.1.1
Boost diode in PFC working in continuous mode . . . . . . . . . . . . . . . . . . 5
2.1.2
Boost diode in PFC working in transition mode . . . . . . . . . . . . . . . . . . . 10
2.1.3
Boost diode working in fixed-off-time (FOT) PFC . . . . . . . . . . . . . . . . . 12
Other applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
3
Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
4
References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
5
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
2/18
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AN3358
General information
1
General information
1.1
Technology information
The Turbo2 families are manufactured using simple rules to insure high quality and
reliability. These diodes are planar structures on epitaxial layers. The wafers are thus
subjected to reduced mechanical stress for planar diodes compared to mesa ones.
The use of epitaxial layers makes the VF/trr trade-off independent of the wafer thickness, the
contrary of homogenous diodes. These properties make the manufacturing of large
diameter wafers possible. So the wafers benefit from state-of-the-art technology on recent
equipment.
Epitaxial diodes, which present good drift area thickness, are particularly suitable for diodes
up to 600 V and exhibit a significantly superior VF/trr trade-off. The lifetime control of the
carriers for the Turbo2 diodes is obtained through platinum (Pt) doping. Pt doping is required
for high junction temperature applications because it results in low reverse current at
elevated temperature and, in this way, presents a low thermal runaway risk.
1.2
VF, Qrr trade-off for the three families
The three families are: STTHxxR06 (R stands for rapid with low Qrr), STTHxx06 (medium VF
and Qrr), and STTHxxL06 (Low forward voltage).
Figure 1 shows where a trade-off occurs in three operational areas. A technology using gold
doping is also shown.
Figure 1.
VF - Qrr trade-off for an 8 A diode
1.7
VF (V)
Typical values
R Family
1.2
IF = 8 A
VR = 400 V
dIF/dt = 200 A/µs
Tj = 125 °C
Gold doping
Medium Family
Platinum doping
L Family
Qrr (nC)
0.7
0
1.3
200
400
600
800
1000
1200
1400
1600
Platinum doping and low leakage current
Figure 2 shows the trade-off between leakage current IR and Qrr in several operational
areas. The faster the diode, the higher the IR is. This rule is true for both gold and platinum
doping. For the same Qrr, IR is approximately 100 times lower with platinum doping. The
corresponding “R” family with gold doping would have a high maximum leakage current
(18 mA at 125 °C and 400 V). As shown later in this Application note, with such a leakage
current thermal instability can be reached for operating junction temperatures higher than
125 °C in a conventional application.
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General information
AN3358
It will also be shown that IR is also a critical parameter for diodes in axial and SMD
packages.
Figure 2.
IR - Qrr trade-off in several operational areas for an 8 A diode
100000
IRmax (µA)
IF = 8 A
VR = 400 V
dIF/dt = 200 A/µs
Tj = 125 °C
10000
1000
R Family
Gold doping
Medium Family
100
L Family
10
Platinum doping
Qrr typ (nC)
1
0
4/18
500
1000
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1500
2000
AN3358
2
Main applications of 600 V ultrafast diodes
Main applications of 600 V ultrafast diodes
This section discusses the trade-offs in a common application. Boost power factor corrector
(PFC) will be widely covered since it is a major application. A typical PFC circuit is shown in
Figure 3.
2.1
Power factor corrector applications
Figure 3.
Boost power factor corrector circuit
Vmains
L
Dboost
Vout
IRM
Vgate
2.1.1
Boost diode in PFC working in continuous mode
Hard switching conditions
PFC applications are mainly designed in continuous mode when the power is greater than
200 W.
In such an application, it is well known that the greatest losses due to the diode are the
switching losses in the transistor (Pontr) when it turns on. The reverse recovery current (IRM)
of the boost diode flows into the MOSFET (Figure 3). Consequently, the best choice in most
cases is the “R” family.
Switching losses due to IRM depend mainly on two parameters: the operating junction
temperature Tj and the mains voltage Vmains.
Figure 4 and Figure 5 show that the switching losses for STTH8R06 quickly increase when
Tj increases and when Vmains decreases. These curves are drawn with a software tool
realized by these authors.
If the PFC only works on 240 V mains, with a low operating junction temperature, switching
losses will be less critical and the best trade-off could be the intermediate trade-off:
STTHxx06.
However, most PFCs are designed to work in a wide mains voltage range (85 V-264 V) with
an operating junction temperature (in the worst case) close to 100 °C. The “R” family will be
the family usually recommended.
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Main applications of 600 V ultrafast diodes
Figure 4.
AN3358
Switching losses versus Tj at turn off of the diode
14
Poff diode + Pontr due to the diode (W)
STTH8R06D
12
10
Vmains = 90 V
dI/dt = 400 A/µs
L = 0.5 mH
Fsw = 100 kHz
VOUT = 400 V
POUT = 400 W
8
6
4
2
Tj (°C)
0
0
Figure 5.
25
50
75
100
125
150
175
Switching losses versus Vmains at turn off of the diode
12
Poff diode + Pontr due to the diode (W)
STTH8R06D
10
8
Tj = 100 °C
dI/dt = 400 A/µs
L = 0.5 mH
Fsw = 100 kHz
VOUT = 400 V
POUT = 400 W
6
4
2
Vmains (V)
0
0
6/18
50
100
150
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200
250
300
AN3358
Main applications of 600 V ultrafast diodes
Tjmax before thermal runaway
The maximum junction temperature Tjmax before thermal runaway can be calculated using
Equation 1, Equation 2 then Equation 3.
Equation 1
δ=1-
2 Vmains peak
π VOUT
Equation 2
IR(VOUT, Tjmax) =
1
VOUT · δ · c Rth(j-a)
Equation 3
Tjmax = 125 +
1
c
· loge
(
IR(VOUT, Tjmax)
IRmax(VOUT, 125 °C)
(
Where:
●
δ is the average duty cycle of the blocking time of the diode given by Equation 3.
●
VOUT is the output voltage.
●
c is a constant with units of °C-1. Each diode has its own “c” coefficient depending on
the technology of the diode and the reverse voltage VR applied. It can be determined
from Equation 3 for two values of leakage current corresponding to application reverse
voltage Vout, for example: IR(Vout,100 °C) and IR(Vout,125 °C).
●
Rth(j-a) is the thermal resistance between junction and ambient (heatsink + diode).
With the following conditions:
VOUT = 400 V, c ≈ 0.055 °C-1 (for the “R” family)
Vmains = 85 V, δ = 0.8, Rth(j-a) = 10 °C/W
Figure 2 gives IRmax(400 V, 125 °C) = 215 µA for an 8 A “R” family diode and 17 mA for the
equivalent diode in gold doping.
Equation 2 and Equation 3 give Tjmax = 184 °C for Turbo2 and 104 °C for the equivalent
diode in gold doping.
Soft switching condition
Designers can use a number of techniques to turn on the MOSFET in soft switching
conditions and reduce the switching losses due to IRM.
Figure 6 and Figure 7 show two solutions, widely used with the associated waveforms
during switching time. In the non-dissipative circuit Figure 6, the smaller transistor T2 turns
on before the main one T1. The dl/dt when Dboost turns off is controlled by Lr (dl/dt = Vout/Lr),
and T1 turns on at zero current. Consequently, the switching-on losses will be close to zero.
With this circuit, the reverse recovery current of the boost diode is less critical. The best
choice, following the application conditions (switching frequency, Lr…) will be “the
intermediate” or the “L” trade-off.
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Main applications of 600 V ultrafast diodes
Figure 6.
AN3358
Non-dissipative soft switching solution
Vmains
Dboost
L
Vout
Lr
Dr
Cr
T1
T2
T1, T2
T1on
I DBoost
I0
t
T 2on
⎛ VOUT ⎞
⎜
L r ⎟⎠
⎝
I Lr
t
I RM
I 0+I RM+I res
I 0+IRM
I Dr
t
VC r
t
t
I T1
t
I RM+I res
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AN3358
Main applications of 600 V ultrafast diodes
The topology shown in Figure 7 is more simple but more dissipative than that in Figure 6.
The waveforms in Figure 7 show the MOSFET turning on at zero current, thus reducing the
switching losses. When the diode turns off, the Lr inductor is charged with the reverse
recovery current of the boost diode. This energy will be dissipated in the resistor.
The higher IRM is the higher the losses in the resistor are. In this application IRM is more
critical than in the previous one. The best choice for the boost diode trade-off will be “R” or
medium family depending on the application conditions.
Figure 7.
Dissipative soft switching solution
Dr
Vmains
L
VRC
DBoost
VOUT
Lr
VDS
T
450 V
20 A
IT
250 V
10 A
VDS
0A
0V
-250 V
180.0
60 V
IDBoost
IRM
Qrr
180.1
180.2
180.3
180.4
t (µs)
180.5
180.6
180.7
VRC
IDr
40 V
-10 A
180.8
20 A
10 A
IDBoost
20 V
0A
0V
180.0
ILr
180.1
180.2
180.3
t (µs)
180.4
180.5
180.6
180.7
-8 A
180.8
Another very interesting alternative soft switching solution is described in the application
note AN3276, “ST solution for efficiency improvement in PFC applications, back current
circuit (BC2)”. AN3276 presents a patented soft switching circuit from STMicroelectronics
offering performance similar to that of SiC Schottky diodes.
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Main applications of 600 V ultrafast diodes
2.1.2
AN3358
Boost diode in PFC working in transition mode
The transition mode (TM) is widely used for low power PFC (<200 W). The particularity of
this control mode working between continuous and transition mode is a simple control and a
few external components. This control mode results in variable frequency operation and a
constant on time of the MOSFET.
Consequently, the current flowing through the Boost inductor is triangular (Figure 8). It
increases through the MOSFET following the slope defined by Vmains/L, and decreases
through the diode following a low dl/dt given in Equation 4.
Equation 4
dI
dt
=
Vmains - VOUT
L
In this case dI/dt may have a value up to 0, the necessary condition for the next cycle.
Figure 8.
Inductor current waveform and MOSFET timing
IPK = 2 2x POUT V
mains
Average
input current
Inductor
current
Vmains
L
Vmains-Vout
L
T=
N
1
=
2 · Fmains FSW
On
MOSFET
Off
ton fixed
TSW variable
The ZCD circuit (zero current detection) turns on the MOSFET at zero current, avoiding high
switching losses in the MOSFET due to the recovery charge of the diode.
Unlike the continuous mode, the Qrr of the diode it is not the key parameter any more. In the
transition mode, the main losses of the diode are due to the forward voltage. It is then
possible to optimize the VF parameter to the detriment of Qrr, due to the low dl/dtoff of the
diode (<1 A/µs) fixed by the inductor.
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AN3358
Main applications of 600 V ultrafast diodes
Nevertheless, an accurate study at switch-off of the diode shows that the Qrr parameter
cannot be indefinitely relaxed. Figure 9 highlights this phase when the current of the diode
reaches 0, and shows that this time is composed of 3 phases:
●
Phase 1 [t0,t1]: The diode is open. There is a resonant circuit between the equivalent
capacitance (Cds MOS + Cj diode) and the boost inductance, which has as its initial
condition the IRM of the diode.
●
Phase 2 [t1,t2]: VDS reaches 0 and the body diode of the MOSFET enters in
conduction and the current linearly increases through the VF of the body diode.
●
Phase 3 at t2: The ZCD circuit turns the MOSFET on and the current continues to
linearly rise through the RDS(on).
Figure 9.
Switch-off comparison between STTH1L06 and a slower diode
t 0 t1 t2
IRM
Vds
V
ds
Slower diode
Vgrille
IDiode
IMos
STTH1L06
It can be observed that the dead time (t0,t2) increases with the IRM of the diode. This time a
negative current flows through the power MOSFET and is the source of additional losses.
This duration depends on the slope (versus Vmains, L) and also on the IRM of the diode (the
initial condition of phase 1). During this time there is no power transferred to the load. In this
way, with a very slow diode, the sum of the losses due to high IRM cannot be negligible
compared to these of the conduction losses. Therefore, there is a limit for Qrr. This limit
appears for the full range PFC at 110 V. In this condition the current in the power MOSFET
takes more time to reach 0 (maximum dead time).
The maximum Qrr of the “L” family has been optimized taking these considerations into
account.
According to the application conditions (Pout, Vmains, dI/dtmax, Fsw, Tj), the medium trade-off
could be also considered. The optimum choice between low forward voltage trade-off
(STTHxxL06) and the medium trade-off (STTHxx06) could be determined by efficiency
measurement.
In transition mode a diode with a small current rating is used. It is generally a small package
(axial or SMD packages) with high thermal resistances. Consequently, the junction
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Main applications of 600 V ultrafast diodes
AN3358
temperature of the diode, which is mainly fixed by the conduction losses, can be high.
Equation 2 in Section 2.1 shows that the thermal resistance is a critical parameter for the
thermal runaway limit. Table 1 compares the thermal runaway limit between Turbo2 and a
gold-doped diode working in a transition mode PFC in the following conditions:
Rth(j-a) = 75 °C/W, c ≈ 0.072 °C-1, VOUT = 400 V, Vmains = 85 V, δ = 0.808
Table 1.
Tjmax comparison between Turbo2 and gold doping diode
STTH3L06
Gold Doping
IRmax 125 °C, 400 V
15 µA
1.5 mA
Tjmax before thermal runaway
limit is reached.
176 °C
112 °C
This comparison shows that gold-doped diodes are limited in high temperature. There is no
thermal runaway risk when Turbo2 uses platinum doping. For all these reasons, in most
cases, the “L” family is recommended for the PFC application working in transition mode.
2.1.3
Boost diode working in fixed-off-time (FOT) PFC
In this third PFC operating mode, instead of maintaining the on-time fixed, such as TM PFC,
the Toff is kept constant and the Ton is free to be changed in order to modulate the power
drained from the source according to the load.
This modulation method, is described in the Application note AN1792, “Design of fixed-offtime controller PFC pre-regulators with L6562”.
As shown in Figure 10 in FOT mode, the PFC works in DCM and CCM modes along the line
semi period.
Figure 10. Inductor, switch and diode currents in a CCM FOT-controlled PFC stage
CCM
Inductor current
peak envelope
ILpk
DCM
DCM
Low frequency
inductor current
Switch current
Diode current
ON
OFF
Switch
θt
12/18
OFF
OFF
Doc ID 018581 Rev 1
π−θt
AN3358
Main applications of 600 V ultrafast diodes
In this operating mode, according to the application conditions the optimal diode will be the
medium trade-off (VF/QRR) or the rapid trade-off (“R” family). The designer should make
some measurements of efficiency to confirm the good trade-off diode in its application.
2.2
Other applications
There are numerous other electronic functions, where 600 V ultrafast diodes are used. For
example, rectification, demagnetization, snubber, bootstrapping, clamping, or East-West
correction in a horizontal deflection circuit for TV or monitor (Figure 11).
Figure 11. Traditional applications of 600 V ultrafast diodes
Clamping diode
Demagnetization diode
Snubber diode
Bootstrap diode
Modulator diode in horizontal deflection circuit
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Main applications of 600 V ultrafast diodes
AN3358
It is not possible in this document to analyze each function in detail. We will focus on the
clamping function used in flyback converters. The function of the clamping circuit is to
protect the MOSFET against the overvoltage due to the energy in the leakage inductance of
the transformer. The associated waveforms are represented in Figure 12.
Figure 12. 600 V ultrafast diode waveforms in clamping function
Vmains
VLf
m
or
DR
VC
VIN
VOUT
VDCL
DCL
VDS
Breakdown voltage of the MOS transistor
VDS
VDS=V IN+VC+V FP
VL f
V IN+(VOUT/m)
VIN
VCEsat
ID CL
IDR
VDCL
ID CL
IDR
Qrr
V FP
V Rmax=V IN+VC+S pike
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AN3358
Main applications of 600 V ultrafast diodes
When the MOSFET turns off, the inductive circuit opens and an overvoltage VLf appears in
addition to the voltage across the primary inductor VOUT/m. The effect of this overvoltage
turns on the clamping diode. Thus, the drain voltage is equal to VDS = VIN + VC +VFP.
VFP is the peak forward voltage across the 600 V diode. VC is a DC voltage realized either
by an RC circuit in parallel or by a clamping diode such as a Transil™.
The first key parameter of the diode is VFP. VDS has to be lower than the breakdown voltage
of the MOSFET. If VFP is too high the designer may be obliged to choose a higher voltage
MOSFET (for example 800 V instead of 600 V).
To avoid thermal runaway problems a low value of leakage current is necessary as the diode
is normally a 1 A device in an SMD or axial package. A low IR will also contribute to the
reduction of consumption in stand-by mode. The forward voltage is not a critical parameter
because the diode conducts about ten nanoseconds every switching period.
When the clamping voltage is made with a Transil, it is generally better to use an ultrafast
type diode. When an RC solution is used, the capacitance is discharged through the reverse
recovery current of the diode, thus reducing the losses in the resistor.
The Turbo2 technology, which allows low leakage current and low peak forward voltage, is
well suited for this application. The best trade-off with a Transil, will be the “R” or the medium
family. With an RC solution the choice will generally be between the “L” and the “medium”
families.
TM: Transil is a trademark of STMicroelectronics
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Conclusion
3
AN3358
Conclusion
This Application note presents the main applications of the 600 V ultrafast diodes. These
applications are numerous, each requiring a slightly different trade-off among the diode
parameters. In order to propose an optimized solution for each one, three trade-offs are
proposed by STMicroelectronics. There are some general rules to define the right trade-off.
For example, the “R” family for PFC working in continuous mode and hard switching
condition and the “L” family for PFC working in transition mode. However, there are also
applications for which a deeper study will be necessary.
An important benefit of the platinum doping implemented in the Turbo2 technology resides
in the use of the diodes at high junction temperature without thermal runaway risk in normal
prescribed condition of use (<175 °C).
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AN3358
4
References
References
[1] ST Application note AN628, “Designing a high power factor switching preregulator with
the l4981 continuous mode”
[2] PCIM, Nuremburg, 2000 “New solution to optimize diode recovery in PFC boost
converter”, B. Rivet.
[3] ST Application note AN667, “Designing a high power factor switching preregulator with
the l6560 transition mode”
[4] ST Application note AN966, “Enhanced transition mode power factor corrector”
[5] ST Application note AN1792, “Design of fixed-off-time controller PFC preregulator with
the L6562”
[6] ST Application note AN3276, “ST solution for efficiency improvement in PFC
applications, back current circuit (BC2)”
5
Revision history
Table 2.
Document revision history
Date
Revision
14- Sep-2011
1
Changes
First issue
Doc ID 018581 Rev 1
17/18
AN3358
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