LT1370 - 500kHz High Efficiency 6A Switching Regulator

LT1370
500kHz High Efficiency
6A Switching Regulator
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FEATURES
DESCRIPTION
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The LT ®1370 is a monolithic high frequency current mode
switching regulator. It can be operated in all standard
switching configurations including boost, buck, flyback,
forward, inverting and “Cuk.” A 6A high efficiency switch
is included on the die, along with all oscillator, control and
protection circuitry.
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■
■
■
■
■
■
■
Faster Switching with Increased Efficiency
Uses Small Inductors: 4.7µH
All Surface Mount Components
Low Minimum Supply Voltage: 2.7V
Quiescent Current: 4.5mA Typ
Current Limited Power Switch: 6A
Regulates Positive or Negative Outputs
Shutdown Supply Current: 12µA Typ
Easy External Synchronization
Switch Resistance: 0.065Ω Typ
The LT1370 typically consumes only 4.5mA quiescent
current and has higher efficiency than previous parts.
High frequency switching allows for very small inductors
to be used.
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APPLICATIONS
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New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an external logic level source. A logic low on the Shutdown pin
reduces supply current to 12µA. Unique error amplifier
circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation techniques. Nonlinear error amplifier transconductance
reduces output overshoot on start-up or overload recovery. Oscillator frequency shifting protects external components during overload conditions.
Boost Regulators
Laptop Computer Supplies
Multiple Output Flyback Supplies
Inverting Supplies
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
5V to 12V Boost Converter
L1*
VIN
OFF
ON
S/S
VOUT†
12V
VSW
R1
53.6k
1%
FB
+
LT1370
+
C1**
22µF
25V
GND
C2
0.047µF
R3
2k
VC
R2
6.19k
1%
C4**
22µF
25V
×2
*COILTRONICS
UP2-4R7 (4.7µH)
UP4-100 (10µH)
**AVX TPSD226M025R0200
†MAX I
OUT
L1
IOUT
4.7µH 1.8A
10µH 2A
C3
0.0047µF
LT1370 • TA01
VIN = 5V
L = 10µH
90
EFFICIENCY (%)
5V
12V Output Efficiency
92
D1
MBRD835L
88
86
84
82
80
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
LOAD CURRENT (A)
LT1370 • TA02
sn1370 1370fs
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LT1370
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AXI U
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ABSOLUTE
RATI GS
Supply Voltage ....................................................... 30V
Switch Voltage
LT1370 ............................................................... 35V
LT1370HV .......................................................... 42V
S/S, SHDN, SYNC Pin Voltage ................................ 30V
Feedback Pin Voltage (Transient, 10ms) .............. ±10V
Feedback Pin Current ........................................... 10mA
Negative Feedback Pin Voltage
(Transient, 10ms) ............................................. ±10V
Operating Ambient Temperature Range ...... 0°C to 70°C
Operating Junction Temperature Range
Commercial .......................................... 0°C to 125°C
Industrial ......................................... – 40°C to 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER I FOR ATIO
FRONT VIEW
7
6
5
4
3
2
1
TAB
IS
GND
VIN
NFB
VSW
GND
S/S
FB
VC
R PACKAGE
7-LEAD PLASTIC DD
ORDER PART
NUMBER
LT1370CR
LT1370HVCR
LT1370IR
LT1370HVIR
FRONT VIEW
7
6
5
4
3
2
1
TAB
IS
GND
TJMAX = 125°C, θJA = 30°C/W, θJC = 4°C/W
VIN
NFB
VSW
GND
S/S
FB
VC
T7 PACKAGE
7-LEAD TO-220
ORDER PART
NUMBER
LT1370CT7
LT1370HVCT7
LT1370IT7
LT1370HVIT7
TJMAX = 125°C, θJA = 50°C/W, θJC = 4°C/W
WITH PACKAGE SOLDERED TO 0.5 INCH2 COPPER
AREA OVER BACKSIDE GROUND PLANE OR INTERNAL
POWER PLANE. θJA CAN VARY FROM 20°C/W TO
> 40°C/W DEPENDING ON MOUNTING TECHNIQUE
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, TA = 25°C unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
VREF
Reference Voltage
Measured at Feedback Pin
VC = 0.8V
1.230
1.225
1.245
1.245
1.260
1.265
V
V
250
550
900
nA
nA
0.01
0.03
%/V
IFB
Feedback Input Current
●
VFB = VREF
●
VNFR
INFB
gm
Reference Voltage Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
Negative Feedback Reference Voltage
Measured at Negative Feedback Pin
Feedback Pin Open, VC = 0.8V
●
– 2.525
– 2.560
– 2.48
– 2.48
– 2.435
– 2.400
V
V
Negative Feedback Input Current
VNFB = VNFR
●
– 45
– 30
– 15
µA
Negative Feedback Reference Voltage
Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
0.01
0.05
%/V
Error Amplifier Transconductance
∆IC = ±25µA
1500
1900
2300
µmho
µmho
200
350
µA
1400
2400
µA
●
1100
700
120
Error Amplifier Source Current
VFB = VREF – 150mV, VC = 1.5V
●
Error Amplifier Sink Current
VFB = VREF + 150mV, VC = 1.5V
●
sn1370 1370fs
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LT1370
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, TA = 25°C unless otherwise noted.
SYMBOL
AV
f
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Error Amplifier Clamp Voltage
High Clamp, VFB = 1V
1.5
1.8
2.30
V
Low Clamp, VFB = 1.5V
0.2
0.3
0.52
Error Amplifier Voltage Gain
VC Pin Threshold
Duty Cycle = 0%
Switching Frequency
2.7V ≤ VIN ≤ 25V
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ ≤ 0°C (I-Grade)
Maximum Switch Duty Cycle
V/ V
0.9
1.1
1.35
V
●
460
440
400
500
500
550
580
580
kHz
kHz
kHz
●
85
95
35
42
40
Switch Current Limit Blanking Time
130
BV
Output Switch Breakdown Voltage
LT1370
●
●
LT1370HVC, 0°C ≤ TJ ≤ 125°C
LT1370HVI, – 40°C ≤ TJ ≤ 0°C (I-Grade)
VSAT
Output Switch ON Resistance
ISW = 6A
●
ILIM
Switch Current Limit
Duty Cycle = 50%
Duty Cycle = 80% (Note 1)
●
∆IIN
∆ISW
%
300
ns
44
47
V
V
V
0.065
0.11
Ω
8
7
10
A
A
Supply Current Increase During Switch ON Time
22
33
mA/A
Control Voltage to Switch Current
Transconductance
10
6
A/V
●
2.4
2.7
2.7V ≤ VIN ≤ 25V
●
4.5
6
mA
Shutdown Supply Current
2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V
●
12
40
µA
Shutdown Threshold
2.7V ≤ VIN ≤ 25V
●
0.6
1.3
2
V
●
4
12
25
µs
Minimum Input Voltage
IQ
V
500
Supply Current
Shutdown Delay
S/S Input Current
Synchronization Frequency Range
The ● denotes specifications which apply over the full operating
temperature range.
0V ≤ S/S ≤ 5V
V
●
–7
10
µA
●
600
800
kHz
Note 1: For duty cycles (DC) between 45% and 85%, minimum switch
current limit is given by ILIM = 2.65(2.7 – DC).
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LT1370
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TYPICAL PERFORMANCE CHARACTERISTICS
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
8.2
550
SWITCH CURRENT LIMIT (A)
75°C
400
25°C
350
300
0°C
250
200
150
100
2.8
7.8
7.6
7.4
7.2
4
2
3
SWITCH CURRENT (A)
5
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
0
6
20
2.0
18
1.8
SHUTDOWN DELAY (µs)
1.4
12
1.2
10
1.0
SHUTDOWN DELAY
0.8
6
0.6
4
0.4
2
0.2
0
–50 –25
0
SHUTDOWN THRESHOLD (V)
1.6
14
8
0
25 50 75 100 125 150
TEMPERATURE (°C)
3.0
2.5
2.0
1.5
1.0
0.5
0
–50 –25
0
–1
–2
–3
–4
2
3
4
VOLTAGE (V)
5
6
7
LT1370 • G07
25°C
–55°C
200
125°C
100
0
–100
–200
–300
25 50 75 100 125 150
TEMPERATURE (°C)
–0.3
VREF
–0.2
–0.1
FEEDBACK PIN VOLTAGE (V)
Error Amplifier Transconductance
vs Temperature
110
2000
100
1800
90
80
70
60
50
40
30
gm =
∆I (VC)
∆V (FB)
1600
1400
1200
1000
800
600
400
200
20
10
0.1
LT1370 • G06
TRANSCONDUCTANCE (µmho)
SWITCHING FREQUENCY (% OF TYPICAL)
INPUT CURRENT (µA)
0
1
300
Switching Frequency
vs Feedback Pin Voltage
2
0
400
LT1370 • G05
S/S Pin Input Current vs Voltage
–1
Error Amplifier Output Current
vs Feedback Pin Voltage
fSYNC = 700kHz
LT1370 • G04
1
25 50 75 100 125 150
TEMPERATURE (°C)
LT1370 • G03
Minimum Synchronization
Voltage vs Temperature
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
Shutdown Delay and Threshold
vs Temperature
SHUTDOWN THRESHOLD
0
LT1370 • G02
LT1370 • G01
16
2.2
1.8
–50 –25
6.6
1
0
2.4
2.0
6.8
50
2.6
7.0
ERROR AMPLIFIER OUTPUT CURRENT (µA)
SWITCH VOLTAGE (mV)
8.0
125°C
450
3.0
INPUT VOLTAGE (V)
500
0
Minimum Input Voltage
vs Temperature
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
FEEDBACK PIN VOLTAGE (V)
LT1370 • G08
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1370 • G09
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LT1370
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TYPICAL PERFORMANCE CHARACTERISTICS
VC Pin Threshold and High
Clamp Voltage vs Temperature
Feedback Input Current
vs Temperature
2.2
1.8
1.6
1.4
1.2
VC THRESHOLD
0
25 50 75 100 125 150
TEMPERATURE (°C)
700
NEGATIVE FEEDBACK INPUT CURRENT (µA)
VC HIGH CLAMP
FEEDBACK INPUT CURRENT (nA)
VC VOLTAGE (V)
0
800
2.0
1.0
–50 –25
Negative Feedback Input Current
vs Temperature
VFB =VREF
600
500
400
300
200
100
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1370 • G10
LT1370 • G11
VNFB =VNFR
–10
–20
–30
–40
–50
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1370 • G12
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PIN FUNCTIONS
VC: The Compensation pin is used for frequency compensation, current limiting and soft start. It is the output of the
error amplifier and the input of the current comparator.
Loop frequency compensation can be performed with an
RC network connected from the VC pin to ground. See
Applications Information.
S/S: Shutdown and Synchronization Pin. The S/S pin is
logic level compatible. Shutdown is active low and the
shutdown threshold is typically 1.3V. For normal operation, pull the S/S pin high, tie it to VIN or leave it floating. To
synchronize switching, drive the S/S pin between 600kHz
and 800kHz. See Applications Information.
FB: The Feedback pin is used for positive output voltage
sensing and oscillator frequency shifting. It is the inverting input to the error amplifier. The noninverting input of
this amplifier is internally tied to a 1.245V reference.
VIN: Bypass Input Supply Pin with a Low ESR Capacitor,
10µF or More. The regulator goes into undervoltage lockout when VIN drops below 2.5V. Undervoltage lockout
stops switching and pulls the VC pin low.
NFB: The Negative Feedback pin is used for negative
output voltage sensing. It is connected to the inverting
input of the negative feedback amplifier through a 100k
source resistor.
VSW: The Switch pin is the collector of the power switch
and has large currents flowing through it. Keep the traces
to the switching components as short as possible to
minimize radiation and voltage spikes.
GND: Tie all ground pins to a good quality ground plane.
See Applications Information.
sn1370 1370fs
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LT1370
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BLOCK DIAGRAM
VIN
SHUTDOWN
DELAY AND RESET
SW
LOW DROPOUT
2.3V REG
ANTI-SAT
S/S
SYNC
LOGIC
OSC
DRIVER
SWITCH
5:1 FREQUENCY
SHIFT
+
100k
NFB
NFBA
–
COMP
50k
–
FB
+
1.245V
REF
+
EA
IA
VC
GND SENSE
AV ≈ 20
0.005Ω
–
GND
LT1370 • BD
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OPERATION
The LT1370 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage. Referring to the block
diagram, the switch is turned ON at the start of each
oscillator cycle. It is turned OFF when switch current
reaches a predetermined level. Control of output voltage is
obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second, it reduces the 90° phase shift at midfrequencies
in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely varying input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator provides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A
500kHz oscillator is the basic clock for all internal timing.
It turns on the output switch via the logic and driver
circuitry. Special adaptive antisat circuitry detects onset of
saturation in the power switch and adjusts driver current
instantaneously to limit switch saturation. This minimizes
driver dissipation and provides very rapid turn-off of the
switch.
A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
brought out for positive output voltage sensing. The error
amplifier has nonlinear transconductance to reduce output overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases 10 times,
which reduces output overshoot. The feedback input also
invokes oscillator frequency shifting, which helps protect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator frequency is reduced 5:1. Lower switching frequency allows
full control of switch current limit by reducing minimum
switch duty cycle.
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LT1370
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OPERATION
Unique error amplifier circuitry allows the LT1370 to
directly regulate negative output voltages. The negative
feedback amplifier’s 100k source resistor is brought out
for negative output voltage sensing. The NFB pin regulates
at – 2.48V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintains ease of use. Consult LTC Marketing for units
that can regulate down to – 1.25V.
The error signal developed at the amplifier output is
brought out externally. This pin (VC) has three different
functions. It is used for frequency compensation, current
limit adjustment and soft starting. During normal regulator operation this pin sits at a voltage between 1V (low
output current) and 1.9V (high output current). The error
amplifier is a current output (gm) type, so this voltage can
be externally clamped for lowering current limit. Likewise, a capacitor coupled external clamp will provide soft
start. Switch duty cycle goes to zero if the VC pin is pulled
below the control pin threshold, placing the LT1370 in an
idle mode.
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APPLICATIO S I FOR ATIO
VOUT
Positive Output Voltage Setting
The LT1370 develops a 1.245V reference (VREF) from the
FB pin to ground. Output voltage is set by connecting the
FB pin to an output resistor divider (Figure 1). The FB pin
bias current represents a small error and can usually be
ignored for values of R2 up to 7k. The suggested value for
R2 is 6.19k. The NFB pin is normally left open for positive
output applications. Positive fixed voltage versions are
available (consult LTC Marketing).
R1
FB
PIN
R2
Dual Polarity Output Voltage Sensing
Certain applications benefit from sensing both positive
and negative output voltages. One example is the “Dual
Output Flyback Converter with Overvoltage Protection”
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as
described above. When both the FB and NFB pins are used,
R1 = R2
VOUT
–1
1.245
VREF
LT1370 • F01
Figure 1. Positive Output Resistor Divider
Negative Output Voltage Setting
The LT1370 develops a – 2.48V reference (VNFR) from the
NFB pin to ground. Output voltage is set by connecting the
NFB pin to an output resistor divider (Figure 2). The
–30µA NFB pin bias current (INFB) can cause output
voltage errors and should not be ignored. This has been
accounted for in the formula in Figure 2. The suggested
value for R2 is 2.49k. The FB pin is normally left open for
negative output applications.
( )
( )
VOUT = VREF 1 + R1
R2
–VOUT
INFB
( )
R1
–VOUT = VNFB 1 + R1 + INFB (R1)
R2
R2
R1 =
NFB
PIN
VNFR
⏐VOUT⏐– 2.48
( )(
2.48 + 30 • 10– 6
R2
)
LT1370 • F02
Figure 2. Negative Output Resistor Divider
the LT1370 acts to prevent either output from going
beyond its set output voltage. For example, in this application if the positive output were more heavily loaded than
the negative, the negative output would be greater and
would regulate at the desired set-point voltage. The positive output would sag slightly below its set-point voltage.
This technique prevents either output from going unregulated high at no load.
sn1370 1370fs
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LT1370
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APPLICATIO S I FOR ATIO
Shutdown and Synchronization
The device has a dual function S/S pin which is used for
both shutdown and synchronization. This pin is logic level
compatible and can be pulled high, tied to VIN or left
floating for normal operation. A logic low on the S/S pin
activates shutdown, reducing the part’s supply current to
12µA. Typical synchronization range is from 1.05 to 1.8
times the part’s natural switching frequency, but is only
guaranteed between 600kHz and 800kHz. A 12µs resetable
shutdown delay network guarantees the part will not go
into shutdown while receiving a synchronization signal.
Caution should be used when synchronizing above 700kHz
because at higher sync frequencies the amplitude of the
internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic
switching only occurs when the duty cycle of the switch
is above 50%. Higher inductor values will tend to eliminate this problem.
Thermal Considerations
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause excessive die temperatures. Typical thermal resistance is
30°C/W for the R package and 50°C/W for the T7 package
but these numbers will vary depending on the mounting
techniques (copper area, airflow, etc.). Heat is transferred
from the package via the tab.
Average supply current (including driver current) is:
IIN = 4.5mA + DC(ISW/45)
ISW = Switch current
DC = Switch duty cycle
Switch power dissipation is given by:
PSW = (ISW)2(RSW)(DC)
RSW = Output switch ON resistance
Total power dissipation of the die is the sum of supply
current times supply voltage, plus switch power:
PD(TOTAL) = (IIN)(VIN) + PSW
Surface mount heat sinks are available which can lower
package thermal resistance by two or three times. One
manufacturer, Wakefield Engineering, offers surface mount
heat sinks for the R package and can be reached at (617)
245-5900 or at www.wakefield.com.
Choosing the Inductor
For most applications the inductor will fall in the range of
2.2µH to 22µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch, which has a 6A limit. Higher values also
reduce input ripple voltage and reduce core loss.
When choosing an inductor you need to consider maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault
current in the inductor, saturation and, of course, cost.
The following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.
1. Assume that the average inductor current for a boost
converter is equal to load current times VOUT / VIN and
decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 3A, for instance, a
3A inductor may not survive a continuous 6A overload
condition. Also be aware that boost converters are not
short-circuit protected and that, under output short
conditions, inductor current is limited only by the
available current of the input supply.
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving
because they saturate softly, whereas ferrite cores
saturate abruptly and other core materials fall in
between. The following formula assumes continuous
mode operation but it errs only slightly on the high side
for discontinuous mode, so it can be used for all
conditions.
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LT1370
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APPLICATIO S I FOR ATIO
) )
V
V (V
–V )
IPEAK = (IOUT) OUT + IN OUT IN
VIN
2(f)(L)(VOUT)
VIN = Minimum input voltage
f = 500kHz switching frequency
3. Decide if the design can tolerate an “open” core geometry, like a rod or barrel, which has high magnetic field
radiation, or whether it needs a closed core, like a
toroid, to prevent EMI problems. One would not want an
open core next to a magnetic storage media, for
instance! This is a tough decision because the rods or
barrels are temptingly cheap and small and there are no
helpful guidelines to calculate when the magnetic field
radiation will be a problem.
4. Start shopping for an inductor that meets the
requirements of core shape, peak current (to avoid
saturation), average current (to limit heating) and fault
current. If the inductor gets too hot, wire insulation will
melt and cause turn-to-turn shorts. Keep in mind that
all good things like high efficiency, low profile and high
temperature operation will increase cost, sometimes
dramatically.
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the LTC Applications Department if
you feel uncertain about the final choice. They have
experience with a wide range of inductor types and can
tell you about the latest developments in low profile,
surface mounting, etc.
Output Capacitor
The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. At 500kHz any polarized capacitor
is essentially resistive. To get low ESR takes volume, so
physically smaller capacitors have high ESR. The ESR
range needed for typical LT1370 applications is 0.025Ω
to 0.2Ω. A typical output capacitor is an AVX type TPS,
22µF at 25V (two each), with a guaranteed ESR less than
0.2Ω. This is a “D” size surface mount solid tantalum
capacitor. TPS capacitors are specially constructed and
tested for low ESR, so they give the lowest ESR for a given
volume. To further reduce ESR, multiple output capacitors can be used in parallel. The value in microfarads is
not particularly critical, and values from 22µF to greater
than 500µF work well, but you cannot cheat mother
nature on ESR. If you find a tiny 22µF solid tantalum
capacitor, it will have high ESR and output ripple voltage
will be terrible. Table 1 shows some typical solid tantalum
surface mount capacitors.
Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current
E CASE SIZE
ESR (MAX Ω)
RIPPLE CURRENT (A)
0.1 to 0.3
0.7 to 0.9
0.7 to 1.1
0.4
0.1 to 0.3
0.9 to 2.0
0.7 to 1.1
0.36 to 0.24
0.2 (Typ)
1.8 to 3.0
0.5 (Typ)
0.22 to 0.17
2.5 to 10
0.16 to 0.08
AVX TPS, Sprague 593D
AVX TAJ
D CASE SIZE
AVX TPS, Sprague 593D
AVX TAJ
C CASE SIZE
AVX TPS
AVX TAJ
B CASE SIZE
AVX TAJ
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and AVX type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the output capacitor. Solid
tantalum capacitors fail during very high turn-on surges,
which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead-shorted, do not harm the capacitors.
Single inductor boost regulators have large RMS ripple
current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:
Output Capacitor Ripple Current (RMS)
DC
IRIPPLE (RMS) = IOUT 1 – DC
= IOUT
VOUT – VIN
VIN
DC = Switch duty cycle
sn1370 1370fs
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LT1370
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APPLICATIO S I FOR ATIO
Input Capacitors
Output Diode
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular and
does not contain large squarewave currents as is found in
the output capacitor. Capacitors in the range of 10µF to
100µF with an ESR of 0.1Ω or less work well up to full 6A
switch current. Higher ESR capacitors may be acceptable
at low switch currents. Input capacitor ripple current for a
boost converter is :
The suggested output diode (D1) is a Motorola MBRD835L.
It is rated at 8A average forward current and 35V reverse
voltage. Typical forward voltage is 0.4V at 3A. The diode
conducts current only during switch OFF time. Peak reverse voltage for boost converters is equal to regulator
output voltage. Average forward current in normal operation is equal to output current.
Frequency Compensation
IRIPPLE =
0.3(VIN)(VOUT – VIN)
(f)(L)(VOUT)
f = 500kHz switching frequency
The input capacitor can see a very high surge current when
a battery or high capacitance source is connected “live”
and solid tantalum capacitors can fail under this condition.
Several manufacturers have developed tantalum capacitors specially tested for surge capability (AVX TPS series,
for instance) but even these units may fail if the input
voltage approaches the maximum voltage rating of the
capacitor during a high surge. AVX recommends derating
capacitor voltage by 2:1 for high surge applications.
Ceramic, OS-CON and aluminum electrolytic capacitors
may also be used and have a high tolerance to turn-on
surges.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic
capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
high ripple current ratings and tolerance of turn-on surges.
Loop frequency compensation is performed on the output
of the error amplifier (VC pin) with a series RC network.
The main pole is formed by the series capacitor and the
output impedance (≈500kΩ) of the error amplifier. The
pole falls in the range of 2Hz to 20Hz. The series resistor
creates a “zero” at 1kHz to 5kHz, which improves loop
stability and transient response. A second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency
ripple on the VC pin. VC pin ripple is caused by output
voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor,
VC pin ripple is:
VC Pin Ripple =
1.245(VRIPPLE)(gm)(RC)
(VOUT)
VRIPPLE = Output ripple (VP–P)
gm = Error amplifier transconductance
(≈1500µmho)
RC = Series resistor on VC pin
VOUT = DC output voltage
To prevent irregular switching, VC pin ripple should be
kept below 50mVP–P. Worst-case VC pin ripple occurs at
maximum output load current and will also be increased if
poor quality (high ESR) output capacitors are used. The
addition of a 0.0047µF capacitor on the VC pin reduces
switching frequency ripple to only a few millivolts. A low
value for RC will also reduce VC pin ripple, but loop phase
margin may be inadequate.
sn1370 1370fs
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APPLICATIO S I FOR ATIO
Layout Considerations
For maximum efficiency, LT1370 switch rise and fall times
are made as short as possible. To prevent radiation and
high frequency resonance problems, proper layout of the
components connected to the switch node is essential. B
field (magnetic) radiation is minimized by keeping output
diode, switch pin and output bypass capacitor leads as
short as possible. Figure 3 shows recommended positions for these components. E field radiation is kept low by
minimizing the length and area of all traces connected to
the switch pin. A ground plane should always be used
under the switcher circuitry to prevent interplane
coupling.
The high speed switching current path is shown schematically in Figure 4. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.
More Help
For more detailed information on switching regulator
circuits, please see Application Note 19. Linear
Technology also offers a computer software program,
SwitcherCADTM, to assist in designing switching converters. In addition, our Applications Department is always
ready to lend a helping hand.
SwitcherCAD is a trademark of Linear Technology Corporation.
FB
VC
GND NFB
S/S VSW VIN
C
D
C
KEEP PATH FROM
VSW, OUTPUT DIODE,
OUTPUT CAPACITORS
AND GROUND RETURN
AS SHORT AS POSSIBLE
LT1370 • F03
Figure 3. Layout Considerations— R Package
L1
SWITCH
NODE
VOUT
VIN
HIGH
FREQUENCY
CIRCULATING
PATH
LOAD
LT1370 • F04
Figure 4
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11
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TYPICAL APPLICATIONS N
Positive-to-Negative Converter with Direct Feedback
VIN
2.7V TO 13V
+
T1*
2
D2
P6KE-15A
D3
1N4148 1 •
C1
100µF
VIN
ON
OFF
VSW
S/S
4
+
•
C4
100µF
×2
R2
2.49k
1%
D1
MBRD835L
LT1370
NFB
VC
C3
0.0047µF
R3
2.49k
1%
GND
C2
0.047µF
R1
2k
–VOUT†
–5V
3
*BH ELECTRONICS 501-0726
†MAX I
OUT
IOUT VIN
1.75A 3V
2.25A 5V
3A 9V
LT1370 • TA03
Dual Output Flyback Converter with Overvoltage Protection
R2
6.19k
1%
R1
68.1k
1%
VIN
2.7V TO 10V
+
OFF
C1
22µF
ON
VIN
VSW
FB
S/S
MBRS360T3
T1*
2, 3
7
+
P6KE-20A •
1N4148
8, 9
•
LT1370
NFB
VC
C3
0.0047µF
•4
10
GND
C2
0.047µF
R3
2k
*DALE LPE-5047-100MB
+
1
MBRS360T3
VOUT
15V
C4
47µF
C5
47µF
–VOUT
–15V
R4
12.1k
1%
R5
2.49k
1%
LT1370 • TA04
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12
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Two Li-Ion Cells to 5V SEPIC Converter**
VIN
4V TO 9V
L1A*
6.8µH
VIN
ON
OFF
•
VSW
S/S
C1
33µF
20V
FB
GND
•
VC
C3
100µF
10V
×2
+
L1B*
6.8µH
R1
2k
C4
0.047µF
VOUT†
5V
R2
18.7k
1%
C2
4.7µF
LT1370
+
D1
MBRD835L
R3
6.19k
1%
C5
0.0047µF
LT1370 • TA05
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E475ZY5U-C304
C3 = AVX TPSD107M010R0100
* BH ELECTRONICS 501-0726
** INPUT VOLTAGE MAY BE GREATER OR
LESS THAN OUTPUT VOLTAGE
†MAX I
OUT
IOUT VIN
2A 4V
2.2A 5V
2.6A 7V
2.8A 9V
Single Li-Ion Cell to 5V
L1*
D1
MBRD835L
VOUT†
5V
VSW
R1
18.7k
1%
FB
+
VIN
OFF
ON
S/S
LT1370
+
SINGLE
Li-Ion
CELL
+
C1**
100µF
10V
GND
VC
R2
6.19k
1%
C2
0.047µF
R3
2k
C4**
100µF
10V
×2
C3
0.0047µF
LT1370 • TA06
*COILCRAFT DO3316P-472
**AVX TPSD107M010R0100
†MAX I
OUT
IOUT
2.5A
3A
3.3A
VIN
2.7V
3.3V
3.6V
sn1370 1370fs
13
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Laser Power Supply
0.01µF
5kV
1800pF
10kV
47k
5W
1800pF
10kV
8
11
L1
1
4
5
HV DIODES
3
2
LASER
+
2.2µF
Q1
0.47µF
150Ω
L2
82µH
MUR405
VIN
12V TO 25V
VSW
10k
VIN
+
Q2
10k
FB
LT1370
2.2µF
VC
GND
0.1µF
VIN
1N4002
(ALL)
190Ω
1%
+
10µF
L1 = COILTRONICS CTX02-11128
L2 = GOWANDA GA40-822K
Q1, Q2 = ZETEX ZTX849
0.47µF = WIMA 3X 0.15µF TYPE MKP-20
HV DIODES = SEMTECH-FM-50
LASER = HUGHES 3121H-P
COILTRONICS (407) 241-7876
LT1370 • TA07
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LT1370
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PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
R Package
7-Lead Plastic DD Pak
(LTC DWG # 05-08-1462)
0.256
(6.502)
0.060
(1.524)
TYP
0.060
(1.524)
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
0.045 – 0.055
(1.143 – 1.397)
15° TYP
0.060
(1.524)
0.183
(4.648)
+0.008
0.004 –0.004
0.059
(1.499)
TYP
0.330 – 0.370
(8.382 – 9.398)
(
+0.203
0.102 –0.102
0.095 – 0.115
(2.413 – 2.921)
0.075
(1.905)
0.300
(7.620)
BOTTOM VIEW OF DD PAK
HATCHED AREA IS SOLDER PLATED
COPPER HEAT SINK
)
0.040 – 0.060
(1.016 – 1.524)
0.026 – 0.036
(0.660 – 0.914)
+0.012
0.143 –0.020
(
+0.305
3.632 –0.508
)
0.013 – 0.023
(0.330 – 0.584)
0.050 ± 0.012
(1.270 ± 0.305)
R (DD7) 0396
T7 Package
7-Lead Plastic TO-220 (Standard)
(LTC DWG # 05-08-1422)
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
0.147 – 0.155
(3.734 – 3.937)
DIA
0.045 – 0.055
(1.143 – 1.397)
0.230 – 0.270
(5.842 – 6.858)
0.460 – 0.500
(11.684 – 12.700)
0.570 – 0.620
(14.478 – 15.748)
0.330 – 0.370
(8.382 – 9.398)
0.620
(15.75)
TYP
0.700 – 0.728
(17.780 – 18.491)
0.152 – 0.202
0.260 – 0.320 (3.860 – 5.130)
(6.604 – 8.128)
0.040 – 0.060
(1.016 – 1.524)
0.095 – 0.115
(2.413 – 2.921)
0.013 – 0.023
(0.330 – 0.584)
0.026 – 0.036
(0.660 – 0.914)
0.135 – 0.165
(3.429 – 4.191)
0.155 – 0.195
(3.937 – 4.953)
T7 (TO-220) (FORMED) 1197
sn1370 1370fs
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT1370
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PART NUMBER
DESCRIPTION
COMMENTS
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Input Voltage May Be Greater or Less Than Battery Voltage
LT1513
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Input Voltage May Be Greater or Less Than Battery Voltage
®
sn1370 1370fs
16
Linear Technology Corporation
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FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com
LT/TP 0198 4K • PRINTED IN THE USA
© LINEAR TECHNOLOGY CORPORATION 1998