AN58

A Product Line of
Diodes Incorporated
AN58
Designing with References - Shunt regulation
Peter Abiodun A. Bode, Snr. Applications Engineer, Diodes Incorporated
Introduction
This application note introduces shunt regulators, sometimes called "reference diodes", "3terminal voltage references", or "adjustable voltage references". Using worked examples we
introduce some applications of adjustable (3-terminal) shunt regulators.
Diodes' range of three-terminal adjustable shunt regulators offer temperature stabilised
reference voltage values ranging from 0.6V to 2.5V with an initial tolerance of 0.5% or 1% and
maximum current sinking capability from 18mA to 200mA. Maximum operating voltage ranges
from 18V to 36V.
Adjustable references can be used as replacements for zener diodes in many applications that
require an improved regulation. Also they can also be used in conjunction with other types of
voltage regulators to improve the initial accuracy and regulation. This is discussed elsewhere
particularly in the references provided towards the end of this document.
The information presented in this Applications Note shows typical basic use of references with
calculated examples.
What are they?
A reference in its basic form is a two terminal device that functionally behaves like a zener diode.
The same circuit symbol is used for both of them as shown in Figure 1. A reference is therefore
a shunt regulator and is quite often referred to as such. Note the polarity and direction of current
flow.
The primary difference between a zener diode and a reference lies in the accuracy of the two
devices. Relative to a zener diode, a reference is a precision component. It is more accurate with
a wider dynamic range of operation and better figures of merit all round. A reference can, for
example, work over a current range of 500µA to 50mA (a dynamic range of 100:1) with little or no
practical change in its key properties. This can never be possible with a zener diode which may
typically work with a dynamic range of 2:1 or less and still perform worse than a reference.
The reference is able to do this because, internally, it is in fact an integrated circuit containing
amplifiers and temperature/stability compensating elements. A zener on the other hand is simply
a specially fabricated p-n junction diode.
R
I
+
V
K
S u p p ly
V OUT
A
Figure 1 - Shunt Regulator using a zener diode or a 2-terminal reference diode
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Adjustable or 3-terminal references
Whereas the standard reference comes in a narrow range of fixed voltages with two terminals,
the adjustable kind has a third terminal which allows the user to set any output voltage in a
prescribed range. It is often referred to as a "three-terminal reference". The third pin is
interchangeably labelled "Ref" (reference pin), "Adj" (adjustment pin) or "FB" (feedback pin). The
usual circuit symbol for an adjustable reference is shown in Figure 2 below.
K
Ref
A
Figure 2 - Adjustable or 3-terminal reference
Applications
Basic Shunt Regulator
Vin
I IN
IL
R3
R1 ⎞
⎛
VOUT = VREF ⎜1 +
⎟
⎝ R2 ⎠
Vout
IR
IKA
R1
REF/FB
REF1
R3 =
VREF
C1
VIN − VOUT
I IN
ΔI L (max) ≤ (I KA(max) − I KA(min) )
R2
GND
Figure 3 Basic shunt regulator
The basic shunt regulator circuit using a 2-terminal device has been shown in Figure 1. The low
dynamic resistance of the diode establishes a voltage across the resistor which determines the
resistor current. The same circuit function using a 3-terminal reference is shown in Figure 3 above. In
response to changes in Vin or the load current, IL, REF1 will adjust how much current it sinks or
"shunts" to maintain a voltage equal to VREF across its feedback (FB) or reference (REF) pin. It will be
noticed that the input current, IIN, is the sum of three components, IKA, IL and IR. Usually R1 and R2
are chosen such that IR is much less than IIN to maximise the current that is available to the load.
REF1 has a minimum bias current requirement, IKA(min), below which it is not able to operate. This is
analogous to the "knee" current of a zener diode. Also it has a maximum current sink capability,
IKA(max), beyond which the device will become unreliable or it will fail. These two parameters define
the limit of use of the reference regulator within its maximum programmable voltage such that, at all
times,
ΔI L (max) ≤ (I KA(max) − I KA(min) )
The equation above for VOUT is accurate for most practical purposes. However it does not include
the effect of the input current at the REF pin of the reference device. A more accurate expression
is:
R1 ⎞
⎛
VOUT = VREF ⎜1 +
⎟ + I REF R1
⎝ R2 ⎠
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where IREF is the REF pin input current. IREF is typically 40nA or less for the TLV431. For example, if
R1=100kohm, VOUT is 4mV greater than that given by the simple formula. For the TLV431, this resistor
value is a good choice because it also gives very low power dissipation in the feedback network.
Calculated Example 1
Requirement
Supply Voltage:
Output voltage:
Load current:
15V to 20V
10V ±1%
5mA
Assume the use of TLV431 .
Discussion
The device needs some overhead current whilst delivering 5mA. The TLV431 can work with as little as
100µA shunt current.
Let's assume R1 will be approximately 100k as discussed above. The voltage divider ratio is 10V/1.24V
or roughly 8 to 1. Then R2 will be of the order of 10 or 12k. Choosing R2 = 10k, this makes the current,
IR, equal to 1.24V/10k = 124µA. The minimum current requirement of the TLV431 is IKA(min) = 100µA.
Therefore, allowance for a minimum overhead current of 224µA (i.e. IKA(min) + IR) must be made.
Allowing for a safety margin, this might be rounded up to 250µA. More margin could be allowed but this
could result in unnecessary wasted power which would especially be expensive for battery powered
applications.
This means the circuit, or more specifically, R3, needs to supply 5.25mA under worst case input condition
which is 15V.
Solution
Therefore,
R3 =
VIN (min) − VOUT
I IN
=
15 − 10
0.00525
R3 = 953
= 952.38
to the nearest E48 value.
Check that the maximum current handling capability of the TLV431 is not exceeded:
Maximum current
I max =
VIN (max) − VOUT
R3
=
20 − 10
953
Imax = 10.5mA
- This is less than 15mA as
required.
Lastly, although this could have been computed first, R1 is determined thus,
R1 ⎞
⎛
VOUT = VREF ⎜1 +
⎟
2⎠
R
⎝
Re-arrange to obtain
Equation 1
⎛V
⎞
R1 = R 2⎜⎜ OUT − 1⎟⎟
⎝ VREF
⎠
⎞
⎛ 10
= 10k ⎜
− 1⎟
⎠
⎝ 1.24
= 70.64k
Or
1
to the nearest E192 value
and within 0.057%.
R1 = 70.6k
The same principles apply for all of Diodes' shunt regulators.
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Component Tolerances
Given the expression in Equation 1, the accuracy, vout, of the output voltage can be expressed in
the reduced form as,
Equation 2
⎛
⎛
⎝
R1
⎞⎞
⎠⎠
α VOUT = ± ⎜⎜α TLV 431 + 2 ⋅ α R ⎜
⎟⎟
R1 + R 2 ⎟
⎝
where
(see Appendix for
proof)
α TLV 431 = Manufacturing tolerance of TLV431
α R = Manufacturing tolerance of resistors R1 and R2
(different values but equal fractional tolerance)
This expression applies provided both R1 and R2 are exact values as calculated. If preferred values, other than calculated ones, have to be used the error is given by
Equation 3
⎡
⎛
R1C
⎝ R1C + R 2 C
α VOUT = ± ⎢α TLV 431 + α RD + 2 ⋅ α RP ⎜⎜
⎣
where, as before,
⎞⎤
⎟⎟⎥
⎠⎦
(see Appendix for
proof)
α TLV 431 = Manufacturing tolerance of TLV431
α RP = Manufacturing tolerance of preferred resistors R1 and R2 (see Appendix for
P
P
R1C, R2C are the Calculated values of R1 and R2 respectively.
And the new term,
more explanation)
α RD = Weighted deviation of resistors R1 and R2 from their
calculated values given by,
⎛
R1C
α RD = ⎜⎜
⎝ R1C + R2C
⎞
⎟(α R1D − α R2D )
⎟
⎠
where α R1 and α R 2 are fractional deviations of R1 and R2 from
their calculated values respectively. Both are sign critical.
Equation 4
Since the problem requires an accuracy of ±1%, only the TLV431B (0.5% tolerance) can be used. This
then leaves 0.5% to be shared by the two resistors and any deviation from their standard values. Also
because the calculated R1 is not a preferred value, Equation 3 rather than Equation 2 has to be used.
Hence from Equation 4
⎛ 70.64k
⎞
α RD = ⎜
⎟( −0.057 − 0)
⎝ 70.64k + 10k ⎠
α RD = −0.05%
Transposing Equation 3
⎡⎛ α VOUT − (α TLV 431 + α RD ) ⎞⎛ R1C + R 2C
⎟⎜⎜
2
R1C
⎠⎝
⎣⎝
α RP = ± ⎢⎜
⎞⎤
⎟⎟⎥
⎠⎦
⎡⎛ 1 − (0.5 − 0.05) ⎞⎛ 70.64 + 10 ⎞⎤
= ± ⎢⎜
⎟⎜
⎟⎥
2
⎠⎝ 70.64 ⎠⎦
⎣⎝
α RP = ±0.31%
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Summary
Using a TLV431B, R1 = 70.6k, R2 = 10k (both 0.31% or better) and R3 = 953 will satisfy the requirement.
Current-boosted Shunt Regulator
R3
IL
R4
I R4
IKA
IB
REF1
ZX T P 2039F
I IN
Vin
Vout
IS H IR
Q1
R1 ⎞
⎛
VOUT = VREF ⎜1 +
⎟
R2 ⎠
⎝
R1
VREF
R3 =
I IN = I R 4 + I SH + I L + I R
R2
C1
VIN − VOUT
I IN
GND
Figure 4 - High current shunt regulator
It may at times be required to shunt-regulate more current than a reference device is capable of
providing. Figure 4 shows how this can be done using a transistor Q1 to provide current amplification.
Calculated Example 2
Requirement
Supply Voltage:
Output voltage:
Load current:
15V to 20V
10V ±1%
20mA
Assume the use of TLV431.
Discussion
This problem is similar to Calculated Example 1 above except that more load current is required. The
values calculated for R1 and R2 are still valid. However, R3 will need to be recalculated. IR remains the
same but IKA now consists of two components (IR4 + IB) whilst there is a new term, ISH. The overhead
current can remain the same at 0.25mA making the total current through R3 20.25mA.
Q1 is under the control of the reference and it is important that the reference can turn it fully off if
necessary. R4 is chosen to enable this. As a minimum requirement therefore, R4 will be chosen to ensure
that the voltage drop across it at IKA(min) is insufficient to turn on Q1. This will be the case if VBE at IKA(min)
is kept to no more than 0.2V. In reality it would make more sense to allow the reference to supply as much
of the shunt current as possible before bringing Q1 into service as this will result in the least strain on Q1.
It will also mean that the reference would not have problem turning Q1 off to maximise power delivery
to load.
Solution
R3 =
VIN (min) − VOUT
I IN
=
15 − 10
0.02025
= 246.9
R3 = 240
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To determine R4, we need to know at what nominal current Q1 will be brought into service. Let
this nominal hand-over current be IHC.
I B (max) < I HC < 20mA
V
R 4 = BE
I HC
Let IHC = 10mA. Therefore,
=
0 .6
10mA
= 60
Hence,
At maximum voltage however,
Maximum current
R4 = 62
I IN (max) =
VIN (max) − VOUT
R3
to the nearest E24 value.
=
20 − 10
240
= 41.67mA
This figure shows the wisdom of allowing the TLV431 to supply as much of the load current as
possible before bringing Q1 into service. Had R4 been selected so that REF1 only supplies
enough current for its own bias (i.e. 100µA), Q1 would be required to carry 41.57mA. With an
output voltage of 10V, this would result in a power dissipation of 415.7mW in Q1 which is too
much for the specified device to handle. As it is, with REF1 carrying 10mA, Q1 only needs to carry
the remaining 31.67A. This would result in a power dissipation of 316.7mW which is more within
the capability of Q12.
Determination of IB
I B(max) =
=
I SH (max)
(h FE (min) +1)
31.67mA
101
I B = 313.6 μA
This represents a very small contribution to the 10mA flowing through R4 making the total
current, IKA = 10.31mA.
Accuracy
The same accuracy considerations as in applies here. Neither R3, R4 nor Q1 have any bearing
on accuracy.
2
The ZXTP2039F is a SOT23 transistor with a VCEO rating of -60V, an IC of -1A and can dissipate up to 350mW when
suitably mounted. Refer to datasheet ZXTP2039F.
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Conclusion
The above examples show basic considerations for use of references and amply demonstrate
how easy it is to design with these versatile components.
Recommended further reading
AN59 - Designing with Shunt Regulators – Series Regulation
AN60 - Designing with Shunt Regulators – Fixed Regulators and Opto-Isolation
AN61 - Designing with Shunt Regulators – Extending the operating voltage range
AN62 - Designing with Shunt Regulators – Other Applications
AN63 - Designing with Shunt Regulators – ZXRE060 Low Voltage Regulator
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Appendix - Calculating output error due to components' preferred values and
tolerances
R1 ⎞
⎛
VOUT = VREF ⎜1 +
⎟
R2 ⎠
⎝
αVOUT =
δVOUT =
Transfer function
δVOUT
- definition of fractional
tolerance
VOUT
∂VOUT
∂V
∂V
⋅ δV REF + OUT ⋅ δR1 + OUT ⋅ δR 2
∂V REF
∂R1
∂R 2
⎛
⎝
δVOUT = δV REF ⎜1 +
δVOUT
Therefore,
αVOUT =
Or
αVOUT = αVREF + ⎜
VOUT
=
R1 ⎞
R1 ⎛ δR1 δR 2 ⎞
−
⎟ + V REF ⋅
⎜
⎟
R2 ⎠
R 2 ⎝ R1 R 2 ⎠
δVREF
VREF
⎛ R1 ⎞⎛ δR1 δR 2 ⎞
+⎜
−
⎟⎜
⎟
⎝ R1 + R 2 ⎠⎝ R1 R 2 ⎠
⎛ R1 ⎞
⎟(α R1 − α R 2 )
⎝ R1 + R 2 ⎠
Equation 5
Equation 6
It can be observed that the result consists of two terms namely the fractional tolerance of the
reference itself and the manufacturing fractional tolerance of the two resistors, R1 and R2. It is
entirely logical that these resistors will be of identical types of equal tolerance. Therefore, their
worst case would be obtained when one resistor's tolerance is to one extreme and the other to
the opposite extreme. This would result in their tolerances being an integer multiple with the
appropriate sign as the case may be
Therefore, Equation 6
becomes
⎡
⎛ R1 ⎞⎤
⎟⎥
⎝ R1 + R 2 ⎠⎦
αVOUT = ± ⎢αVREF + 2 ⋅ α R ⎜
⎣
where α R
= tolerance of resistors
Equation 7
This represents the worst case error using the calculated resistor values. Very often, the calculated value would not be exactly the same as a preferred value, resulting in additional deviation error. This deviation of actual resistor from calculated value needs to be accounted for which means
Equation 6 needs to be modified. This is done by defining another error term representing the
combined deviation of both resistors from their calculated values such that the overall error
would then be
⎡
⎤
α VOUT = ± ⎢α VREF + αRD + αRPE ⎥
⎣
⎦
where α RD = error due to preferred value and
Equation 8
α RPE = error due to tolerance of the preferred value
resistor
The task then is determining RD and RPE as follows.
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Consider the second term in the RHS of Equation 5. Each R can be assumed to consist of two
components namely,
1. An error due to choosing a preferred resistor value different from the calculated one, and
2. The basic tolerance of the preferred resistor itself.
Let RnC = Calculated value of Rn, and
RnP = Preferred value of Rn.
Let
α RnD =
RnP − RnC δRnC
=
RnC
RnC =Error due to changing from RnC to RnP,
and
α RnP = Tolerance of RnP.
Thus, from
Equation 5,
Substituting
Equation 10
into Equation 5
gives
Re-arranging
gives
⎛ δR1 δR 2 ⎞ ⎛ δR1C + α R1P ⋅ R1P
−
⎜
⎟≡⎜
R1C
⎝ R1 R 2 ⎠ ⎜⎝
⎞ ⎛ δR 2 C + α R 2 P ⋅ R 2 P
⎟⎟ − ⎜
⎜
R2 C
⎠ ⎝
⎛ δR1 δR 2 ⎞ ⎛⎜ δR1C α R1P ⋅ R1P
+
−
⎟=
⎜
R1C
⎝ R1 R 2 ⎠ ⎜⎝ R1C
⎞ ⎛ δR 2 C α R 2 P ⋅ R 2 P
⎟⎟ − ⎜⎜
+
R2C
⎠ ⎝ R2 C
α VOUT =
α VOUT =
δV REF
V REF
δV REF
V REF
⎛
R1C
+ ⎜⎜
R
1
⎝ C + R2 C
⎞⎛ ⎛ δR1C α R1P ⋅ R1P
⎟⎟⎜ ⎜⎜
+
⎜
R1C
⎠⎝ ⎝ R1C
⎛
R1C
+ ⎜⎜
R
1
⎝ C + R2 C
⎞⎛ δR1C δR 2 C
⎟⎟⎜⎜
−
R2 C
⎠⎝ R1C
⎛
R1C
R
1
⎝ C + R2 C
α VOUT = α VREF + ⎜⎜
Again assuming worst case conditions with
Equation 9
⎞
⎟⎟
⎠
Equation 10
⎞ ⎛ δR 2 C α R 2 P ⋅ R 2 P
⎟⎟ − ⎜⎜
+
R2 C
⎠ ⎝ R2 C
⎞ ⎛
R1C
⎟⎟ + ⎜⎜
R
1
⎠ ⎝ C + R2 C
⎞
⎛
R1C
⎟⎟(α R1D − α R 2 D ) + ⎜⎜
R
1
⎠
⎝ C + R2 C
α R1P
⎞
⎟
⎟
⎠
⎞⎞
⎟⎟ ⎟
⎟
⎠⎠
⎞⎛ α R1P ⋅ R1P α R 2 P ⋅ R 2 P
⎟⎟⎜⎜
−
R2 C
⎠⎝ R1C
⎞⎛ α R1P ⋅ R1P α R 2 P ⋅ R 2 P
⎟⎟⎜⎜
−
R2 C
⎠⎝ R1C
⎞
⎟⎟
⎠
⎞
⎟⎟
⎠
Equation 11
= α R 2P (= α RP ) but opposite in sign,
⎡
⎛ α ⋅ R1C
⎞
⎛
R1C
⎟⎟(α R1D − α R 2D ) + ⎜⎜ RP
αVOUT = ± ⎢αVREF + ⎜⎜
R
1
R
2
+
C ⎠
⎝ R1C + R 2C
⎝ C
⎣
⎞⎛ R1P R 2 P
⎟⎟⎜⎜
+
⎠⎝ R1C R 2C
⎞⎤
⎟⎟⎥
⎠⎦
Equation 12
Check:
If both calculated R1 and R2 are the same as the preferred values, the expression reduces to
⎡
⎛
R1C
⎝ R1C + R 2C
α VOUT = ± ⎢α VREF + 2 ⋅ α RP ⎜⎜
⎣
⎞⎤
⎟⎟⎥
⎠⎦
Same as
Equation
7Equation 7
Also, comparing Equation 12 and Equation 8, it can be seen that,
⎛
R1
⎞
C
⎟⎟(α R1D − α R 2D )
α RD = ⎜⎜
⎝ R1C + R 2C ⎠
and,
⎛ α ⋅ R1C
α RPE = ⎜⎜ RP
⎝ R1C + R 2C
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© Diodes Incorporated, 2008
⎞⎛ R1P R 2 P
⎟⎟⎜⎜
+
⎠⎝ R1C R 2C
⎞
⎟⎟
⎠
As required.
9
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In practice, one resistor (usually R2) is chosen by the user, effectively making its RC = RP, whilst
the other is calculated. Also, if the remaining resistor's RP is very close to its RC, as is more likely
than not, then is a valid assumption. Applying these facts, Equation 12 becomes,
⎡
⎛
R1C
R
1
⎝ C + R 2C
α VOUT = ± ⎢α VREF + α RD + 2 ⋅ α RP ⎜⎜
⎣
⎞⎤
⎟⎟⎥
⎠⎦
Equation 13
As required.
Summary
The cumulative error caused by variation in values of the component parts is given by Equation
12 which holds true under all conditions. However, in the special but realistic conditions given
preceding it, Equation 13 produces the same result as Equation 12 with negligible difference.
The latter is simpler and easier to remember than the former and is the one used in Calculated
Example 1.
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Intentionally left blank
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requirements with respect to the use of hazardous substances. Numerous successful programs have been implemented to reduce the use
of hazardous substances and/or emissions.
All Diodes Zetex components are compliant with the RoHS directive, and through this it is supporting its customers in their compliance
with WEEE and ELV directives.
Product status key:
“Preview”
Future device intended for production at some point. Samples may be available
“Active”
Product status recommended for new designs
“Last time buy (LTB)”
Device will be discontinued and last time buy period and delivery is in effect
“Not recommended for new designs” Device is still in production to support existing designs and production
“Obsolete”
Production has been discontinued
Datasheet status key:
“Draft version”
This term denotes a very early datasheet version and contains highly provisional information, which
may change in any manner without notice.
“Provisional version”
This term denotes a pre-release datasheet. It provides a clear indication of anticipated performance.
However, changes to the test conditions and specifications may occur, at any time and without notice.
“Issue”
This term denotes an issued datasheet containing finalized specifications. However, changes to
specifications may occur, at any time and without notice.
Diodes Zetex sales offices
Europe
Americas
Asia Pacific
Corporate Headquarters
Diodes Zetex GmbH
Kustermann-park
Balanstraße 59
D-81541 München
Germany
Telefon: (49) 89 45 49 49 0
Fax: (49) 89 45 49 49 49
[email protected]
Zetex Inc
700 Veterans Memorial Highway
Hauppauge, NY 11788
USA
Diodes Zetex (Asia) Ltd
3701-04 Metroplaza Tower 1
Hing Fong Road, Kwai Fong
Hong Kong
Diodes Incorporated
15660 N Dallas Parkway
Suite 850, Dallas
TX75248, USA
Telephone: (1) 631 360 2222
Fax: (1) 631 360 8222
[email protected]
Telephone: (852) 26100 611
Fax: (852) 24250 494
[email protected]
Telephone: (1) 972 385 2810
www.diodes.com
© 2008 Published by Diodes Incorporated
Issue 1 - September 2008
© Diodes Incorporated, 2008
12
www.zetex.com
www.diodes.com
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