AN1093

Application Note 1093
Application Notes for AP3765A System Solution
Prepared by Li Yunrong
System Engineering Dept.
1. Introduction
The AP3765A uses Pulse Frequency Modulation (PFM)
method to realize Discontinuous Conduction Mode (DCM)
operation for Flyback power supplies. The operating
principle of PFM is different with Pulse Width Modulation
(PWM), so the design of transformer is also different.
number of external system components. Fixed cable
compensation (6%) is used to adapt the voltage drop on
output cable and good CV regulation is achieved. Besides,
audio noise is reduced by the creative audio suppression
technique.
The AP3765A can provide accurate constant voltage (CV),
constant current (CC) regulation with Primary Side
Regulation (PSR) structure. It uses internal line
compensation and cable compensation to reduce the
1
The AP3765A is designed for driving bipolar transistor in
Flyback converter, with more driving current of about
40mA. With system parameters properly designed,
AP3765A can achieve standby power less than 150mW.
FR1
L1
T1
BG1
RST1
VINAC
C1
R2
Np
RST2
DS
Ns
R1
+
D1
CIN1
+
CIN2
COUT1
+
Da
Ra
+
COUT2
RDUMMY
CN1
5V/1.2A
Na
+
CVCC
L2
Q1
U1
AP3765A
VCC
CCPC
GND
RFB1
OUT
CPC
FB
CS
RFB2
RLINE
RCS
Figure 1. Typical Application Circuit of AP3765A
Figure 1 is the typical application circuit of AP3765A,
which is a conventional Flyback converter with a
3-winding transformer---primary winding (NP), secondary
winding (NS) and auxiliary winding (NA). The auxiliary
winding is used for providing VCC supply voltage for IC
and sensing the output voltage feedback signal to FB pin.
the parameters are defined as following.
Vdri---The driving signal of primary power switch
Ip---The primary side current
Is ---The secondary side current
IPK---Peak value of primary side current
IPKS---Peak value of secondary side current
VSEC---The transient voltage at secondary winding
VS---The stable voltage at secondary winding when
1
Figure 2 shows the typical waveforms which demonstrate
the basic operating principle of AP3765A application. And
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Application Note 1093
tONP ---The conduction time when primary side switch is
“ON”
tONS ---The conduction time when secondary side diode is
“ON”
tOFF ---The dead time when neither primary side switch nor
secondary side diode is “ON”
tOFFS --- The time when secondary side diode is “OFF”
rectification diode is in conducting status, which equals the
sum of output voltage VOUT and the forward voltage drop
of diode
VAUX---The transient voltage at auxiliary winding
VA--- The stable voltage at auxiliary winding when
rectification diode is in conducting status, which equals the
sum of voltage VCC and the forward voltage drop of
auxiliary diode
tSW ---The period of switching frequency
t SW
Vdri
IPK
IP
IPKS
t OFFS
IS
VA
VAUX
VS
VSEC
t ONP
tONS
t OFF
Figure 2. Operation Waveforms of Flyback PSR Control System
(1)
2. Guideline of System Design
t START = (R ST1 + R ST 2 ) ⋅ C vcc ⋅ VTH _ ST / VINDC _ MIN
1.
2.
3.
4.
5.
Where VTH_ST is the Startup Threshold of VCC, and
VINDC_MIN is the rectified DC voltage from the lowest AC
input.
Low Standby Power Design
Switching Frequency Design
Transformer and Power Devices Design
Feedback Resistors Design
Line Compensation Design
Besides, the selection of dummy load resistor is a tradeoff
between standby power and I-V curve. The recommended
value of dummy load resistor RDUMMY is 4.7kΩ to 10kΩ for
an application with 5V output voltage.
2.1 Low Standby Power Design
In order to achieve low standby power, AP3765A decreases
the minimum operating voltage. And the startup resistors
RST1+RST2 should be high enough to further lower the
power loss. However, there is a tradeoff between low
standby power PST and small startup time tSTART, which is
Oct. 2012
2.2 Switching Frequency Design
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Application Note 1093
As we know, in DCM Flyback converter, the stored energy
of primary side will be transferred to secondary side at the
time when the primary switch is turned off. And assume
the current transfer efficiency from primary to secondary
is ηi , then
2 ⋅ VDD
Vcpc
=
IO
N PS ⋅ ηi ⋅ I PK
If ηT is efficiency of power transmission from transformer
primary to the output, then
(2)
Ipks = Ipk ⋅ N PS ⋅ ηi
PO = VO ⋅ I O =
Here, NPS is the turn ratio of primary winding to secondary
winding.
t
1
Ipks ⋅ ONS
2
t SW
f SW
2 ⋅VO
=
2
IO
LP ⋅ I pk ⋅ ηT
(7)
(8)
(3)
When voltage at the sense resistor reaches the reference
voltage set by AP3765A, the switch will be turned off and
primary current reaches its maximum value,
Then,
Io =
1
2
⋅ LP ⋅ I pk ⋅ f SW ⋅ ηT
2
Where, fSW is the switching frequency. So,
It is obvious in Figure 2 that the output current “IO” is the
average current of secondary side “IS”,
Io =
(6)
t
1
Ipk ⋅ N PS ⋅ η i ⋅ ONS
2
t SW
(4)
I PK =
Always voltage of CPC pin (VCPC) is determined by,
t
Vcpc = VDD ⋅ ONS
t SW
Vcs _ ref
(9)
Rcs
When the constant reference VCS_REF is used, the peak
current IPK is constant. From formula (6) and (8), it is
obvious that VCPC and fSW increases linearly with the output
current IO.
(5)
Here VDD is a constant voltage generated by IC. Then,
VH=0.5V
VCS_REF
1.4V
VCPC
fSW
fSW
47.6kHz
20kHz
IO
42%IO
Figure 3. Relationship Between VCPC, fSW and IO at Constant Peak Current Mode
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Application Note 1093
decreased to 0.5V/1.5 when IO<42%* IO_MAX, as follows in
Figure 4.
In AP3765A, in order to realize audio noise suppression,
two-segmented of current reference voltage VCS_REF is used.
The reference is about 0.5V when IO>=42%*IO_MAX and is
VLOAD
VCPC
VH=0.5V
VCS_REF
VL=0.5V/1.5
0.42хIO_MAX
IO_MAX
fSW
55kHz
52kHz
fSW
23.1kHz
8.89kHz
3.95kHz
20kHz
0.42хIO_MAX
IO_MAX
ISOURCE
IO_MAX
Figure 4. Relationship Between VCPC, fSW and IO at Variable Peak Current Mode
Then from formula (6) and (8), we can see the VCPC and
fSW both has a leap at about 42% of maximum load. At the
leap point, if the peak current is decreased by 1.5 times, the
voltage of CPC pin at low IPK will be increased to 1.5 times,
and the switching frequency fSW at low IPK will be
increased to 1.52 times. So the load range in audio is
largely narrowed.
VCS_REF
VH=0.5V
VL=0.5V/1.5
39%IO
42%IO
IO
Figure 5. Hysteresis at Conversion Between Low IPK and High IPK
In order to avoid unstable operation, a hysteresis is added
at the conversion between low IPK and high IPK.
Considering the relationship between audio noise and flux
density of transformer, deltaB≤2500 gauss is better for
audio noise suppression.
the AP3765A can be up to 120kHz. But this is only the
limit of the IC; the finally designed maximum switching
frequency is determined by the tradeoff between the
efficiency,
mechanical
dimensions
and
thermal
performance.
The low limitation of maximum switching frequency is
given by audio noise suppression. And the upper limit of
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Application Note 1093
There is an oscillating signal on FB waveform after
secondary Schottky diode current decrease to zero, which
is caused by primary inductance and equivalent output
capacitance of primary switch. Then some margin is added
to tONS as
2.3 Transformer and Power Devices Design
In the design of AP3765A, constant current control
function will keep a fixed proportion between on-time tONS
and off-time tOFFS of rectifier D1 (in Figure 1) by
discharging or charging a capacitor embedded in the IC.
The fixed proportion is
tONS 1
=
t SW
2
tONS = I pks ⋅
Vs ⋅ Io =
1
1
I O = ⋅ I PKS = ⋅ N PS ⋅ ηi ⋅ I PK
k
k
(18)
Then,
t SW =
(12)
The turn ratio of transformer should be designed first,
which ensures the power converter operating in DCM
within the whole conditions,
tSW ≥ tONP + tONS
2
2 ⋅VS ⋅ I O
2
⋅ ηi
Lp ⋅ I pk
2 ⋅ VS ⋅ I O
(13)
Ls =
For the primary side current,
2
≥ I pks ⋅
Lp
Ls
⋅ 1.1 + I pk ⋅
Vs
Vindc_min
(20)
Lp
N PS
(21)
2
At full load, the system will work in the boundary of CC
regulation. IO can be given by formula (12)，the following
can be obtained,
Lp
(14)
N PS ≤ N PS _ MAX =
Where
LP is the inductance of primary winding.
Vindc is the rectified DC voltage of input.
When Vindc is the minimum value, the maximum tONP can
be obtained. So,
Vindc_min ⋅ ηi
VS
k
⋅ ( − 1.1)
2
(22)
Then designed turns ratio NPS should be no more than
NPS_MAX defined in formula (22).
2.3.2 Check Stress Voltage of Primary Side Switch and
Reverse Voltage of Secondary Diode
Lp
Vindc _ min
(19)
Relationship between inductance of primary side and
secondary side is,
As we know, if equation (13) is met at minimum input
voltage and full load, it can ensure that the power converter
operates in DCM in all conditions.
Vindc
2
L p ⋅ I pk
⋅ ηi
tONP , tONS and tSW in (13) are replaced with (15), (16) and
(19), then
2.3.1 Calculate Turn Ratio of Transformer (NPS)
(15)
If NPS is fixed by customer according to design step 2.3.1,
real stress voltage of primary side switch and reverse
voltage of secondary diode can be calculated.
For the secondary side current, LS is the inductance of
secondary winding, Vd is the forward voltage of secondary
diode.
Oct. 2012
1
1 Lp
1
⋅ Ls ⋅ Ipks 2 ⋅ fsw = ⋅
⋅ ( Ipk ⋅ N PS ⋅ ηi ) 2 ⋅ fsw = ⋅ Lp ⋅ Ipk 2 ⋅ fsw ⋅ ηi 2
2
2 N PS 2
2
(11)
Then the output constant-current value IO is
t ONP_MAX = I pk ⋅
(17)
From formula (4) and formula (16), we can get
2 ⋅ t SW
=4
tONS
t ONP = I pk ⋅
(16)
VS = VO + Vd
(10)
It is assumed
k=
LS
⋅ 1. 1
VS
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Application Note 1093
The maximum stress voltage of primary side switch is,
Vds _ switch = Vdc_spike + Vindc_max +
VS ⋅ N P
NS
Where, fSW was set by the user based on definite
requirement. Then, LP can be gotten by,
(23)
LP =
Be careful that the value of Vdc_spike is determined by the
snubber circuit design.
2 ⋅ PS
1
⋅ 2
I ⋅ f SW ηi
2
PK
(27)
2.3.5 Calculate the Turns of Primary, Secondary and
Auxiliary (NP, NS, NA)
Maximum reverse voltage of secondary side，
The turns of primary winding,
Vdr = VS +
Vindc_max ⋅ N S
NP
(24)
Np =
For Flyback converter design, higher turns ratio NPS brings
higher stress voltage of primary side switch, higher
transforming efficiency, and the lower reverse voltage of
secondary diode. Finally, in design of turns ratio NPS,
formula (22), (23) and (24) should be totally considered.
LP ⋅ I PK
LP ⋅ I PK
≥
Ae ⋅ ∆B Ae ⋅ B max
As NPS and NP are fixed, we can get NS by
NS =
NP
N PS
(29)
2.3.3 Calculate the Peak Current of Primary Side and
Current Sensed Resistor (IPK & RCS)
Turns of auxiliary winding is,
IPK can be calculated by the output current.
NA =
I pk =
k ⋅ IO
N PS ⋅ ηi
(25)
Considering the Volt-second balance between magnetizing
and de-magnetizing, the formula of duty cycle is
D max =
2.3.4 Calculate the Inductance of Primary Side---LP
( VO + Vd ) ⋅ N PS t ons
⋅
Vindc ⋅ η i
t sw
(31)
2.3.7 Check Reverse Voltage of Auxiliary Diode
The primary side inductance LP is relative with the stored
energy. LP should be big enough to store enough energy, so
that PO_MAX can be obtained from this system.
If NP and NA is fixed according to design step 2.3.5, real
reverse voltage of auxiliary diode can be calculated by
formula (32).
According to formula (18), the output power can be given
by,
Oct. 2012
(30)
After turn ratio of primary side and secondary side is
designed, the maximum duty cycle of primary side at low
line voltage can be calculated again.
So RCS can be obtained by formula (9) and selected with a
real value from the standard resistor series. We
recommended using 1% tolerance resistors for RCS. After
RCS is selected, IPK should be modified based on the
selected RCS.
1
2
2
⋅ L p ⋅ I pk
⋅ f SW ⋅ ηi
2
N S ⋅ VA
VS
2.3.6 Check the Maximum Duty Cycle of Primary Side
In AP3765A, 0.5V is an internal reference voltage. If the
sensed voltage VCS_REF reaches 0.5V, the power switch will
shut down and tONP will be ended.
PS = VS ⋅ I O =
(28)
Vdar = VA +
Vindc_max ⋅ N A
NP
(32)
(26)
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Application Note 1093
2.4 Feedback Resistors Design
Figure 6. Feedback Resistors Circuit
From above Figure 6,
Vo = VFB ⋅
(R FB1 + R FB 2 ) N S
⋅
− VD
R FB 2
NA
R FB1 Vo + VD
=
⋅ NA −1
RFB 2 N S ⋅ VFB
RFB2 are within 5kΩ to 100kΩ.
(33)
2.5 Line Compensation Design
The internal line compensation function in AP3765A is
shown in Figure 7. S1 is closed when the primary switch is
“ON”. The line voltage can be detected from the FB pin.
The detected voltage internally compensates the peak
current. So the line compensation is determined by RLINE.
In different applications, the value of RLINE is different.
(34)
Through adjusting RFB1 and RFB2, a suitable output voltage
can be achieved. The recommended values of RFB1 and
Figure 7. Line Compensation Circuit
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Application Note 1093
Figure 8. Waveform of FB Pin
Other setting by users:
Switching frequency: fSW=65kHz
Forward voltage of secondary diode: Vd=0.4V
Forward voltage of auxiliary diode: Vda=1.1V
VCC voltage: VCC=14V
Core_type: RM5 (Ae=23.7mm2), Bmax<3000GS
Vdc_spike=50V (with snubber circuit)
The negative voltage VN of FB pin (in Figure 8) is linear to
line voltage. The AP3765A samples VN to realize the line
compensation.
VN = Vindc ⋅
NA
R FB 2
⋅
N P R FB1 + R FB 2
(35)
The compensated voltage of line compensation (VCS_LINE)
can be calculated by the following formula,
VCS _ LINE
1
= VN ⋅
⋅ 0.8 ⋅ RLINE
670k
Design Steps:
1) Calculate turn ratio of transformer (NPS)
(36)
N PS ≤ N PS _ MAX =
This is designed to compensate the additional voltage of
VCS introduced by tdelay, which is the delay time of internal
drivers of IC and primary side switch.
Vindc_min ⋅ η i
VS
k
⋅ ( − 1.1) = 15.8
2
(39)
(40)
Vindc_min = Vinac_min ⋅ 2 − 40
(37)
Considering some margin for Flyback PSR control, we
choose NPS=15.5.
Then RLINE can be adjusted to achieve excellent line
2) Check stress voltage of primary side switch and
reverse voltage of secondary diode
regulation of output current.
According to formulas (23) (24) and the selected NPS,
proper power devices could be chosen.
Vdelta = Vindc ⋅
R LINE = (
t delay
t delay
Lp
LP
⋅ Rcs
⋅ R cs ) /(
NA
R FB 2
0.8
⋅
⋅
)
N P R FB1 + R FB 2 670k
(38)
Vds _ switch = Vdc_spike + Vindc_max +
Design Example (for 5V/1.2A application):
Vdr = VS +
Specification:
Input voltage: 85VAC to 265VAC
Output voltage @ cable: VO_CABLE=5V
Output current: IO=1.2A
Output voltage @ PCB: VO=5.13V, (AWG22 Cable, Length
of cable=100cm)
Oct. 2012
Vindc_max ⋅ N S
NP
VS ⋅ N P
= 510V < 700V
NS
(41)
= 29V < 40V (42)
3) Calculate the peak current of primary side and
current sense resistor (IPK & RCS)
I pk =
Rev. 1. 0
I pks
N PS ⋅ η i
=
k ⋅ IO
= 330mA
N PS ⋅ η i
(43)
BCD Semiconductor Manufacturing Limited
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Application Note 1093
RCS =
VCS
= 1.5 Ω
I pk
(44)
D=
4) Calculate the inductance of primary side---LP
LP =
2 ⋅ VS ⋅ I O
= 1.9 mH (45) 2
I ⋅ f SW ⋅ ηi
Vdar = VA +
2
PK
LP ⋅ I PK
LP ⋅ I PK
≥
= 89.8 T
Ae ⋅ ∆B Ae ⋅ B max
(49)
7) Check reverse voltage of auxiliary diode
8)
5) Calculate the turns of primary, secondary and
auxiliary (NP, NS, NA)
Np =
(VO + Vd ) ⋅ N PS ⋅ 0.4
= 0.49
Vindc ⋅ η i
Vindc_max ⋅ N A
NP
Feedback Resistors
RFB1 Vo + VD
=
⋅ N A − 1 = 2.56
RFB 2 N S ⋅ VFB
(46)
(50)
= 79V
(51)
RFB1=24.9kΩ, RFB2=9.85kΩ
We choose NP=93T
N
NS = P = 6 T
N PS
NA =
N S ⋅VA
= 16 T
VS
9)
Line Compensation Resistors
(47)
RLINE = (
(48)
tdelay
Lp
⋅ Rcs ) /(
NA
RFB 2
0.8
⋅
⋅
) = 3. 4 k Ω
N P RFB1 + RFB 2 670k
(52)
Where VO_NL=5V. Therefore, the output voltage at cable
terminal at full load is a little higher than the voltage at
no load.
6) Check the maximum duty cycle of primary side
The maximum duty cycle of primary side is calculated as
following:
Design Results Summary:
1.Maximum peak current of primary side and RCS
IPK
330
mA
Peak current of primary side
RCS
1.5
Ω
Current sensed resistor
2.Transformer
LP
1.90
mH
Inductance of primary side
NPS
15.5
Turn ratio of primary and secondary
NP
93
T
Turns of primary side
NS
6
T
Turns of secondary side
NA
16
T
Turns of auxiliary side
DMAX
0.49
Maximum duty cycle of primary side at VINDC=80V
3. Primary power switch and diode
Vds_switch
510
V
Voltage stress of primary power switch
Vdr
29
V
Maximum reverse voltage of secondary diode
Vdar
79
V
Maximum reverse voltage of auxiliary diode
4. Voltage feedback resistors
RFB1
24.9k
Ω
Feedback resistor at upside from auxiliary side to FB pin
RFB2
9.85k
Ω
Feedback resistor at downside from FB pin to GND
5. Line compensation resistor
RLINE
3.4k
Ω
Line compensation resistor
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Application Note 1093
3. Summary
In order to get good performance of AP3765A, it is
important to correctly design standby power, switching
frequency, transformer parameters, feedback resistance
and line compensation resistance. This application note
only gives a preliminary design guideline about these
aspects and considers ideal conditions, so some
parameters need to be adjusted slightly on the basis of the
calculated results.
4. Application of AP3765A with AP4340
FR1
L1
BG1
RST1
VINAC
C1
R2
Np
RST2
R1
+
CIN2
Ns
R3
AP4340
OUT VCC
D1
CIN1
DS
T1
R4
+
GND
+
COUT
Da
CN1
5V/1.2A
Ra
Na
+
CVCC
L2
+
Q1
U1
AP3765A
VCC
CCPC
RFB1
OUT
CPC
GND
FB
CS
RFB2
RLINE
RCS
Figure 9. Typical Application Circuit of AP3765A with AP4340
In Primary Side Regulation of AP3765A application, if
AP4340 is used at secondary side as the output voltage
regulator, excellent dynamic response and low standby
power can be achieved. When detecting the output
voltage lower than a certain level, the AP4340 outputs
periodical signals which will be coupled to auxiliary side
and detected by AP3765A. By fast response and
Oct. 2012
cooperation, AP4340 and AP3765A can effectively
improve the transient performance for Primary Side
Regulation power system. Besides, dummy load is not
needed at secondary side and as a result standby power
will be decreased. For more detailed operating principles,
please refer to Application Note of AP4340 (Application
Note 1078_BCD).
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