AN2

Application Note 2
Issue 2 February 1996
Portable Ni-Cd Battery Charger
Inexpensive Load Tolerant Step-Up Energy Source
David Bradbury
Many portable systems such as
radio/tape players and video recording
equipment are commonly powered by a
12V rechargeable Nickel-Cadmium
(Ni-Cd)battery pack, which rarely has the
capacity to power the system for more
than a few hours. It would be convenient
if these batteries could be recharged
from a readily available source such as
an automotive battery.
The circuit shown in Figure 1 was
designed to charge 12V, 2 ampere-hour
Ni-Cd battery packs from a 12V source,
but was made versatile enough to
charge packs in the range of 4.8 to 15.6V
to increase possible applications. To
charge batteries that may reach a higher
end voltage than the 12V supply input
(even 12V Ni-Cd packs reach an end
voltage of about 14.5V), a voltage
12V
T1
R1
1M
W1
W2
W3
W4
D3
+ve
BYV27-5
C1
100µF
25V
C4
100µF Output
25V
R2
R3
120
10
D1
-ve
D2
C2
Q1
1N4148
0.01µF
ZTX650
0V
Figure 1
Flyback Topology based Ni-Cd Charger.
AN2 - 1
D2 - BYV27-5
T1 DETAILS
Core
FX3437 With Gap/
Spacer of 0.08mm
R4
Former
DT2492
Winding order W2, W4, W3
39
then W1
W2
40T
30swg.
C3
W4
20T
30swg.
4700pF
W3
13T
36swg.
W1
12T
36swg.
Application Note 2
Issue 2 February 1996
converter of some kind is required. This
converter must be short and open circuit
protected, be able to operate efficiently
over a wide output voltage range and
must be insensitive to input voltage
variations.
A self oscillating flyback converter will
meet most of these requirements, being
particularly suited to operating over a
wide output voltage range. However, the
standard circuit needs modifications to
cope satisfactorily with short and open
circuit loads. These changes include a
special biasing circuit and an energy
recovery winding on the converter
transformer.
Circuit Operation
When power is first applied to the circuit,
a small bias current supplied by R1 via
w in ding W1 starts to turn on the
transistor Q1. This forces a voltage
across W2 and the positive feedback
given by the coupling of W1 and W2
causes the transistor to turn hard on,
applying the full supply voltage across
W2. The base drive voltage induced
across W1 makes the junction between
R1 and R2 become negative with respect
to the 0V supply, forward biasing diode
D1 to provide the necessary base current
to hold Q1 on.
With the transistor on, a magnetising
current builds up in W2 which eventually
saturates the ferrite core of transformer
T1. This results in a sudden increase on
the collector current flowing through Q1,
causing its collector-emitter voltage to
rise and thus reducing the voltage
across W2. With the positive feedback
given by windings W1 and W2, a falling
voltage across W2 causes Q1 to turn off
rapidly.
The current flowing in W2 now forces
the collector voltage of Q1 to swing
positive until restricted by transformer
output loading. During this “flyback”
period, the voltage induced across W4
forward biases the diode D3 to charge
the output capacitor C4. Energy stored in
the core of the transformer while the
transistor was on is dumped into this
capacitor which feeds the load. The
collector voltage of Q1 remains high
until the current flowing in W4 falls to
zero.The voltage then falls until the
positive feedback given by W1 causes
Q1 to turn on hard again to start the next
cycle of oscillation. If a load (Ni-Cd
battery) is not connected across C4, the
energy dumped into this capacitor will
charge it to an ever increasing voltage.
To restrict the maximum output voltage,
an extra transformer winding, W3, has
been added which will return stored
energy into the input supply via D2 if the
output exceeds 20V.
A resistor-capacitor network comprised
of R4 and C3 has been added to the
circuit to limit the turn off transient to
within the ratings of Q1. A second
network, R3 and C2, was added to
maintain the loop gain of the circuit
when the diode D1 is not conducting i.e.,
during start up and switching. Without
sufficient gain, the circuit will not
oscillate.
The capacitor C2 also has an important
effect on the operation of the converter
when its output is shorted. During the
conduction period of Q1, C2 is charged
to a negative voltage by winding W1,
and this charge remains during the
flyback period. This negative bias will
inhibit continuous oscillation unless the
transformer “rings” sufficiently at the
AN2 - 2
Application Note 2
Issue 2 February 1996
end of the flyback period to produce a
transient base drive voltage large
enough to overcome the bias. Since an
output voltage of at least 1.5V is required
to pro duc e s u ffic ien t tra ns former
ringing, a short circuit load causes the
converter to run in an intermittent mode,
consuming very little power.
Converter Design
The converter is to charge a 12V, 2Ah
battery pack, which has a recommended
charge current rating of 220mA. The
typical power source is a 12V car battery.
Firstly, a transformer core must be
chosen that will give the necessary
throughput without the need to operate
at an excessive frequency. The choice
will be controlled by the peak current
passed through winding W2 and the
inductance required. The efficiency of
the converter can be expected to be
around 75%, so with a 12V supply and
12V output load, the average supply
current will be:
transistor for a given output voltage,
resulting in a shorter flyback period.
Reducing the flyback period allows a
given output power to be achieved with
a small peak current in the switching
transistor, helping to minimise losses.
However this is at the expense of
requiring a higher voltage transistor. A
compromise duty cycle of 70% was
chosen for this design. This gives Ipeak as:
Ipeak =
The E-Line ZTX650 transistor will yield a
high gain at this current and so was
chosen as the switching transistor. To
k e e p t h e c o n v e r t e r i n a u d i b l e y et
minimize switching losses, an
oscillation frequency in the range of 20
to 50kHz was chosen. This gives a
transistor on time (current build up time)
o f 3 5µs t o 1 4µs respectively.The
inductance of transformer winding W2
can now be calculated using:
Lmax =
Io x Vo 0.22
Is =
=
= 0.29A
Vs x Eff 0.75
The actual supply current taken by the
converter will be a linear ramp from zero
to Ipeak followed by a period of no current
flow. The ratio of the ramp period to the
whole cycle period is the duty cycle.
Because of the simple current
waveform, once the duty cycle is
known, the peak current in W2 can be
calculated from the average supply
current. The duty cycle is dependent on
the input to output turns ratio of the
transformer. The smaller the number of
turns on the output winding, the higher
the flyback voltage across the switching
Is x 2
0.29 x 2
=
= 0.83A
0.75
Duty Cy.
Vs x Ton 12 x 35E−6
=
= 0.5 mH
Ipeak
0.83
Similarly, Lmin was calculated to be
approximately 0.2mH.
The energy storage capability of the
suitable RM range of transformer cores
are described in the form of Hanna
curves. These curves relate I2 x L, I x N
and core spacer. The I2 x L value that is
required for this transformer is 0.33E-3
to 0.17E-3. The smallest core in the RM
range will meet this specification. An RM
type FX3437 pair of cores with a 0.08mm
spacer will give an I2x L factor of 0.25E-3.
This factor is in the required range and
also corresponds to a pre-gapped RM
AN2 - 3
Application Note 2
Issue 2 February 1996
core type LA14376 which thus can be
used as a convenient substitute. The
inductance of W2 will be (0.25E-3/(0.832)
= 0.36mH, requiring 38.5 turns according
to the Hanna curves for this core and gap
(rounded to 40 turns).
The output winding W4 is determined
by:
W4 =
Vo x Toff x W2 12.7 x 30% x 40
=
Vs x Ton
12 x 70%
= 18.4, rounded to 20
Note the ratio of Ton to Toff was used
above.
The output voltage must be limited to
20V by winding W3 and so this gives:
W4 =
( Vs + VREC ) x W 4 12.7 x 20
=
20.7
( Vmax + Vrec)
= 12.2, rounded to 13
The base winding W1, is a compromise
between providing sufficient base
current for a low gain transistor
operating with minimum supplies, and
avoiding losses caused by overdriving
the
transistor
under
normal
circumstances. The transistor is required
to pass 0.83A peak and a minimum gain
device at low temperature will need
approximately 15mA to achieve this. A
base drive voltage of at least 1.4V is
needed to pass any current at all through
the biasing circuit adopted and a voltage
of twice this is desirable if the base
current is to be insensitive to supply
voltage variations. A base winding of 12
turns will give a drive of 3.3V with a
minimum supply of 10.5V. A base
resistor of 120Ω gives the required base
current from this drive.
Finally, the starting base bias resistor
value must be calculated. This resistor
must cause the circuit to have sufficient
gain to oscillate yet not cause excessive
power dissipation if oscillation does not
occur due to incorrect winding phasing
or some other fault. To oscillate, the loop
gain of the circuit must be greater than
one. The feedback gain is 12/40 or about
0.3, so the transistor must give a voltage
gain of at least 1/0.3 or 3.3. The transistor
voltage gain is mainly dependent on the
ratio of collector and emitter loads. The
small signal collector loading is the
result of the tuned circuits made by the
transformer windings and associated
capacitors, turning out to be of the order
of 2k. The emitter loading is the intrinsic
emitter resistance, given by re = 25E-3/Ie.
For a voltage gain of 3.3, re must be less
than 2k/3.3 or 600. The minimum emitter
current during start up must be more
than 25E-3/600 or 42µA. The hFE of the
ZTX650 transistor is not specified at such
low currents but it is not expected to be
less than 30, so this would set the
m i n i m u m b a s e c u r r e n t a t 1 . 5µA.
However, because of this uncertainty,
the bias was raised to 10µA to ensure
reliable starting. This value will only
cause a worst case power dissipation of
about 70mW in the transistor if the circuit
fails to oscillate under fault conditions.
Performance
Over the intended operational range, the
circuit was found to give an efficiency
exceeding 70%, providing a useful
output from a supply as low as 9V. Full
AN2 - 4
Application Note 2
Issue 2 February 1996
input and output characteristics of the
converter are given in Figures 2 and 3.
Figure 2 shows the output current given
by the circuit for various load voltages.
Note the output current at 12V is very
close to the design aims. This diagram
also shows the efficiency of the
converter when operating into these
loads. Figure 3 shows how the output
current given into a 12V load varies with
input supply voltage.
converter was designed to charge the
2Ah power pack in about 14 hours. At
this charge rate, these vented battery
packs will safely stand continuous over
charging. If the converter is used to
charge a different battery pack, Figure 2
should be used to find the output current
of the circuit which can then be used to
calculate the charge time necessary. In
cases where the charge rate is greater
than C/10 for vented cells or C/50 for
button cells, it is recommended that a
timer be included in the circuit to ensure
that accidental overcharging does not
occur.
The time required to fully charge the
load batteries will depend on their
voltage and ampere-hour capacity. The
.5
90
Normal Operation Mode
Output Current
80
70
Efficiency
.3
60
Efficiency (%)
Output Current (Amps)
.4
50
40
.20
30
.10
20
Short Circuit Operation
Mode Output Current
10
0
0
2
4
6
8
10
12
14
16
18
Output Voltage (Volts)
Figure 2
Output Current and Efficiency against Output Voltage for Flyback Converter.
AN2 - 5
Application Note 2
Issue 2 February 1996
.275
.25
90
Output Current
.225
80
70
Efficiency
.175
Efficiency (%)
Output Current (Amps)
.20
60
.15
50
.125
40
.10
30
.075
20
.05
10
7
8
9
10
11
12
13
14
15
Figure 3
Input Voltage (Volts)
Output Current and Efficiency against Input Voltage for Flyback Converter.
Partial Characterisation of ZTX650. Full characterisation available upon request.
Absolute Maximum Rating
Unit
ICM
6
A
IC
2
A
1
W
Parameter
Symbol
Peak Pulse Current
Continuous Collector Current
Power Dissipation
at Tamb =25°C
Parameter
Symbol
Min
Collector-Base
Breakdown Voltage
V (BR)CBO
Collector-Emitter
Breakdown Voltage
V (BR)CEO
Collector-Emitter
Saturation Voltage
V CE(sat)
Static Forward Current
Transfer Ratio
hFE
P tot
Typ
Unit
Conditions
60
V
IC=100µA
45
V
IC=10mA
V
V
IC=1A, IB=100mA
IC=2A, IB=200mA
IC=50mA, V CE=2V
IC=500mA, V CE=2V
IC=1A, V CE=2V
IC=2A, V CE=2V
0.12
0.23
70
100
80
40
200
200
170
80
AN2 - 6
Max
0.3
0.5
300
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